One problem is that older GPS receiver spec sheets give numbers with SA on,
even though SA has been off for quite some time, so their numbers will be
inherently higher.

Probably the most accurate GPS receiver today would be the Motorola M12M
Timing. I would go into more detail but I'm about to head out the door. I'm
sure others will chime in on this soon.

There were some posts a few days ago about some new papers that were
published, there is some good info on the various recent versions of the
Motorola receivers... Might want to check that out.

Jason

GPS

M12+.

the

available? I

Jack,

Jason is right. I think the M12M is currently at the top of the heap.
Rick Hambly and Tom Clark have developed some circuitry that knocks the
jitter down quite a lot.

Basically, for a "normal" timing receiver (please don't flame me guys, I
know there is other stuff out there) the jitter is dependent on the
receiver micro-controller's clock frequency. For instance, the old UT+
timing receiver had a jitter spec of +/- 45ns. If you were to compare
the short term 1PPS output from a UT+ to a rubidium, OCXO, or any
"stable" reference oscillator with a 1PPS output you would see that one
pulse might show up about 45ns ahead of the reference 1PPS output, the
next one would be about 45ns behind the reference, etc., etc......

The reason for this particular "jitter" of 90ns is because the UT+ used
a crystal running at about 11MHz, giving a period of about 90ns per
clock cycle. Since the receiver can only place 1PPS pulses with a 90ns
granularity it places the pulse as close to the what it thinks is the
actual UTC time tick. One time it will be on one side of the UTC tick,
the next time it may occur after the tick. This is where the term
"sawtooth" comes from. In one of the timing messages the receiver puts
out its best guess of how far off the NEXT pulse is going to be. If you
plot this data it looks roughly like a sawtooth as on average 50% of the
pulses are ahead of the time tick and the other 50% are behind it. I
have attached a screenshot from Winoncore12 showing this waveform.

Note that in this screenshot the jitter is about +/-15ns. That's because
the receiver I used for this plot is an M12+ timer. The reason for the
increased resolution is that the M12+T uses a 16.384MHz clock, and that
it can place the pulse on both the rising and falling edges of the
clock. This means that the M12+T has a 30ns granularity, resulting in a
+/-15ns amplitude of the sawtooth (jitter).

Note that these are all "perfect world" numbers, which is why they are
expressed as 1 Sigma values. Actual realizable performance ultimately
depends on how good the underlying code running the GPS receiver is.
This is why we all ultimately rely on the experts in the timing
community (many of whom are on this list) to test and publish "real
world" numbers, instead of some marketeer's spin on the performance of
the receiver he happens to hawking at the time.....

Sorry for the long post....

Randy Warner
Senior Applications Engineer
Synergy Systems, LLC
randy@synergy-gps.com

-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com] On
Behalf Of Jason Rabel
Sent: Wednesday, December 13, 2006 9:39 AM
To: 'Discussion of precise time and frequency measurement'
Subject: Re: [time-nuts] Best GPS 1PPS Accuracy

One problem is that older GPS receiver spec sheets give numbers with SA
on, even though SA has been off for quite some time, so their numbers
will be inherently higher.

Probably the most accurate GPS receiver today would be the Motorola M12M
Timing. I would go into more detail but I'm about to head out the door.
I'm sure others will chime in on this soon.

There were some posts a few days ago about some new papers that were
published, there is some good info on the various recent versions of the
Motorola receivers... Might want to check that out.

Jason

GPS

M12+.

the

available? I

time-nuts mailing list
time-nuts@febo.com
https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts

Mit freundlichen Grüssen

Paul Klöckler
Coral links Gmbh
Postfach 6955
CH-3001 Bern
Schweiz

-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com] On
Behalf Of Randy Warner
Sent: Mittwoch, 13. Dezember 2006 19:27
To: Discussion of precise time and frequency measurement
Subject: Re: [time-nuts] Best GPS 1PPS Accuracy

Jack,

Jason is right. I think the M12M is currently at the top of the heap.
Rick Hambly and Tom Clark have developed some circuitry that knocks the
jitter down quite a lot.

Basically, for a "normal" timing receiver (please don't flame me guys, I
know there is other stuff out there) the jitter is dependent on the receiver
micro-controller's clock frequency. For instance, the old UT+ timing
receiver had a jitter spec of +/- 45ns. If you were to compare the short
term 1PPS output from a UT+ to a rubidium, OCXO, or any "stable" reference
oscillator with a 1PPS output you would see that one pulse might show up
about 45ns ahead of the reference 1PPS output, the next one would be about
45ns behind the reference, etc., etc......

The reason for this particular "jitter" of 90ns is because the UT+ used a
crystal running at about 11MHz, giving a period of about 90ns per clock
cycle. Since the receiver can only place 1PPS pulses with a 90ns granularity
it places the pulse as close to the what it thinks is the actual UTC time
tick. One time it will be on one side of the UTC tick, the next time it may
occur after the tick. This is where the term "sawtooth" comes from. In one
of the timing messages the receiver puts out its best guess of how far off
the NEXT pulse is going to be. If you plot this data it looks roughly like a
sawtooth as on average 50% of the pulses are ahead of the time tick and the
other 50% are behind it. I have attached a screenshot from Winoncore12
showing this waveform.

Note that in this screenshot the jitter is about +/-15ns. That's because the
receiver I used for this plot is an M12+ timer. The reason for the increased
resolution is that the M12+T uses a 16.384MHz clock, and that it can place
the pulse on both the rising and falling edges of the clock. This means that
the M12+T has a 30ns granularity, resulting in a
+/-15ns amplitude of the sawtooth (jitter).

Note that these are all "perfect world" numbers, which is why they are
expressed as 1 Sigma values. Actual realizable performance ultimately
depends on how good the underlying code running the GPS receiver is.
This is why we all ultimately rely on the experts in the timing community
(many of whom are on this list) to test and publish "real world" numbers,
instead of some marketeer's spin on the performance of the receiver he
happens to hawking at the time.....

Sorry for the long post....

Randy Warner
Senior Applications Engineer
Synergy Systems, LLC
randy@synergy-gps.com

-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com] On
Behalf Of Jason Rabel
Sent: Wednesday, December 13, 2006 9:39 AM
To: 'Discussion of precise time and frequency measurement'
Subject: Re: [time-nuts] Best GPS 1PPS Accuracy

One problem is that older GPS receiver spec sheets give numbers with SA on,
even though SA has been off for quite some time, so their numbers will be
inherently higher.

Probably the most accurate GPS receiver today would be the Motorola M12M
Timing. I would go into more detail but I'm about to head out the door.
I'm sure others will chime in on this soon.

There were some posts a few days ago about some new papers that were
published, there is some good info on the various recent versions of the
Motorola receivers... Might want to check that out.

Jason

GPS

M12+.

the

available? I

time-nuts mailing list
time-nuts@febo.com
https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts

Hi Brendan,
I would contact Brooks Shera directly before you go any further as the span
of his controller is set at 4.5e-8 and the maximum span on an LPRO is about
5e-9. He can reprogram the controller by changing the filter gain and loop
time so it is about 8X more sensitive so it will work with a direct
connection to the LPRO. I have a controller modified with the filter gain is
scaled up 1 step and the loop time set to 120 seconds and this works well
with an LPRO attached directly to the DAC.
Enjoy!
Richard

----- Original Message -----
From: "Brendan Minish" ei6iz.brendan@gmail.com
To: "Discussion of precise time and frequency measurement"
time-nuts@febo.com
Sent: Wednesday, December 13, 2006 9:13 AM
Subject: [time-nuts] LPRO-101 with Brooks Shera's GPS locking circuit

If I'm totally missing something here please correct and enlighten me.

On the subject of Brooks Shera's design, the one thing that troubles me is the
use of a 24 MHz oscillator to count the width of the 1PPS signal.
This yields a precision of 4.16e-8, but does it really?
This oscillator is uncontrolled and any drift would exist as noise that would
have to be filtered (He uses a software low pass filter).

Question: Why not multiply the VCXO or OCXO output by 5 or 10 and run that into
24 or 32 bit counter? OR just sample the counter on every 10th PPS?

Thank,
Jack

-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com] On Behalf
Of Richard H McCorkle
Sent: Wednesday, December 13, 2006 4:44 PM
To: Discussion of precise time and frequency measurement
Subject: Re: [time-nuts] LPRO-101 with Brooks Shera's GPS locking circuit

Hi Brendan,
I would contact Brooks Shera directly before you go any further as the span
of his controller is set at 4.5e-8 and the maximum span on an LPRO is about
5e-9. He can reprogram the controller by changing the filter gain and loop
time so it is about 8X more sensitive so it will work with a direct
connection to the LPRO. I have a controller modified with the filter gain is
scaled up 1 step and the loop time set to 120 seconds and this works well
with an LPRO attached directly to the DAC.
Enjoy!
Richard

----- Original Message -----
From: "Brendan Minish" ei6iz.brendan@gmail.com
To: "Discussion of precise time and frequency measurement"
time-nuts@febo.com
Sent: Wednesday, December 13, 2006 9:13 AM
Subject: [time-nuts] LPRO-101 with Brooks Shera's GPS locking circuit

time-nuts mailing list
time-nuts@febo.com
https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts

the

No, with averaging it's much better than that.

would

No, when an oscillator is used as a timebase for what
is essentially a short period time interval counter the
XO drift rate does not affect the result like you think.

Suppose you use a cheap XO with a huge drift rate of
100 ppm per year or even 1 ppm per day to make TI
measurements between the OCXO and GPS. So an
average measurement that is, say 12.34 ns today,
will be off by 1 ppm tomorrow: it will be 12.34001 ns
instead. Do you see now why it doesn't matter how
bad the XO is?

Secondly, someone can double check me here -- but
it seems to me that any GPSDO that uses a built-in TIC
to monitor the deviation between the GPS 1PPS and
the OCXO 1PPS is a closed loop system and so the
actual accuracy of the TIC timebase has no effect on
the function of the GPSDO. I mean, the 24 MHz clock
could drift down to 20 MHz or up to 30 MHz and the
GPSDO would still work fine (hey, maybe even better).

/tvb

Tom Van Baak wrote:

Tom

Yes, frequency changes in the TIC oscillator only change the phase
detector gain, if the loop attempts to lock somewhere near zero phase
shift then the phase error at lock will be relatively unaffected by
phase detector gain changes of a few percent. However if the phase
detector gain changes by too much the loop dynamics will be compromised.
If the loop locks at say 90 degrees phase shift then changes in the
phase detector gain will affect the static phase error when the loop is
locked. Even then the frequency offset will be zero for a second order loop.

Bruce

Hi folks,

When i was a newbie in time nuts business the Shera design was the only
one available in amateur radio literature and i studied it in detail. It
were exactly these two questions that bothered me too. I would like to
explain my nowadays view of these two questions and by the way direct
your interest into some subtle details of the Shera design that not all
of you may be aware of:

In the Shera design the GPS's 1 pps is compared to a down-divided
version of the OCXO frequency. Shera explains "This (the comparison is
meant) can be done with less ambiguity if the measurement is done at a
frequency lower than 5 Mhz say 300 KHz". In his circuit the 5 MHz OCXO
signal is divided by 16 to give a frequency of 312.5 KHz. At 312.5 KHz
the ambiguity-free measurement range of the phase comparator is 1/312.5
KHz or 3.2 microseconds.

This small measurement range of the phase comparator is one of the
reasons for the complex procedure that is necessary to get the OCXO
locked for the first time because its frequency has to be close enough
to the setpoint that the phase difference to the GPS's 1pps stays long
enough within the measuring range of 3.2 microseconds to get it measured
ambiguity-free and to get the loop to lock. Since digital dividers are
cheap you may ask yourself why Shera did not use additional dividers to
get to even a lower frequency and a bigger ambiguity-free phase
measurement range which would have eased the oscillator lock setup. With
one of the signals being a 1pps, why not divide down the second signal
also to a 1pps raising the ambiguity-free measurement range of the phase
comparator to 500 ms (!) ?

The answer to this question is directly related to the properties of the
oscillator used for the time interval measurement. If you measure a time
interval of 3.2 microseconds length (or less) with a timebase having 24
MHz you may get 76 or 77 counts (or less) depending on the phase
relation. Now consider what circumstances are necessary to give you a
result of 75 or 78 counts. The period length of the 24 MHz oscillator
needs to change by more than +/- 1/77!!! That is almost 1.3 % or 13000
ppm!!! Even for a very simple packaged xtal oscillator having a tempco
of some ppm/K it is almost impossible that it changes its frequency by
13000 ppm due to environmental reasons. We see: If we measure short time
intervals where the period length of the timebase comes in the percent
range of the time interval to be measured the drift and noise of the
timebase are not of concern because they do not lead to a different
count value. That is the reason why the Shera design may use a cheap
canned xtal oscillator for the time interval measurement but is also the
reason why the Shera design depends heavily on measureing short times.

Due to what I have read in this newsgroup before I believe that not all
of you are aware of the influence of the measurement apparatus on time
stability measurements. Let me try to explain it with a thought
experiment:

Consider two perfect oscillators with no frequency and/or phase
fluctuations at all and a perfect time interval counter with infinite
measurement resolution and no measurement errors. I know, stuff like
this does not exist but it is a good idea to start with. With equipment
like this we would measure ALWAYS THE SAME phase delay between the two
oscillators. Let us call this time 't'. If we compute the Allan
Deviation from a number of identical values t we will always get a
result of zero regardless of the ovservation time Tau. That is
completely correct for the perfect scenario assumed.

Now consider the same situation with only one slight change: The time
interval counter shall not have a infinite resolution but shall be
limited to a certain resolution value. This value is the number that you
may find under 'single shot resolution' in the TIC's specs. Let us
assume a modern design like the Agilent 51131 universal counter. That
one has a single shot resolution of 500 ps. Let this number be
'delta_t'. If we now use this real world TIC to measure the phase delay
between the otherwise perfect oscillators we will sometimes get the
result 't' but - with some statistical probability - we will somtimes
get the result 't+delt_t' and also 't-delta_t'.

If we feed a number of these values into a Allen Deviation computation
the pure mathematics will of course NOT be aware of the fact that this
slight changes in the values are due to the measurement equipment.
Instead the mathematics will kind of 'believe' that the slight changes
are really due to the oscillators under test und will compute us
non-zero Allan Deviations for all observation times Tau. That is why we
can say that the simple fact (and only this!) that the measurement
equipment has a certain limited resolution this generates a 'noise
floor' that we are not able to measure Allan Deviations smaller than
that.

The amplitude of this noise floor is directly related to the time
resolution related to the times measured. Assume the phase delay between
two 1pps signals is measured with a TIC like this you get a noise floor
of 500ps/s = 5E-10. You will never be able to measure Allan Deviations
below 5E-10 @ 1s with a TIC like that!!! Now consider the case of the
famous HP5370 having a single shot resolution of 20 ps! A big
improvement but you will never be able to measure Allan Deviation below
2E-11 @ 1 s with this TIC!!! Good (for the amateur achieveable) xtal
oscillators feature Allan Deviations below 1E-12 @ 1s and (not so easy
achieveable) BVA resonator based oscillators may even be better than
that by an order of magnitude. I hope this has shown that most of us do
not own the necessary equipment to measure low Allan Deviations at short
observation times Tau.

In this situation the usual argument to be heard is: But i can take the
MEAN over some measurements and hereby improve the resolution a lot.
That is what it is done in the Shera design. By computing the mean over
30 s the resolution is improved by a factor of 30! Or it is not?

If there is no 'special rule' for when we measure 't' or 't+delt_t' or
't-delta_t' it is not unreasonable to assume that it is completely at
random when we measure which value. Perhaps I am going to simplify a lot
now, but effects that are 'at pure random' indeed tend to cancel out the
more measurement values one has available to compute the mean from. In a
Tau-Sigma-Diagram a effect that is 'at pure random' displays itself as
straight line having a slope of -1 and having a starting point that is
determined by the 'randomness' at the basic measurement interval. For
the 53131 its noise floor would be a straight line starting at 5E-10 @ 1
s and having a slope of -1.

For the TIC employed in the Shera design the noise floor is 4.2E-8 @ 1s
and it has slope of -1. And yes, you may run down this line for whatever
you want. Run to a observation time of 30 s and you get a value that is
a factor 30 more precise than the 1 second value, run to a observation
time of 1000 s and you get a value that is more precise by a factor of
1000 than the 1 second value, run to whatever you want and you will get
an improvement of whatever you want.

But is it really an improvement that you get out of it? The answer is
NO! He, why not? The answer is: Because you have to PAY the increase in
precision with the increase in observation time. For every increase of
10 in precision you need to increase the observation time by 10!

A real improvement would have been if the starting point of the line had
been lower than the 4.2E-8 @ 1s of the TIC employed in the Shera design.
Would the Shera design make use of a Agilent 51151 as a phase comparator
its noise floor would start at 5E-10 @ 1s which is a REAL improvement by
a factor of almost 100!!! For any given precision the Shera TIC will
need 100 X the time that the 51131 needs.

A lot of people may perhaps agree to this argument now. However they may
not immediately see how this effect really improves a frequency
standard. In order to understand this you have to get an idea on what
the Tau-Sigma-Diagram of an OXCO is. The Tau-Sigma-Diagram of a (good)
OCXO is a banana-like figure that starts at 1E-12 @ 1s, drops down to
say 3E-13 @ 10-100 s and increases from that. In a GPSDO we would draw
the Tau-Sigma of the OCXO alone into the diagram and the Tau-Sigma of
the GPS receiver into the same diagram. Where the -1 slope Tau-Sigma of
the receiver meets the banana one of the OCXO is a prominent point: We
need to make this the time constant of the loop because below this time
the OCXO has more stability than the GPS and above this point the GPS
has more stability than the OCXO. Below the loop's time constant the
frequency standard's stability is excusively determined by the LO and
above it it is exclusively determined by the GPS receiver.

Because the OCXO's banana like figure is already on its ASCENDING branch
where the lines meet each other it becomes clear immediatly that we want
to meet the lines AS EARLY AS POSSIBLE to make the overall stability at
whatever observation time as low as possible. That in conclusion is the
reason why we need a TIC measurement resolution that fits what is
available from a good GPS receiver. The 4.2E-8 @ 1s of the Shera design
does surely not fit the 2E-9 @ 1s resolution of a M12+ (including
sawtooth correction).

Everything written in this mail before has already been treated
absolutely correct by Dr. Bruce Griffiths on earlier occasions. I have
just tried to express it in a semi-scientifical language.

It depends on how fast the TIC's clock frequency changes. Every change
in clock frequency that is big enough to be noticed at all (and my first
point has shown that the clock frequency has to change by more 13000 ppm
for a change to be noticed) then the controller will measure a different
time interval between the LO and the GPS. This will lead the loop to
change the LO frequency until the original time interval is reached
again. After this equilibrium has being achieved again the LO will again
be on the correct setpoint. In so far TVB's argument is completely
correct: The absolute value of the TIC's clock frequency is of no
concern at all. However, changes in the clock frequency lead to time
limited changes in the LO frequency and it depends on the loop time
constant and some parameters more whether the so induced LO's frequency
changes stay within allowable bounds or not.

Best regards
Ulrich Bangert, DF6JB

Ulrich,

Thanks for the long contribution. One minor correction:
you imply that increase in observation time is a bad or
undesirable thing. This is usually true. But not really in
the case of a GPSDO. Due to GPS receiver 1PPS noise
you must average over many minutes anyway so this
greatly relaxes the requirements on the TIC.

What you say later about the sigma-tau lines is all
correct. I just wanted to point out, for example, that
a picosecond accurate TIC is a complete waste for
a GPSDO when the 1PPS jitter is on the order of
several nanoseconds.

Don't mislead yourself. At 1 s you are limited by GPS
1PPS noise. Having a better TIC doesn't fix this. If your
GPS noise is 2e-9 at 1 s you don't really need a TIC
that is good to 5e-10 at 1 s. So the gain isn't as useful
as you might think.

I should remind readers that the Shera design came
from an era of S/A and Oncore VP receivers where the
modern numbers you throw around do not apply.

/tvb

Tom Van Baak wrote:

Tom

A TIC with 0.5ns jitter at 1 second isn't actually too much in the way
of overkill when the PPS signal has 2ns of jitter.
If the TIC has 1ns of jitter and the TIC and PPS jitters are
statistically independent then the system jitter will be around 2.36ns
which is about 12% greater than the jitter of the PPS signal itself.
Conservative engineering practice usually dictates a jitter degradation
of no more than 5% corresponding to a TIC measurement jitter of about
1/3 the PPS jitter, that is a TIC jitter of 666ps  of less is advisable
when the  PPS jitter is 2ns.

If one's GPS timing receiver has a jitter of 2ns then the phase
detector/TIC used in the phase lock loop to discipline the OCXO/Rubidium
standard should have  subnanosecond jitter and resolution to avoid
significantly degrading the phase measurement jitter. This requirement
will (if one wishes to improve the performance of one's GPSDXO) become
even more stringent as the GPS system is upgraded and the Galileo system
becomes operational.

Thus devising inexpensive phase detectors/TICs with subnanosecond
performance allows one to take advantage of improvements in GPS timing
receiver performance when they occur.

The possibility of utilising GPS carrier phase tracking techniques in a
timing receiver offers a potential timing resolution and jitter in the
picosecond range which would allow enhanced  GPSDO performance.
Alternatively one could then achieve much better performance with less
expensive oscillators. Currently dual frequency GPS geodetic receivers
achieve subnanosecond resolution and stability when the data is
processed, albeit not in real time.

Bruce

On Thu, December 14, 2006 23:07, Dr Bruce Griffiths said:

I have had an idea for some time, even have the hardware pieces since a
year or more. Wish there were more time to spend on realizing projects...
:-( Its not very original, but I have not seen it explored in any GPSDOs.
Why not do the phase detection/frequency measurement inside the GPS
receiver?

Find a GPS that can be driven by your VCOCXO. (Zarlink's GP4020 accept
10MHz. CMCs Allstar and Superstar receivers are still available.) Control
the oscillator softly enough that the tracking loops will not unlock.

PLL augmented code measurement noise is in the low dm region for a good
receiver design. Using a known surveyed position this would give one sub
ns measurement for each satellite tracked. And then there is the phase
measurements that should be usable in some sense. Other information that
is available for a static receiver with internet connection is the
ultra-rapid ephemeris and clock-parameters that are available for
surveying use. These are much better than the broadcasted ephemeris. It
appears as if this concept would open performance enhancement
opportunities that are not used by the current OEM GPS timing receivers.

This structure would not need to generate the 1PPS from the GPS, and there
is no need for an external phase/TIC. It does instead ad an adaptable
amount of software complexity to solve the GPS time error outside the GPS.

What is the catch, that leads every(?) GPSDO designer along a different path?

--

Björn

bg@lysator.liu.se wrote:

Björn

You will need a dual frequency receiver to more accurately correct for the ionospheric delay.
Currently non approved users do not have access to the codes required to use the L2 frequency signals in an optimum way.
Various kludges are required to extract some info from the L1 frequency carrier phase if one doesn't have the despreading codes.
However this should change as the GPS system is enhanced to provide 2 or more frequencies to civilian users.

Multipath effects become even more critical as one attempts subnanosecond timing performance.
One may have to use either a choke ring ground plane antenna (as used in geodetic receivers) or resort to phased array techniques.

There is at least one commercially available GPS disciplined OCXO system
that uses carrier phase measurement techniques to discipline a crystal
oscillator.
A fractional standard deviation of 1E-11 for a measurement time of
around 1 second is claimed. When using carrier phase measurement
techniques it is advantageous to use the oscillator being disciplined to
generate all the receiver local oscillator frequencies as well as the
correlator clock frequencies.

Bruce

bg@lysator.liu.se wrote:

Björn

You will need a dual frequency receiver to more accurately correct for the ionospheric delay.
Currently non approved users do not have access to the codes required to use the L2 frequency signals in an optimum way.
Various kludges are required to extract some info from the L1 frequency carrier phase if one doesn't have the despreading codes.
However this should change as the GPS system is enhanced to provide 2 or more frequencies to civilian users.

Multipath effects become even more critical as one attempts subnanosecond timing performance.
One may have to use either a choke ring ground plane antenna (as used in geodetic receivers) or resort to phased array techniques.

There is at least one commercially available GPS disciplined OCXO system
that uses carrier phase measurement techniques to discipline a crystal
oscillator.
A fractional standard deviation of 1E-11 for a measurement time of
around 1 second is claimed. When using carrier phase measurement
techniques it is advantageous to use the oscillator being disciplined to
generate all the receiver local oscillator frequencies as well as the
correlator clock frequencies.

Bruce

From: Dr Bruce Griffiths bruce.griffiths@xtra.co.nz
Subject: Re: [time-nuts] LPRO-101 with Brooks Shera's GPS locking circuit
Date: Fri, 15 Dec 2006 12:55:19 +1300
Message-ID: 4581E467.8070706@xtra.co.nz

Bruce,

HP wrote a paper on that which I seem to have missplaced. Should be on my
laptop but I can't seem to find it. Their trick was to compare the code and
carrier trackings on standard Oncore receivers. There was some sound reasoning
behind it, but it doesn't handle the full error. I do not know if it ever made
it into any commercial product, but it should not supprice me. In effect it
used the fact that the ionospheric dispersion is such that the carrier delay
shift and code delay shift has opposite signs (and I beleive amplitude, but I
am not sure) for L1. They had made measurements to compare between sites and
relative cesiums. The improvement was there. I beleive it was a PTTI paper.

Otherwise you could go codeless to a combined L1 & L2 tracking using say
Z-tracking.

With now three GPS IIR-M sats in orbit, a modern receiver capable of L2C should
be able to do it. Good luck finding one on the second hand market. Actually
finding one cheap would be a challenge. I am tempted thought.

L1 choke rings can be found fairly cheaply. L1 and L2 is a bit harder. Also
notice that modern sats have a bigger footprint than they used to. Infact, your
ability to mitigate multipath improves with a higher bandwidth.

This was exactly what Björn and I was looking at. By replaing the 10 MHz TCXO
with a 10 MHz OVCXO and some additional circuit all the mixer frequencies,
sampling frequencies etc. becomes synchronous, the steering of the OCXO has the
goal of making the PPS output synchronous. Unfortunatly there is a little
design-bug in the correlator design which makes the PPS jump about in a 7
second cycle. However, this can be mitigated with external circuit.

BTW. Does anyone have FW for the CRC Allstar? I need it.

Cheers,
Magnus

From: bg@lysator.liu.se
Subject: Re: [time-nuts] LPRO-101 with Brooks Shera's GPS locking circuit
Date: Fri, 15 Dec 2006 00:21:25 +0100 (CET)
Message-ID: 2268.212.181.149.215.1166138485.squirrel@webmail.lysator.liu.se

The Internet aspect have not been fully adapted, but augmentation as such
isn't new. GPSes for mobiles could be augmented with information sent from the
BSC and can thus be adapted to the area for the BSC.

If you do your own processing of carrier and code information, which some
receiver allow, you can naturally shake out a different processing from it.

There are many uses for a good PPS even if it is not to be used for the GPSDO
as such.

The many choices available? :-)
The many ways to make a design cheaper?

Cheers,
Magnus

On Fri, December 15, 2006 0:55, Dr Bruce Griffiths said:

Sure, that is an improvement. But how large is really the time rate of
change of the ionosphere? (Depends on the solar activity naturally)

I do not think the kludges are not impacting GPS use on a good stationary
antenna location that much.

MP is the big remaining problem. What correlator types are really used in
OEM timing GPS receivers today? How "narrow" correlators do the different
receivers in the Oncore family use?

Chokering antennas for a few hundred dollars are available on ebay, so
thats  not outside the reach.

:-)

--

Björn

Bruce,

Can you clarify about the jitter, though. The TIC jitter
that was quoted (500 ps) is the single-shot resolution
for the 53131A. The "2 ns" M12+ jitter is an rms value,
no? The short-term or "single-shot" M12+ jitter, if you
could call it that, is more like +/- 20 ns. Averaging it,
over many minutes, gets you below 10 ns. Also the
sawtooth correction helps even further but that isn't
being done with Shera's board.

Maybe we're all agreeing even if with different words.

/tvb

Tom,

i believe that Bruce as well as me is always referring to what the
receiver CAN do i.e. not the raw but always the sawtooth corrected
signal. That is indeed 2 ns (1 sigma).

Thank you for clarifying this again! While i have been referring to the
measurement apparatus's noise floor for which my statements are correct,
one might indeed get into believing that every increase in resolution
leads to a increase in performance in a GPSDO. Clearly once that you are
below a certain point the GPS's jitter is the limiting number.

I second Bruces's opinion about what is an overshot or not. When ps
reolution is ready available then why not use it? I attach a online
output from my DIY GPSDO from a few minutes ago that shows the M12+'s
signal properties when measured with abt. 110 ps resolution against a
FTS1200. The yellow line reperesents a prefiltered version of the
sawtooth corrected values (blue). The filter time constant is 1/3 of the
loop time constant as in a PRS-10. The yellow values are the ones to
feed the regulation loop.

What I wanted to explain is the Shera concept noise floor is a factor 20
above what a modern receiver can deliver (again inc. the sawtoth
correction). And yes, you are right: There were different numbers when
this concept was thought out! And exactly because different number were
there when this concept was thougt out I am going to ask why people
still built it today.

Best regards
Ulrich Bangert, DF6JB

Hi Ulrich:

I think the answer is what other low cost options are available?  I
would like to have a more modern TIC capability to add to the clock I'm
working on.  But although there's been a lot of discussion about
different ways of making TIC measurements, it's not clear to me how to
do it on a budget.

For example the TIC232 circuit by Richard H McCorkle is easy to
implement, but how good is it's noise floor.  See:
http://www.piclist.com/techref/member/RHM-SSS-SC4/TIC232.htm

Then there's the low cost frequency counting TIC that appeared in QEX
that we know trades performance for low cost so it's not a candidate.

Any ideas on what circuits have a noise floor that's compatible with the
M12+T or it's newer equivalents and at the same time are in the low cost
category?

Have Fun,

Brooke Clarke

w/Java http://www.PRC68.com
w/o Java http://www.pacificsites.com/~brooke/PRC68COM.shtml
http://www.precisionclock.com

Ulrich Bangert wrote:

Brooke Clarke wrote:

Brooke

I agree that most will tend to use an available circuit particularly if
they are not too experienced/adventurous.

The noise/resolution of the TIC232 will be a little worse than that of
the Brooks Shera circuit.
It would appear to use the internal counter timer which is clocked at 16MHz.
Also this timer has no hardware for latching its count on the leading
edge of an external signal so there must be some software component used
to do this.
This will almost inevitably add extra noise/uncertainty due to
variations in the delay in reading the timer.
The quoted resolution of 1.04ns for a 1 minute average is probably
derived from a 62.5ns resolution for each individual measurement.

One can only achieve the subnanosecond resolution required to avoid
degrading the performance of an M12+ by using a clock frequency of 1GHz
or more.
Thus this method is probably too expensive and difficult to implement.

Perhaps there would be some demand for a higher resolution replacement
for the Brooks Shera system for those who have M12+ or equivalent
performance timing receivers and high performance OCXOs or Rubidium
standards who wish to achieve the best performance they can without
breaking the bank.

If so then perhaps we can collectively design such a system.
Breaking the task down into more manageable parts will help ensure that
the design is more quickly implemented

As I see it the following methods can achieve the desired phase
measurement resolution

1. Use a commercial TDC chip as the phase detector.
   Range 4millisec ( can be extended almost indefinitely by using a
   synchroniser and counter implemented in a gate array or its functional
   equivalent)
   Noise 65ps rms
   Cost ~ 100 euro
   Advantages someone has already designed and debugged the chip as long as
   the circuit layout recommendations are adhered to there should be no
   unforeseen problems.

2. Use an ADC to sample a sinewave formed by dividing down the OCXO
   frequency and filtering the resultant square wave
   Range half period of the sinewave frequency
   Noise (rms) < 0.0005 of the sinwave period (500ps with a 1MHz sinewave)
   Cost ~ $US20 ??

3. Use dual simultaneous sampling ADCs to sample quadrature phased
   sinewaves derived by dividing down the OCXO frequency filtering the
   resulting square wave and using a quadrature hybrid to produce the
   quadrature phase sinwave pair. Extend range to as much as 1 second or
   more using dual synchroniser to sample a continuously running digital
   counter/timer.
   Range to several days or centuries if required, depending on counter lenght
   Noise (rms) < 0.0005 of the sinewave period (500ps with a 1MHz sinewave)
   Cost ~ $US40 ??

Bruce

----- Original Message -----
From: "Ulrich Bangert" df6jb@ulrich-bangert.de
To: "'Discussion of precise time and frequency measurement'"
time-nuts@febo.com
Sent: Friday, December 15, 2006 05:47
Subject: Re: [time-nuts] LPRO-101 with Brooks Shera's GPS locking circuit

.......

I believe the sawtooth correction is of little or no value for a GPSDO,
which typically requires a low pass filter between the GPS 1pps and the
disciplined oscillator.  This filter is quite effective in removing the
sawtooth quantization introduced by the GPS rcvr clock, just as it removes
the similiar quantization caused by my phase detector.

For example, reading from your graph I averaged the raw data (as best I
could by reading the blue line).  The running average of the raw data over
40 sec (starting at 12:31:30) was -4.5 nsec, after 60 sec it was -4.2 nsec.
These values appear to be indistinguishable from the values you get by
averaging the "sawtooth corrected" data (the yellow line).

It appears from your plot that the sawtooth correction has contributed very
little or nothing that averaging does not already provide.  Have I
misunderstand something?

I believe that your "noise floor is a factor 20 above what a modern receiver
can deliver" statement is incorrect.  With an HP 5720B (about 100 psec
resolution), I have measured the phase difference between the GPS 1pps and
the phase of a 5 MHz oscillator controlled by my controller. This data has
been compared with simultaneous phase serial output from the controller as
determined its maligned 24 MHz asynchronous internal phase measurement
circuitry.

ADEV Stable 32 plots of both data sets are essentially identical.  From this
I conclude that nothing would be gained, for the purpose of discipling an
oscillator, by using a more elaborate and expensive phase detector  (the
phase detector in my controller costs $6.61, including $5.35 for the dual 24
MHz osc that is shared as the PIC clock).  It was my goal when I designed
the controller was to make the design transparent to the builder and to use
as few parts as necessary consistant with performance limited only by
available GPS receivers and VCXOs.  When I wrote the QST article I was
totally ignorant of the fact that I could buy the HP58503 with similiar
performance for $5400.

Your earlier comment about the capture range of the phase detector is well
taken.  For the past several years the PIC software I provide has included
an option designed for use with inexpensive TCVCXOs.  It requires only an
external 128 divider chip and produces EFC voltages suitable for inexpensive
oscillators.  It works very well and provides sufficient performance for
many applications.

Regards,  Brooks

Brooks Shera wrote:

Brooks

Your comparison of your circuit with measurements taken with the "5270"
(is this a typo? did you mean 5370? which is known to have  differential
non linearities well in excess  of  100 picoseconds,  at least according
to the designers - some later modifications  to the circuitry reduced
this effect somewhat) demonstrates very little unless the measurements
were corrected for the sawtooth error.

The only true test is to compare a sawtooth corrected GPSDOCXO alongside
a sawtooth corrected GPSDOXO. Both should of course use equivalent
performance oscillators and GPS timing receivers.

The short plot that Ulrich furnished doesn't include any hanging bridges
which occur when the GPS oscillator drifts through a harmonic of 1Hz.
Most M12+ GPS timing receivers produce sawtooth correction errors in
which such "hanging bridges" are not infrequent.

No one is disputing that with an low performance oscillator its not
worth going to too much trouble in improving the phase detector performance.
However some of us have oscillators with much better performance than
such cheap oscillators. We also have a need to achieve an oscillator
instability of less than a few parts in 1E12 which your circuit cannot
reliably provide in the presence of hanging bridges and aberrant PPS
pulses which do occur from time to time.

The existence of a commercial GPSDOCXO that achieves an Allan variance
of 2E-13 or better from tau = 1 sec to 1 year, indicates that it is
possible to do much better than your circuit is capable of. All we are
doing is exploring cheaper ways of approaching this performance within a
factor of 10 or so.

Bruce

Brooks, Brooke and Bruce,

1. I do not want to talk bad Brooks Shera's design. In fact i admire it
   a lot for its simpicity. It was the first to be published in amateur
   literature and that makes it easily the best available in amateur use
   for a long time. And I learned lots from it. Indeed i needed weeks to
   understand how some subtle ingredients go ahead hand in hand to make the
   whole thing work, the short measurement times that i talked about being
   one of them. The original QEX publication was surely a breakthrough in
   amateur technology.

2. One of the things that the original publication lacks is a in-depth
   rule on how to set the loop time constant correct for a given LO. When i
   was new into this topic it was kind of my belief that choosing this
   parameter correct was the 'black art' in constructing a good frequency
   standard and I wanted to learn more about it. Today i know, that only
   ONE SIMPLE RULE applies to this question despite the fact that some math
   for drawing tau-sigma-diagrams is indispensable.

3. This rule is: An OXCO has a banana like tau-sigma-diagram with the
   lowest ADEV anywhere between 10-100 s. A GPS receiver's
   tau-sigma-diagram is a straight line with a slope of -1 starting
   anywhere from ADEV 2E-8 @ 1 s to 1E-7 @ 1 s depending on the receiver.
   Note that these receiver figures apply to raw, not sawtooth corrected
   values. Now have a look to where the lines meet each other. Left from
   that point the OCXO's ADEV is smaller then the GPS receiver's. Right
   from that point the GPS receiver's ADEV is smaller the the OCXO's. There
   is no guessing or speculating at all: The loop time constant MUST be set
   to where the meeting point is. If it is set to anything else this will
   make the ADEV of the standard's output more worse than is necessary.
   Note that the OCXO's tau-sigma is already on its ASCENDING slope where
   the lines meet.

4. From that simple rule a complete briefing for the construction of a
   good frequency standard may be deduced:

a) Because left of the meeting point the standard's output stability is
only a function of the OCXO's stability and NOTHING ELSE choose the best
available LO in terms of ADEV up to the expected meeting point of the
lines. For this purpose a GOOD xtal oscillator may by all means be
better than a Rb! Perhaps the people that are going to discipline a Rb
with GPS may be disappointed: While the Rb is much easier to discipline
due to its smaller sensibility to environmental changes a good xtal
oscillator (the key word is: good. And good means: better than a
HP10811) may outperform a Rb based standard in terms of ADEV for short
observation times. @1 to some 10 s the HP10811 is better in ADEV than
most Rbs. However up from there its ADEV goes up steeper than that of an
thermically better managed USO like a FTS1000/1200. An even better
choice but beyond the financial scope of most of us were a BVA based
oscillator.

b) Because the meesting point is always on the the OCXO's ascending
slope choose the best available receiver in terms of how high it's -1
slope tau-sigma is located. The less high the absolute position of his
tau-sigma is, the more left (=earlier) the meeting point will be. The
more left the meeting point is the less the overall ADEV of the
standard's output will be deteriored by the OCXO for observation times
near the meeting point due to its ascending ADEV slope.

c) The TIC measurement resolution must be high enough to not deterior
the phase measurements by the sheer measurement 'granularity'.

Some graphics might be helpfull in understanding this. Have a look to
page 22 of

http://www.ulrich-bangert.de/AMSAT-Journal.pdf

which i wrote for the German AMSAT journal. Don't worry over the German,
just look to the pictures. In this graphic both the tau-sigma of a
HP10811 and a M12+ are drawn into the same diagram and according to 4)
it becomes immediatly clear why we want the OCXO as stable as possible
before the meeting point and the receivers tau-sigma as low as possible
to make the meeting point as early as possible.

Exactly this is the point where i fear that you, Brookes, are the victim
of a basic misconception, at least your comment makes me think so:

This indicates that you are believing that it can all be done with low
pass filtering. And this is wrong for two reasons:

a) As Bruce and TVB have pointed out there are 'anomalies' in a GPS
receiver's raw pps (well documented on TVB's web pages) where the idea
that lowpass filtering the raw phase data will do the job is simply
unsustainable.

b) Low pass filtering is a trade with nature: You can get better
precision due to low pass filtering but you have to pay for it in terms
of time that you have to wait for the samples to avarage over. Look
again at page 22 of

http://www.ulrich-bangert.de/AMSAT-Journal.pdf

and ask yourself what the noisefloor of you circuit would look like in
this diagram. I tell you: Even if you had the best current GPS receiver
available your phase measurements would be dominated by a noisefloor
induced by the 4E-8s single shot resolution of your TIC giving a
straight line starting at 4E-8 @ 1 s and having a slope of -1 i.e. a
line that runs parallel to the M12+ graph but a factor 2 higher in
absolute terms. Low pass filtering = averaging means nothing else than
running up and down the line. Go to any point of time on the horizontal
axis and draw a vertical line there. Where this line meets your
noisefloor draw a horizontal line and on the vertical axis read the
precision that you gain if you average over that time. It is as easy as
that. And to find out when you reach a certain precision go to that
precision on the vertical axix and draw a horizontal line. Where this
line meets your noisefloor draw a vertical line and read the necessary
averiging time on the horizontal axis. And note that this horizontal
line drawn in the last example has crossed the M12+'s line by a factor
of 2 earlier! That is: the sheer measurement resolution of 4E-8 s has
DOUBLED the averaging time necessary to come to a certain given
precision.

At a first glance this may be not so impressive: Instead of 10 s we have
to wait 20 s with your circuit to get the precision that the receiver
alone has already after 10 s. Why do I make that heck out of it? Don't
we have these 10 additional seconds? Please read on: The M12+'s
sigma-tau shown un the diagram is the one for the raw phase data. If the
sawtooth correction is taken into account the line starts at an ADEV of
2E-9 @ 1 s. Unfortunately its slope is less than -1 so the factor 10
increase in precision does not hold for all oservation times. At
observation times of app. 1 day the two lines meet, giving an
improvement in using the sawtooth corrected values only for observation
times < 1 day. In

http://www.ulrich-bangert.de/html/photo_gallery_44.html

you can see the sigma-taus for the raw and the corrected data from a
M12+ drawn into the same diagram. With a good OCXO the meeting point
between receiver tau-sigma and OCXO tau-sigma is in to order of 1000 s.
1000 s are small against a day, that means that almost the full possible
improvement in ADEV by using the sawtooth corrected values apply to the
case of a loop time constant of 1000. This factor of 10 in conjunction
with the factor of 2 that we had before results in the factor of 20 that
i claim that the noisefloor ot your circuit is inferior to that of a
modern GPS receiver. And of course my claim stays intact!

Some of you may now scratch your head and say: "Well...yes 20 is a
handfull! With the Shera circuit we will have to wait 20 times the time
that is necessary due to GPS 1pps jitter alone, but isn't it more
important that we reach this precision/stability (in a sense these two
words are synonyms in this discussion) AT ALL with the Shera circuit?"

This is EXACTLY where the misconception starts. If someone is claiming:
"I can average over 30s to get an improved measurement precision." I am
going to ask him: "Hey, why don't you average over 300 s, giving you an
additional factor 10 improvement." The answer might be: "Yes, perhaps I
could do. It depends.." My next question were: "Depends? Depends on
what? If every factor 10 in measurement averaging results in a factor 10
in measurement precision, why not even average over 30000 or 300000 s
??" I know the next answer very well in advance: "Oh no, i can't do
THAT. While the argument of improving the measurement precision is
right, i can't make use of this precision because my OCXO has drifted
too much away if I wait THIS long!" AAHH! You have to take your OCXO
into account? And yes, that is correct, but it is correct in a different
sense than you may think!

It is correct in the sense that i tried to explain before: The
tau-sigmas of the OCXO and the receiver meet each other and where they
meet depends ONLY on

a) receiver quality in terms of ADEV

b) OCXO quality in terms of ADEV

c) TIC's noise floor

In reality you are not free to choose "I want to average over 30 s" or
"I want to average over 100 s". Instead the simple rule DICTATES that
you HAVE to set your averaging time to the meeting point's x-axis value
and to nothing else. There is simply no use in saying: "But with such
and such averaging times i would reach a precision of such and such".
You cannot choose! The physical properties of your receiver, your ocxo
and your TIC dictate it!

Since we now know what 'averaging' is all about let us now consider
again at which ADEV the two tau-sigmas meet. Clearly we want to make the
ADEV at this point as small as possible as it represents a local maximum
in the overall tau-sigma of the standard's output. Since we are on the
ascending slope of the OCXO our interest must be that the lines meet AS
EARLY as possible. Since we cannot do anything on the -1 slope of the
receiver's tau-sigma we achieve this only by shifting the absolute
position of the tau-sigma as low as possible. This in turn is achieved
by using the best available receiver AND using the sawtooth correction.
A TIC resolution of 4E-8 shifts the meeting point a factor of 20 more to
the right than would be necessary with a good receiver. Since I admire
it a lot what you do, Brookes, i would be glad if you could gain the
insight that averaging over raw phase data is something VERY DIFFERENT
from using sowtooth corrected values.

Brooke, looking at the web page and the circuit diagram I second
everything that Bruce has already said to it. This one uses a 16 MHz TIC
time base and therefore its performance is even worse compared to
Brooks's circuit. This one has its tau-sigma starting point at 62E-9 @ 1
s, abt. 30 times worse than the M12+.

If it can be done 'on a budget' as you say depends a bit on what you
would call 'a budget' but it can surely not being done better if you
have the Shera design prices in your head! In my own DIY GPDSO I do it
using a delay chain out of the fastest interconnection elements
available in a ALTERA Flex10K10 gate array, giving 110 ps resolution.
That chip is surely not more than 50 US$ in single quantities.
Unfortunately the delay of a single element of this delay line depends
on chip temperature and supply voltage so that the lines need to be
'calibrated' on a cyclic base. While this is done in the controllers
firmware it makes the whole circuit a bit tricky. I currently try to
migrate the design into a XILINX Spartan III fpga XC3S400 worth 25 US$
in single quantities. Let us see what 2007 has to bring for us.

Bruce, the clue is NOT to go out for a high clock frequency. Instead
search for sub-clock interpolation schemes. Lots of them are available!

Best regards
Ulrich Bangert, DF6JB

In message 000001c7211b$d4766130$0202fea9@athlon, "Ulrich Bangert" writes:

Ulrich,

The real challenge is to build an algorithm which finds this point dynamically.

--
Poul-Henning Kamp      | UNIX since Zilog Zeus 3.20
phk@FreeBSD.ORG        | TCP/IP since RFC 956
FreeBSD committer      | BSD since 4.3-tahoe
Never attribute to malice what can adequately be explained by incompetence.

Poul,

i appreciate your comments always a lot! But dynamical methods are
especially usefull when the input parameters are subject of change,
aren't they? I have seen algorithms that use a a-priori knowledge of the
LO's ADEV properties to estimate the GPS's jitter and to adjust the loop
time constant according to that. Ok, if you are referring to something
like that. But how would you solve the system of equations with TWO
unknowns (LO and GPS jitter) if you have only ONE information?

Best regards
Ulrich Bangert, DF6JB

In message 000001c72126$ecc89270$0202fea9@athlon, "Ulrich Bangert" writes:

They are also very useful for amateur projects where the users do
not have the necessary measurement facilities and likely use random
components bought on ebay :-)

The way you do this is by measuring the ADEV between your two sources
and how it changes with changes in your timeconstant.

In my experience, the better way is to start with a short timeconstant
and increase it, until the ADEV shows signs of detoriation.

--
Poul-Henning Kamp      | UNIX since Zilog Zeus 3.20
phk@FreeBSD.ORG        | TCP/IP since RFC 956
FreeBSD committer      | BSD since 4.3-tahoe
Never attribute to malice what can adequately be explained by incompetence.