BK
Bob kb8tq
Tue, Jun 21, 2022 12:04 AM
Hi
Ok, single mixer phase noise basics:
First thing is to womp the mixer up to the point it almost smokes. Putting +7 dbm into
both ports on a “7 dbm” mixer is very normal in this case. Watching for the fact that the
mixer likely is not a 50 ohm load is part of the process ( = pads might help out) as well
as understanding that it does not have a monster amount of isolation ( = isolation amps
may be needed ).
Next one generates a beat note by offsetting the two oscillators a bit. This gives you a nice
360 degree sweep function ( 360 degrees per cycle :) ). From that you can work out the
system sensitivity in volts per degree ( or better yet per radian since that’s what you actually
want as the “magic number” …. it’s phase modulation so radian is king …).
Next you lock the two oscillators together via a DC feed out of the mixer to one or the
other of them. You adjust the “lock point” so that it is at zero volts out of the mixer. This
puts the two oscillators in quadrature. Yes, there is that messy 2X the input frequency RF
output and the inevitable leakage. Those are handled with a lowpass filter.
The output of the mixer is now “just noise”. There is no nasty carrier to deal with. There is
no messy fold over to wonder about. What you get is the DSB noise ( so both sides of
carrier) from the sum of the two oscillators.
Output of the mixer goes up if you terminate it in an “high” load. Something like 500 ohms
on a 50 ohm mixer or 5K ohms on an RPD-1 is often used. The isolation seems to be ok
either way and the added gain / better floor is “free”.
Simply put you add 3 db when you look at DSB if it’s uncorrelated, and another 3 db if the
oscillators are identical. Your “output” is 6 db higher than the single sideband / single oscillator
phase noise. You can argue that close in noise is likely correlated due to it being a modulation
on the carrier. The standard convention is to use 3 db.
Amplify the noise up and you can measure very low levels of phase noise. Low noise
audio op-amps are pretty easy to find spec sheets on. With anything these days finding
them on the shelf may be “interesting”. The OP-27 / OP-37 with low resistance in the
feedback path go way back for this application. There are a lot of other candidates.
The cutoff of the lock signal typically is adjustable to keep it below the lowest point
of interest for your noise testing. If that is impractical, there are ways to calibrate and
read “inside the loop”.
The HP 3048 phase noise analyzer was based on this approach. The original app note most
folks started from came from Fluke back in the early 1970’s. I have not (yet) found a good
copy of it on the internet.
Fun !!!
Bob
On Jun 20, 2022, at 9:43 AM, Erik Kaashoek via time-nuts time-nuts@lists.febo.com wrote:
Bob, Magnus,
Thanks, clear. A counter is for ADEV, not for phase noise.
I made a test setup to learn how to use the mixer/PLL approach.
First using 10MHz from both outputs of a DSS (Rigol DG990) to observe the DC shift with shifting the phase between the two signal.
Then by modulating one output with FM or PM.
There is a low pass filter after the mixer to get rid of the 10 MHz and its harmonics but the LPF is measured flat till about 10kHz.
The output signal from the mixer was kept within 10% of the full voltage swing to stay in the (hopefully) linear range.
Using PM creates a low frequency output from the mixer that is proportional to the phase shift (region 0-1 degree) and constant in amplitude with change of frequency. Also when using external modulation from an audio signal generator created the expected behavior with drive level and no frequency impact
Using FM with 0.1 Hz frequency deviation the mixer output amplitude decreases very fast with increasing frequency (range 0.1 to 10 Hz)
Also when using 1 Hz or more frequency deviation. The higher frequency deviation leads to higher output levels as expected.
Can someone help me understand how this FM signal (0.1 to 1000 Hz modulation and 0.1 to 1 Hz frequency deviation) translates to the calibration example mentioned in the document on phase noise measurement as linked by Bob. (0.1 Hz deviation at 1 kHz rate leading to a sideband (at 1kHz?) level of -86 dBc)
At a 1kHz rate I see (yet) no output from the mixer where at 1Hz there is a lot of output. Why is this output frequency dependency?
Is this a problem with the signal generator? Or the mixer?
Then I tried to use the modulated signal from the SG PLL locked to a 10MHz VCO. Results where the same. FM output signal is frequency dependent, PM not.
Erik.
time-nuts mailing list -- time-nuts@lists.febo.com
To unsubscribe send an email to time-nuts-leave@lists.febo.com
Hi
Ok, single mixer phase noise basics:
First thing is to womp the mixer up to the point it almost smokes. Putting +7 dbm into
both ports on a “7 dbm” mixer is very normal in this case. Watching for the fact that the
mixer likely is *not* a 50 ohm load is part of the process ( = pads might help out) as well
as understanding that it does not have a monster amount of isolation ( = isolation amps
may be needed ).
Next one generates a beat note by offsetting the two oscillators a bit. This gives you a nice
360 degree sweep function ( 360 degrees per cycle :) ). From that you can work out the
system sensitivity in volts per degree ( or better yet per radian since that’s what you actually
want as the “magic number” …. it’s phase modulation so radian is king …).
Next you lock the two oscillators together via a DC feed out of the mixer to one or the
other of them. You adjust the “lock point” so that it is at zero volts out of the mixer. This
puts the two oscillators in quadrature. Yes, there is that messy 2X the input frequency RF
output and the inevitable leakage. Those are handled with a lowpass filter.
The output of the mixer is now “just noise”. There is no nasty carrier to deal with. There is
no messy fold over to wonder about. What you get is the DSB noise ( so both sides of
carrier) from the sum of the two oscillators.
Output of the mixer goes up if you terminate it in an “high” load. Something like 500 ohms
on a 50 ohm mixer or 5K ohms on an RPD-1 is often used. The isolation seems to be ok
either way and the added gain / better floor is “free”.
Simply put you add 3 db when you look at DSB if it’s uncorrelated, and another 3 db if the
oscillators are identical. Your “output” is 6 db higher than the single sideband / single oscillator
phase noise. You can argue that close in noise is likely correlated due to it being a modulation
on the carrier. The standard convention is to use 3 db.
Amplify the noise up and you can measure very low levels of phase noise. Low noise
audio op-amps are pretty easy to find spec sheets on. With anything these days finding
them on the shelf may be “interesting”. The OP-27 / OP-37 with low resistance in the
feedback path go way back for this application. There are a lot of other candidates.
The cutoff of the lock signal typically is adjustable to keep it below the lowest point
of interest for your noise testing. If that is impractical, there are ways to calibrate and
read “inside the loop”.
The HP 3048 phase noise analyzer was based on this approach. The original app note most
folks started from came from Fluke back in the early 1970’s. I have not (yet) found a good
copy of it on the internet.
Fun !!!
Bob
> On Jun 20, 2022, at 9:43 AM, Erik Kaashoek via time-nuts <time-nuts@lists.febo.com> wrote:
>
> Bob, Magnus,
> Thanks, clear. A counter is for ADEV, not for phase noise.
> I made a test setup to learn how to use the mixer/PLL approach.
> First using 10MHz from both outputs of a DSS (Rigol DG990) to observe the DC shift with shifting the phase between the two signal.
> Then by modulating one output with FM or PM.
> There is a low pass filter after the mixer to get rid of the 10 MHz and its harmonics but the LPF is measured flat till about 10kHz.
> The output signal from the mixer was kept within 10% of the full voltage swing to stay in the (hopefully) linear range.
> Using PM creates a low frequency output from the mixer that is proportional to the phase shift (region 0-1 degree) and constant in amplitude with change of frequency. Also when using external modulation from an audio signal generator created the expected behavior with drive level and no frequency impact
> Using FM with 0.1 Hz frequency deviation the mixer output amplitude decreases very fast with increasing frequency (range 0.1 to 10 Hz)
> Also when using 1 Hz or more frequency deviation. The higher frequency deviation leads to higher output levels as expected.
> Can someone help me understand how this FM signal (0.1 to 1000 Hz modulation and 0.1 to 1 Hz frequency deviation) translates to the calibration example mentioned in the document on phase noise measurement as linked by Bob. (0.1 Hz deviation at 1 kHz rate leading to a sideband (at 1kHz?) level of -86 dBc)
> At a 1kHz rate I see (yet) no output from the mixer where at 1Hz there is a lot of output. Why is this output frequency dependency?
> Is this a problem with the signal generator? Or the mixer?
> Then I tried to use the modulated signal from the SG PLL locked to a 10MHz VCO. Results where the same. FM output signal is frequency dependent, PM not.
> Erik.
>
> _______________________________________________
> time-nuts mailing list -- time-nuts@lists.febo.com
> To unsubscribe send an email to time-nuts-leave@lists.febo.com
EK
Erik Kaashoek
Tue, Jun 21, 2022 9:26 AM
Bob, Using the approach you described I was able to verify the noise
floor of my initial measurement setup to be at -80dBc at 10Hz offset
from carrier and -100dBc above 1kHz with a strong peaks at 50Hz, 60Hz
and harmonics. This all with an RBW in the FFT of around 4Hz.
This level of noise is way to high to do phase noise measurement so I'm
now going to work on removing ground loops and adding low noise
amplification.
Erik.
On 21-6-2022 2:04, Bob kb8tq wrote:
Hi
Ok, single mixer phase noise basics:
First thing is to womp the mixer up to the point it almost smokes. Putting +7 dbm into
both ports on a “7 dbm” mixer is very normal in this case. Watching for the fact that the
mixer likely is not a 50 ohm load is part of the process ( = pads might help out) as well
as understanding that it does not have a monster amount of isolation ( = isolation amps
may be needed ).
Next one generates a beat note by offsetting the two oscillators a bit. This gives you a nice
360 degree sweep function ( 360 degrees per cycle :) ). From that you can work out the
system sensitivity in volts per degree ( or better yet per radian since that’s what you actually
want as the “magic number” …. it’s phase modulation so radian is king …).
Next you lock the two oscillators together via a DC feed out of the mixer to one or the
other of them. You adjust the “lock point” so that it is at zero volts out of the mixer. This
puts the two oscillators in quadrature. Yes, there is that messy 2X the input frequency RF
output and the inevitable leakage. Those are handled with a lowpass filter.
The output of the mixer is now “just noise”. There is no nasty carrier to deal with. There is
no messy fold over to wonder about. What you get is the DSB noise ( so both sides of
carrier) from the sum of the two oscillators.
Output of the mixer goes up if you terminate it in an “high” load. Something like 500 ohms
on a 50 ohm mixer or 5K ohms on an RPD-1 is often used. The isolation seems to be ok
either way and the added gain / better floor is “free”.
Simply put you add 3 db when you look at DSB if it’s uncorrelated, and another 3 db if the
oscillators are identical. Your “output” is 6 db higher than the single sideband / single oscillator
phase noise. You can argue that close in noise is likely correlated due to it being a modulation
on the carrier. The standard convention is to use 3 db.
Amplify the noise up and you can measure very low levels of phase noise. Low noise
audio op-amps are pretty easy to find spec sheets on. With anything these days finding
them on the shelf may be “interesting”. The OP-27 / OP-37 with low resistance in the
feedback path go way back for this application. There are a lot of other candidates.
The cutoff of the lock signal typically is adjustable to keep it below the lowest point
of interest for your noise testing. If that is impractical, there are ways to calibrate and
read “inside the loop”.
The HP 3048 phase noise analyzer was based on this approach. The original app note most
folks started from came from Fluke back in the early 1970’s. I have not (yet) found a good
copy of it on the internet.
Fun !!!
Bob
On Jun 20, 2022, at 9:43 AM, Erik Kaashoek via time-nuts time-nuts@lists.febo.com wrote:
Bob, Magnus,
Thanks, clear. A counter is for ADEV, not for phase noise.
I made a test setup to learn how to use the mixer/PLL approach.
First using 10MHz from both outputs of a DSS (Rigol DG990) to observe the DC shift with shifting the phase between the two signal.
Then by modulating one output with FM or PM.
There is a low pass filter after the mixer to get rid of the 10 MHz and its harmonics but the LPF is measured flat till about 10kHz.
The output signal from the mixer was kept within 10% of the full voltage swing to stay in the (hopefully) linear range.
Using PM creates a low frequency output from the mixer that is proportional to the phase shift (region 0-1 degree) and constant in amplitude with change of frequency. Also when using external modulation from an audio signal generator created the expected behavior with drive level and no frequency impact
Using FM with 0.1 Hz frequency deviation the mixer output amplitude decreases very fast with increasing frequency (range 0.1 to 10 Hz)
Also when using 1 Hz or more frequency deviation. The higher frequency deviation leads to higher output levels as expected.
Can someone help me understand how this FM signal (0.1 to 1000 Hz modulation and 0.1 to 1 Hz frequency deviation) translates to the calibration example mentioned in the document on phase noise measurement as linked by Bob. (0.1 Hz deviation at 1 kHz rate leading to a sideband (at 1kHz?) level of -86 dBc)
At a 1kHz rate I see (yet) no output from the mixer where at 1Hz there is a lot of output. Why is this output frequency dependency?
Is this a problem with the signal generator? Or the mixer?
Then I tried to use the modulated signal from the SG PLL locked to a 10MHz VCO. Results where the same. FM output signal is frequency dependent, PM not.
Erik.
time-nuts mailing list -- time-nuts@lists.febo.com
To unsubscribe send an email to time-nuts-leave@lists.febo.com
Bob, Using the approach you described I was able to verify the noise
floor of my initial measurement setup to be at -80dBc at 10Hz offset
from carrier and -100dBc above 1kHz with a strong peaks at 50Hz, 60Hz
and harmonics. This all with an RBW in the FFT of around 4Hz.
This level of noise is way to high to do phase noise measurement so I'm
now going to work on removing ground loops and adding low noise
amplification.
Erik.
On 21-6-2022 2:04, Bob kb8tq wrote:
> Hi
>
> Ok, single mixer phase noise basics:
>
> First thing is to womp the mixer up to the point it almost smokes. Putting +7 dbm into
> both ports on a “7 dbm” mixer is very normal in this case. Watching for the fact that the
> mixer likely is *not* a 50 ohm load is part of the process ( = pads might help out) as well
> as understanding that it does not have a monster amount of isolation ( = isolation amps
> may be needed ).
>
> Next one generates a beat note by offsetting the two oscillators a bit. This gives you a nice
> 360 degree sweep function ( 360 degrees per cycle :) ). From that you can work out the
> system sensitivity in volts per degree ( or better yet per radian since that’s what you actually
> want as the “magic number” …. it’s phase modulation so radian is king …).
>
> Next you lock the two oscillators together via a DC feed out of the mixer to one or the
> other of them. You adjust the “lock point” so that it is at zero volts out of the mixer. This
> puts the two oscillators in quadrature. Yes, there is that messy 2X the input frequency RF
> output and the inevitable leakage. Those are handled with a lowpass filter.
>
> The output of the mixer is now “just noise”. There is no nasty carrier to deal with. There is
> no messy fold over to wonder about. What you get is the DSB noise ( so both sides of
> carrier) from the sum of the two oscillators.
>
> Output of the mixer goes up if you terminate it in an “high” load. Something like 500 ohms
> on a 50 ohm mixer or 5K ohms on an RPD-1 is often used. The isolation seems to be ok
> either way and the added gain / better floor is “free”.
>
> Simply put you add 3 db when you look at DSB if it’s uncorrelated, and another 3 db if the
> oscillators are identical. Your “output” is 6 db higher than the single sideband / single oscillator
> phase noise. You can argue that close in noise is likely correlated due to it being a modulation
> on the carrier. The standard convention is to use 3 db.
>
> Amplify the noise up and you can measure very low levels of phase noise. Low noise
> audio op-amps are pretty easy to find spec sheets on. With anything these days finding
> them on the shelf may be “interesting”. The OP-27 / OP-37 with low resistance in the
> feedback path go way back for this application. There are a lot of other candidates.
>
> The cutoff of the lock signal typically is adjustable to keep it below the lowest point
> of interest for your noise testing. If that is impractical, there are ways to calibrate and
> read “inside the loop”.
>
> The HP 3048 phase noise analyzer was based on this approach. The original app note most
> folks started from came from Fluke back in the early 1970’s. I have not (yet) found a good
> copy of it on the internet.
>
> Fun !!!
>
> Bob
>
>
>
>> On Jun 20, 2022, at 9:43 AM, Erik Kaashoek via time-nuts <time-nuts@lists.febo.com> wrote:
>>
>> Bob, Magnus,
>> Thanks, clear. A counter is for ADEV, not for phase noise.
>> I made a test setup to learn how to use the mixer/PLL approach.
>> First using 10MHz from both outputs of a DSS (Rigol DG990) to observe the DC shift with shifting the phase between the two signal.
>> Then by modulating one output with FM or PM.
>> There is a low pass filter after the mixer to get rid of the 10 MHz and its harmonics but the LPF is measured flat till about 10kHz.
>> The output signal from the mixer was kept within 10% of the full voltage swing to stay in the (hopefully) linear range.
>> Using PM creates a low frequency output from the mixer that is proportional to the phase shift (region 0-1 degree) and constant in amplitude with change of frequency. Also when using external modulation from an audio signal generator created the expected behavior with drive level and no frequency impact
>> Using FM with 0.1 Hz frequency deviation the mixer output amplitude decreases very fast with increasing frequency (range 0.1 to 10 Hz)
>> Also when using 1 Hz or more frequency deviation. The higher frequency deviation leads to higher output levels as expected.
>> Can someone help me understand how this FM signal (0.1 to 1000 Hz modulation and 0.1 to 1 Hz frequency deviation) translates to the calibration example mentioned in the document on phase noise measurement as linked by Bob. (0.1 Hz deviation at 1 kHz rate leading to a sideband (at 1kHz?) level of -86 dBc)
>> At a 1kHz rate I see (yet) no output from the mixer where at 1Hz there is a lot of output. Why is this output frequency dependency?
>> Is this a problem with the signal generator? Or the mixer?
>> Then I tried to use the modulated signal from the SG PLL locked to a 10MHz VCO. Results where the same. FM output signal is frequency dependent, PM not.
>> Erik.
>>
>> _______________________________________________
>> time-nuts mailing list -- time-nuts@lists.febo.com
>> To unsubscribe send an email to time-nuts-leave@lists.febo.com
BK
Bob kb8tq
Tue, Jun 21, 2022 5:31 PM
Hi
With an audio spectrum analyzer, and an RPD-1 mixer, and +10 dbm on
each side of the mixer …. you should be able to get away with about 20 db
of “preamp gain”. Yes, that’s dependent on exactly what analyzer you have.
The typical sound card may well need closer to 50 db of gain. That’s better
done with two op-amp stages than trying to get it all in one shot. Again there’s
an assumption involved about wanting 40 to 100 KHz sort of “top end” for the
measurement.
Ground loops ( and power line noise ) are indeed a very big thing in all these
measurements. ADEV / phase noise / whatever. Being able to see the crud makes
getting rid of it much easier. That’s not to in any way to say it’s easy, only that it’s
easier.
Fun !!!
Bob
On Jun 21, 2022, at 1:26 AM, Erik Kaashoek erik@kaashoek.com wrote:
Bob, Using the approach you described I was able to verify the noise floor of my initial measurement setup to be at -80dBc at 10Hz offset from carrier and -100dBc above 1kHz with a strong peaks at 50Hz, 60Hz and harmonics. This all with an RBW in the FFT of around 4Hz.
This level of noise is way to high to do phase noise measurement so I'm now going to work on removing ground loops and adding low noise amplification.
Erik.
On 21-6-2022 2:04, Bob kb8tq wrote:
Hi
Ok, single mixer phase noise basics:
First thing is to womp the mixer up to the point it almost smokes. Putting +7 dbm into
both ports on a “7 dbm” mixer is very normal in this case. Watching for the fact that the
mixer likely is not a 50 ohm load is part of the process ( = pads might help out) as well
as understanding that it does not have a monster amount of isolation ( = isolation amps
may be needed ).
Next one generates a beat note by offsetting the two oscillators a bit. This gives you a nice
360 degree sweep function ( 360 degrees per cycle :) ). From that you can work out the
system sensitivity in volts per degree ( or better yet per radian since that’s what you actually
want as the “magic number” …. it’s phase modulation so radian is king …).
Next you lock the two oscillators together via a DC feed out of the mixer to one or the
other of them. You adjust the “lock point” so that it is at zero volts out of the mixer. This
puts the two oscillators in quadrature. Yes, there is that messy 2X the input frequency RF
output and the inevitable leakage. Those are handled with a lowpass filter.
The output of the mixer is now “just noise”. There is no nasty carrier to deal with. There is
no messy fold over to wonder about. What you get is the DSB noise ( so both sides of
carrier) from the sum of the two oscillators.
Output of the mixer goes up if you terminate it in an “high” load. Something like 500 ohms
on a 50 ohm mixer or 5K ohms on an RPD-1 is often used. The isolation seems to be ok
either way and the added gain / better floor is “free”.
Simply put you add 3 db when you look at DSB if it’s uncorrelated, and another 3 db if the
oscillators are identical. Your “output” is 6 db higher than the single sideband / single oscillator
phase noise. You can argue that close in noise is likely correlated due to it being a modulation
on the carrier. The standard convention is to use 3 db.
Amplify the noise up and you can measure very low levels of phase noise. Low noise
audio op-amps are pretty easy to find spec sheets on. With anything these days finding
them on the shelf may be “interesting”. The OP-27 / OP-37 with low resistance in the
feedback path go way back for this application. There are a lot of other candidates.
The cutoff of the lock signal typically is adjustable to keep it below the lowest point
of interest for your noise testing. If that is impractical, there are ways to calibrate and
read “inside the loop”.
The HP 3048 phase noise analyzer was based on this approach. The original app note most
folks started from came from Fluke back in the early 1970’s. I have not (yet) found a good
copy of it on the internet.
Fun !!!
Bob
On Jun 20, 2022, at 9:43 AM, Erik Kaashoek via time-nuts time-nuts@lists.febo.com wrote:
Bob, Magnus,
Thanks, clear. A counter is for ADEV, not for phase noise.
I made a test setup to learn how to use the mixer/PLL approach.
First using 10MHz from both outputs of a DSS (Rigol DG990) to observe the DC shift with shifting the phase between the two signal.
Then by modulating one output with FM or PM.
There is a low pass filter after the mixer to get rid of the 10 MHz and its harmonics but the LPF is measured flat till about 10kHz.
The output signal from the mixer was kept within 10% of the full voltage swing to stay in the (hopefully) linear range.
Using PM creates a low frequency output from the mixer that is proportional to the phase shift (region 0-1 degree) and constant in amplitude with change of frequency. Also when using external modulation from an audio signal generator created the expected behavior with drive level and no frequency impact
Using FM with 0.1 Hz frequency deviation the mixer output amplitude decreases very fast with increasing frequency (range 0.1 to 10 Hz)
Also when using 1 Hz or more frequency deviation. The higher frequency deviation leads to higher output levels as expected.
Can someone help me understand how this FM signal (0.1 to 1000 Hz modulation and 0.1 to 1 Hz frequency deviation) translates to the calibration example mentioned in the document on phase noise measurement as linked by Bob. (0.1 Hz deviation at 1 kHz rate leading to a sideband (at 1kHz?) level of -86 dBc)
At a 1kHz rate I see (yet) no output from the mixer where at 1Hz there is a lot of output. Why is this output frequency dependency?
Is this a problem with the signal generator? Or the mixer?
Then I tried to use the modulated signal from the SG PLL locked to a 10MHz VCO. Results where the same. FM output signal is frequency dependent, PM not.
Erik.
time-nuts mailing list -- time-nuts@lists.febo.com
To unsubscribe send an email to time-nuts-leave@lists.febo.com
Hi
With an audio spectrum analyzer, and an RPD-1 mixer, and +10 dbm on
each side of the mixer …. you should be able to get away with about 20 db
of “preamp gain”. Yes, that’s dependent on exactly what analyzer you have.
The typical sound card may well need closer to 50 db of gain. That’s better
done with two op-amp stages than trying to get it all in one shot. Again there’s
an assumption involved about wanting 40 to 100 KHz sort of “top end” for the
measurement.
Ground loops ( and power line noise ) are indeed a *very* big thing in all these
measurements. ADEV / phase noise / whatever. Being able to see the crud makes
getting rid of it much easier. That’s not to in any way to say it’s easy, only that it’s
easier.
Fun !!!
Bob
> On Jun 21, 2022, at 1:26 AM, Erik Kaashoek <erik@kaashoek.com> wrote:
>
> Bob, Using the approach you described I was able to verify the noise floor of my initial measurement setup to be at -80dBc at 10Hz offset from carrier and -100dBc above 1kHz with a strong peaks at 50Hz, 60Hz and harmonics. This all with an RBW in the FFT of around 4Hz.
> This level of noise is way to high to do phase noise measurement so I'm now going to work on removing ground loops and adding low noise amplification.
> Erik.
>
> On 21-6-2022 2:04, Bob kb8tq wrote:
>> Hi
>>
>> Ok, single mixer phase noise basics:
>>
>> First thing is to womp the mixer up to the point it almost smokes. Putting +7 dbm into
>> both ports on a “7 dbm” mixer is very normal in this case. Watching for the fact that the
>> mixer likely is *not* a 50 ohm load is part of the process ( = pads might help out) as well
>> as understanding that it does not have a monster amount of isolation ( = isolation amps
>> may be needed ).
>>
>> Next one generates a beat note by offsetting the two oscillators a bit. This gives you a nice
>> 360 degree sweep function ( 360 degrees per cycle :) ). From that you can work out the
>> system sensitivity in volts per degree ( or better yet per radian since that’s what you actually
>> want as the “magic number” …. it’s phase modulation so radian is king …).
>>
>> Next you lock the two oscillators together via a DC feed out of the mixer to one or the
>> other of them. You adjust the “lock point” so that it is at zero volts out of the mixer. This
>> puts the two oscillators in quadrature. Yes, there is that messy 2X the input frequency RF
>> output and the inevitable leakage. Those are handled with a lowpass filter.
>>
>> The output of the mixer is now “just noise”. There is no nasty carrier to deal with. There is
>> no messy fold over to wonder about. What you get is the DSB noise ( so both sides of
>> carrier) from the sum of the two oscillators.
>>
>> Output of the mixer goes up if you terminate it in an “high” load. Something like 500 ohms
>> on a 50 ohm mixer or 5K ohms on an RPD-1 is often used. The isolation seems to be ok
>> either way and the added gain / better floor is “free”.
>>
>> Simply put you add 3 db when you look at DSB if it’s uncorrelated, and another 3 db if the
>> oscillators are identical. Your “output” is 6 db higher than the single sideband / single oscillator
>> phase noise. You can argue that close in noise is likely correlated due to it being a modulation
>> on the carrier. The standard convention is to use 3 db.
>>
>> Amplify the noise up and you can measure very low levels of phase noise. Low noise
>> audio op-amps are pretty easy to find spec sheets on. With anything these days finding
>> them on the shelf may be “interesting”. The OP-27 / OP-37 with low resistance in the
>> feedback path go way back for this application. There are a lot of other candidates.
>>
>> The cutoff of the lock signal typically is adjustable to keep it below the lowest point
>> of interest for your noise testing. If that is impractical, there are ways to calibrate and
>> read “inside the loop”.
>>
>> The HP 3048 phase noise analyzer was based on this approach. The original app note most
>> folks started from came from Fluke back in the early 1970’s. I have not (yet) found a good
>> copy of it on the internet.
>>
>> Fun !!!
>>
>> Bob
>>
>>
>>
>>> On Jun 20, 2022, at 9:43 AM, Erik Kaashoek via time-nuts <time-nuts@lists.febo.com> wrote:
>>>
>>> Bob, Magnus,
>>> Thanks, clear. A counter is for ADEV, not for phase noise.
>>> I made a test setup to learn how to use the mixer/PLL approach.
>>> First using 10MHz from both outputs of a DSS (Rigol DG990) to observe the DC shift with shifting the phase between the two signal.
>>> Then by modulating one output with FM or PM.
>>> There is a low pass filter after the mixer to get rid of the 10 MHz and its harmonics but the LPF is measured flat till about 10kHz.
>>> The output signal from the mixer was kept within 10% of the full voltage swing to stay in the (hopefully) linear range.
>>> Using PM creates a low frequency output from the mixer that is proportional to the phase shift (region 0-1 degree) and constant in amplitude with change of frequency. Also when using external modulation from an audio signal generator created the expected behavior with drive level and no frequency impact
>>> Using FM with 0.1 Hz frequency deviation the mixer output amplitude decreases very fast with increasing frequency (range 0.1 to 10 Hz)
>>> Also when using 1 Hz or more frequency deviation. The higher frequency deviation leads to higher output levels as expected.
>>> Can someone help me understand how this FM signal (0.1 to 1000 Hz modulation and 0.1 to 1 Hz frequency deviation) translates to the calibration example mentioned in the document on phase noise measurement as linked by Bob. (0.1 Hz deviation at 1 kHz rate leading to a sideband (at 1kHz?) level of -86 dBc)
>>> At a 1kHz rate I see (yet) no output from the mixer where at 1Hz there is a lot of output. Why is this output frequency dependency?
>>> Is this a problem with the signal generator? Or the mixer?
>>> Then I tried to use the modulated signal from the SG PLL locked to a 10MHz VCO. Results where the same. FM output signal is frequency dependent, PM not.
>>> Erik.
>>>
>>> _______________________________________________
>>> time-nuts mailing list -- time-nuts@lists.febo.com
>>> To unsubscribe send an email to time-nuts-leave@lists.febo.com
>
EK
Erik Kaashoek
Sat, Jun 25, 2022 6:07 PM
Thanks to all the great help from people on this list I was able to make
some progress in doing close-in phase noise measurements.
The setup consists of a VC-OCXO going into the LO port of an ADE-1
mixer, The DUT into the RF port, the IF port is low pass filtered and
used to steer the VC-OCXO and is send to a high quality 24 bit USB audio
capture unit connected to a PC running ARTA.
The ADE-1 mixer was selected because that all ports are completely
isolated from each other, there is no common ground, which helps to
reduce ground loop problems a bit.
The log plots from ARTA confirm a 130dB dynamic range and the resolution
bandwidth is supposed to be about 1Hz. Each plot was averaged over 10
measurements
All input signals where normalized by attenuators to have their carrier
at 0dB in the plot.
Using a generator at variable frequency offset it was confirmed the
audio input is flat down to 1Hz.
Using a generator with phase modulation down to 0.001 degree the
sensitivity of the measurement chain was checked. (20dB level reduction
with every factor 10 reduction in phase modulation depth)
It is expected to have at least +/-5dB level inaccuracy.
The DUTS measured where:
- a fairly clean XO (PN_XO.JPG)
- a rather bad GPSDO output (PN_GPSDO.JPG)
- The not so famous cheap Chinese TCXO (PN_TCXO.KPG)
- The output of a Rigol SG (PN_Rigol.JPG)
The XO is the cleanest
The TCXO shows odd spurs between 10 and 40 Hz and the PN does not drop
down as it should (spec states: -135dBc/Hz at 1kHz offset)
The GPSDO is terrible, this demonstrates you can have a 1e-10 ADEV at 1s
tau from a bad oscillator.
The Rigol is not so clean and a PLL shoulder seems to be present just
above 1kHz.
Next step is to add low noise gain close to the mixer LPF output to get
more dynamic range and a better VC-OCXO (Morion MV170 (PN -100dBc/Hz at
1Hz offset) to lower the impact of the reference VC-OCXO
Erik.
Thanks to all the great help from people on this list I was able to make
some progress in doing close-in phase noise measurements.
The setup consists of a VC-OCXO going into the LO port of an ADE-1
mixer, The DUT into the RF port, the IF port is low pass filtered and
used to steer the VC-OCXO and is send to a high quality 24 bit USB audio
capture unit connected to a PC running ARTA.
The ADE-1 mixer was selected because that all ports are completely
isolated from each other, there is no common ground, which helps to
reduce ground loop problems a bit.
The log plots from ARTA confirm a 130dB dynamic range and the resolution
bandwidth is supposed to be about 1Hz. Each plot was averaged over 10
measurements
All input signals where normalized by attenuators to have their carrier
at 0dB in the plot.
Using a generator at variable frequency offset it was confirmed the
audio input is flat down to 1Hz.
Using a generator with phase modulation down to 0.001 degree the
sensitivity of the measurement chain was checked. (20dB level reduction
with every factor 10 reduction in phase modulation depth)
It is expected to have at least +/-5dB level inaccuracy.
The DUTS measured where:
- a fairly clean XO (PN_XO.JPG)
- a rather bad GPSDO output (PN_GPSDO.JPG)
- The not so famous cheap Chinese TCXO (PN_TCXO.KPG)
- The output of a Rigol SG (PN_Rigol.JPG)
The XO is the cleanest
The TCXO shows odd spurs between 10 and 40 Hz and the PN does not drop
down as it should (spec states: -135dBc/Hz at 1kHz offset)
The GPSDO is terrible, this demonstrates you can have a 1e-10 ADEV at 1s
tau from a bad oscillator.
The Rigol is not so clean and a PLL shoulder seems to be present just
above 1kHz.
Next step is to add low noise gain close to the mixer LPF output to get
more dynamic range and a better VC-OCXO (Morion MV170 (PN -100dBc/Hz at
1Hz offset) to lower the impact of the reference VC-OCXO
Erik.
BK
Bob kb8tq
Sun, Jun 26, 2022 5:01 PM
Hi
Ok, so mixer phase noise calibration:
Set things up with “full blast” inputs to both sides of the mixer. Keep them
at the same level through all tests. This might be +7 on both ports, it might
be +13, it could be +20 dbm. ( yes, mixers do get fried this way ….).
First cal step, you have the radians to volts factor. The “zero db” point for phase
noise is one radian. It is unlikely that you come up with exactly one volt per
radian. Oddly enough …. it can happen. This is done by looking at a free running
beat note and measuring the slope at a zero crossing. ( 0.1 volts over 10 degrees
maybe …). If you have a pre-amp involved, it needs to be fiddled to keep it out
of saturation.
Back in the day of analog devices, you had a bunch of fun with bandwidths
and averaging factors. These days you run a program and it gives you a 1 hertz
normalized number. You are missing a lot of fun there …. :)
As previously mentioned, your result is at least 3 db higher than the single
sideband phase noise. ( Yes, that’s the definition … single sideband). Back
before the modern era folks used 6 db close in. Not so much anymore.
Finally, you get whatever the combination factor is for your two sources. If
they are identical, you get another 3 db.
Spurs are a bit of an issue. Since you have done this or that for a 1 Hz
normalization, that can mess up the spur levels. They are an absolute number
rather than bandwidth normalized. For home / basement use, you typically ignore
this. On a “pro” device, they have fancy software that guesses what is a spur and
de-normalizes that section.
Some folks do the whole slope measurement thing every time. Others do a
(unsaturated preamp) beat note and then use a “known” correction factor. Since
the beat note is not a sine wave, it can not be used directly as a reference. The “right”
answer is to do the slope every time. If you are measuring a hundred same / same
devices that day, you go with the beat note.
Bob
On Jun 25, 2022, at 10:07 AM, Erik Kaashoek via time-nuts time-nuts@lists.febo.com wrote:
Thanks to all the great help from people on this list I was able to make some progress in doing close-in phase noise measurements.
The setup consists of a VC-OCXO going into the LO port of an ADE-1 mixer, The DUT into the RF port, the IF port is low pass filtered and used to steer the VC-OCXO and is send to a high quality 24 bit USB audio capture unit connected to a PC running ARTA.
The ADE-1 mixer was selected because that all ports are completely isolated from each other, there is no common ground, which helps to reduce ground loop problems a bit.
The log plots from ARTA confirm a 130dB dynamic range and the resolution bandwidth is supposed to be about 1Hz. Each plot was averaged over 10 measurements
All input signals where normalized by attenuators to have their carrier at 0dB in the plot.
Using a generator at variable frequency offset it was confirmed the audio input is flat down to 1Hz.
Using a generator with phase modulation down to 0.001 degree the sensitivity of the measurement chain was checked. (20dB level reduction with every factor 10 reduction in phase modulation depth)
It is expected to have at least +/-5dB level inaccuracy.
The DUTS measured where:
- a fairly clean XO (PN_XO.JPG)
- a rather bad GPSDO output (PN_GPSDO.JPG)
- The not so famous cheap Chinese TCXO (PN_TCXO.KPG)
- The output of a Rigol SG (PN_Rigol.JPG)
The XO is the cleanest
The TCXO shows odd spurs between 10 and 40 Hz and the PN does not drop down as it should (spec states: -135dBc/Hz at 1kHz offset)
The GPSDO is terrible, this demonstrates you can have a 1e-10 ADEV at 1s tau from a bad oscillator.
The Rigol is not so clean and a PLL shoulder seems to be present just above 1kHz.
Next step is to add low noise gain close to the mixer LPF output to get more dynamic range and a better VC-OCXO (Morion MV170 (PN -100dBc/Hz at 1Hz offset) to lower the impact of the reference VC-OCXO
Erik.
<PN_XO.JPG><PN_GPSDO.JPG><PN_TCXO.JPG><PN_Rigol.JPG>_______________________________________________
time-nuts mailing list -- time-nuts@lists.febo.com
To unsubscribe send an email to time-nuts-leave@lists.febo.com
Hi
Ok, so mixer phase noise calibration:
Set things up with “full blast” inputs to both sides of the mixer. Keep them
at the same level through all tests. This might be +7 on both ports, it might
be +13, it could be +20 dbm. ( yes, mixers *do* get fried this way ….).
First cal step, you have the radians to volts factor. The “zero db” point for phase
noise is one radian. It is unlikely that you come up with exactly one volt per
radian. Oddly enough …. it can happen. This is done by looking at a free running
beat note and measuring the slope at a zero crossing. ( 0.1 volts over 10 degrees
maybe …). If you have a pre-amp involved, it needs to be fiddled to keep it out
of saturation.
Back in the day of analog devices, you had a bunch of fun with bandwidths
and averaging factors. These days you run a program and it gives you a 1 hertz
normalized number. You are missing a *lot* of fun there …. :)
As previously mentioned, your result is at least 3 db higher than the single
sideband phase noise. ( Yes, that’s the definition … single sideband). Back
before the modern era folks used 6 db close in. Not so much anymore.
Finally, you get whatever the combination factor is for your two sources. If
they are identical, you get another 3 db.
Spurs are a bit of an issue. Since you have done this or that for a 1 Hz
normalization, that can mess up the spur levels. They are an absolute number
rather than bandwidth normalized. For home / basement use, you typically ignore
this. On a “pro” device, they have fancy software that guesses what is a spur and
de-normalizes that section.
Some folks do the whole slope measurement thing every time. Others do a
(unsaturated preamp) beat note and then use a “known” correction factor. Since
the beat note is not a sine wave, it can not be used directly as a reference. The “right”
answer is to do the slope every time. If you are measuring a hundred same / same
devices that day, you go with the beat note.
Bob
> On Jun 25, 2022, at 10:07 AM, Erik Kaashoek via time-nuts <time-nuts@lists.febo.com> wrote:
>
> Thanks to all the great help from people on this list I was able to make some progress in doing close-in phase noise measurements.
> The setup consists of a VC-OCXO going into the LO port of an ADE-1 mixer, The DUT into the RF port, the IF port is low pass filtered and used to steer the VC-OCXO and is send to a high quality 24 bit USB audio capture unit connected to a PC running ARTA.
> The ADE-1 mixer was selected because that all ports are completely isolated from each other, there is no common ground, which helps to reduce ground loop problems a bit.
> The log plots from ARTA confirm a 130dB dynamic range and the resolution bandwidth is supposed to be about 1Hz. Each plot was averaged over 10 measurements
> All input signals where normalized by attenuators to have their carrier at 0dB in the plot.
> Using a generator at variable frequency offset it was confirmed the audio input is flat down to 1Hz.
> Using a generator with phase modulation down to 0.001 degree the sensitivity of the measurement chain was checked. (20dB level reduction with every factor 10 reduction in phase modulation depth)
> It is expected to have at least +/-5dB level inaccuracy.
> The DUTS measured where:
> - a fairly clean XO (PN_XO.JPG)
> - a rather bad GPSDO output (PN_GPSDO.JPG)
> - The not so famous cheap Chinese TCXO (PN_TCXO.KPG)
> - The output of a Rigol SG (PN_Rigol.JPG)
> The XO is the cleanest
> The TCXO shows odd spurs between 10 and 40 Hz and the PN does not drop down as it should (spec states: -135dBc/Hz at 1kHz offset)
> The GPSDO is terrible, this demonstrates you can have a 1e-10 ADEV at 1s tau from a bad oscillator.
> The Rigol is not so clean and a PLL shoulder seems to be present just above 1kHz.
>
> Next step is to add low noise gain close to the mixer LPF output to get more dynamic range and a better VC-OCXO (Morion MV170 (PN -100dBc/Hz at 1Hz offset) to lower the impact of the reference VC-OCXO
> Erik.
> <PN_XO.JPG><PN_GPSDO.JPG><PN_TCXO.JPG><PN_Rigol.JPG>_______________________________________________
> time-nuts mailing list -- time-nuts@lists.febo.com
> To unsubscribe send an email to time-nuts-leave@lists.febo.com
MD
Magnus Danielson
Sun, Jun 26, 2022 6:52 PM
Hi Erik!
Great progress! Sure interesting to look at them phase-noise plots,
right? It's a really good tool in addition to the stability of ADEV and
friends.
As I recall it, the ADE-1 is not documented to be isolated, but it is
very obvious when you look down the backside of it. However, it has
capacitive coupling and one should consider both common mode rejection
and common mode loading it down for these to work well.
Word of caution when it comes to levels, as the windowing filter used
causes shifts in noise-levels, so estimation of noise-levels becomes a
little bit tricky as you try to get the nitty gritty right, but getting
the overall shape view you already gained a lot with the things you
achieved.
A technique used to push further down into lower noise-levels is the
cross-correlation technique, where you split the signal into two
channels, each being exactly what you have now, and then rather than
squaring the output of the FFT from each channel, you multiply one with
the completment of the other, then average on those. This allows you to
supress the noise of each reference oscillator. You do not have to go
there from start, as you already make very useful measurements, but I'm
just suggesting what may lie up ahead.
Compared to some of the other sources, the Rigol SG does fairly well,
but then again, things can be even more quiet. For the XO you can see
the 15 dB/Oct slope as expected for flicker frequency. Try to locate the
source of the peaks you see and see if you can clean it up. The XO seems
to be a fairly good DUT for doing that.
Cheers,
Magnus
On 2022-06-25 20:07, Erik Kaashoek via time-nuts wrote:
Thanks to all the great help from people on this list I was able to
make some progress in doing close-in phase noise measurements.
The setup consists of a VC-OCXO going into the LO port of an ADE-1
mixer, The DUT into the RF port, the IF port is low pass filtered and
used to steer the VC-OCXO and is send to a high quality 24 bit USB
audio capture unit connected to a PC running ARTA.
The ADE-1 mixer was selected because that all ports are completely
isolated from each other, there is no common ground, which helps to
reduce ground loop problems a bit.
The log plots from ARTA confirm a 130dB dynamic range and the
resolution bandwidth is supposed to be about 1Hz. Each plot was
averaged over 10 measurements
All input signals where normalized by attenuators to have their
carrier at 0dB in the plot.
Using a generator at variable frequency offset it was confirmed the
audio input is flat down to 1Hz.
Using a generator with phase modulation down to 0.001 degree the
sensitivity of the measurement chain was checked. (20dB level
reduction with every factor 10 reduction in phase modulation depth)
It is expected to have at least +/-5dB level inaccuracy.
The DUTS measured where:
- a fairly clean XO (PN_XO.JPG)
- a rather bad GPSDO output (PN_GPSDO.JPG)
- The not so famous cheap Chinese TCXO (PN_TCXO.KPG)
- The output of a Rigol SG (PN_Rigol.JPG)
The XO is the cleanest
The TCXO shows odd spurs between 10 and 40 Hz and the PN does not drop
down as it should (spec states: -135dBc/Hz at 1kHz offset)
The GPSDO is terrible, this demonstrates you can have a 1e-10 ADEV at
1s tau from a bad oscillator.
The Rigol is not so clean and a PLL shoulder seems to be present just
above 1kHz.
Next step is to add low noise gain close to the mixer LPF output to
get more dynamic range and a better VC-OCXO (Morion MV170 (PN
-100dBc/Hz at 1Hz offset) to lower the impact of the reference VC-OCXO
Erik.
time-nuts mailing list -- time-nuts@lists.febo.com
To unsubscribe send an email to time-nuts-leave@lists.febo.com
Hi Erik!
Great progress! Sure interesting to look at them phase-noise plots,
right? It's a really good tool in addition to the stability of ADEV and
friends.
As I recall it, the ADE-1 is not documented to be isolated, but it is
very obvious when you look down the backside of it. However, it has
capacitive coupling and one should consider both common mode rejection
and common mode loading it down for these to work well.
Word of caution when it comes to levels, as the windowing filter used
causes shifts in noise-levels, so estimation of noise-levels becomes a
little bit tricky as you try to get the nitty gritty right, but getting
the overall shape view you already gained a lot with the things you
achieved.
A technique used to push further down into lower noise-levels is the
cross-correlation technique, where you split the signal into two
channels, each being exactly what you have now, and then rather than
squaring the output of the FFT from each channel, you multiply one with
the completment of the other, then average on those. This allows you to
supress the noise of each reference oscillator. You do not have to go
there from start, as you already make very useful measurements, but I'm
just suggesting what may lie up ahead.
Compared to some of the other sources, the Rigol SG does fairly well,
but then again, things can be even more quiet. For the XO you can see
the 15 dB/Oct slope as expected for flicker frequency. Try to locate the
source of the peaks you see and see if you can clean it up. The XO seems
to be a fairly good DUT for doing that.
Cheers,
Magnus
On 2022-06-25 20:07, Erik Kaashoek via time-nuts wrote:
> Thanks to all the great help from people on this list I was able to
> make some progress in doing close-in phase noise measurements.
> The setup consists of a VC-OCXO going into the LO port of an ADE-1
> mixer, The DUT into the RF port, the IF port is low pass filtered and
> used to steer the VC-OCXO and is send to a high quality 24 bit USB
> audio capture unit connected to a PC running ARTA.
> The ADE-1 mixer was selected because that all ports are completely
> isolated from each other, there is no common ground, which helps to
> reduce ground loop problems a bit.
> The log plots from ARTA confirm a 130dB dynamic range and the
> resolution bandwidth is supposed to be about 1Hz. Each plot was
> averaged over 10 measurements
> All input signals where normalized by attenuators to have their
> carrier at 0dB in the plot.
> Using a generator at variable frequency offset it was confirmed the
> audio input is flat down to 1Hz.
> Using a generator with phase modulation down to 0.001 degree the
> sensitivity of the measurement chain was checked. (20dB level
> reduction with every factor 10 reduction in phase modulation depth)
> It is expected to have at least +/-5dB level inaccuracy.
> The DUTS measured where:
> - a fairly clean XO (PN_XO.JPG)
> - a rather bad GPSDO output (PN_GPSDO.JPG)
> - The not so famous cheap Chinese TCXO (PN_TCXO.KPG)
> - The output of a Rigol SG (PN_Rigol.JPG)
> The XO is the cleanest
> The TCXO shows odd spurs between 10 and 40 Hz and the PN does not drop
> down as it should (spec states: -135dBc/Hz at 1kHz offset)
> The GPSDO is terrible, this demonstrates you can have a 1e-10 ADEV at
> 1s tau from a bad oscillator.
> The Rigol is not so clean and a PLL shoulder seems to be present just
> above 1kHz.
>
> Next step is to add low noise gain close to the mixer LPF output to
> get more dynamic range and a better VC-OCXO (Morion MV170 (PN
> -100dBc/Hz at 1Hz offset) to lower the impact of the reference VC-OCXO
> Erik.
>
> _______________________________________________
> time-nuts mailing list -- time-nuts@lists.febo.com
> To unsubscribe send an email to time-nuts-leave@lists.febo.com
EK
Erik Kaashoek
Mon, Jun 27, 2022 9:43 AM
Magnus, Bob,
When the mixer is operating in the linear region for the DUT input (0dBm
or lower), would it be possible to use a calibrated noise sources to do
an extra verification of the noise level measurement?
Of course with a noise source you get 3dB as both sidebands fold.
Verification steps:
Verify the DUT output level is correctly brought to 0dB (using
attenuators) using a calibrated spectrum analyzer
Connect the DUT to the phase measurement setup and set the reference to
a 500Hz offset to get a beat note and verify the beat note is registered
at 0dB, change the DUT level some dB up and down to confirm its in a
linear region.
Measure the per Hz output power of a noise source using a calibrated
spectrum analyzer and a noise marker set to 10MHz.
Connect the noise source to the phase measurement setup and check if the
noise level is measured at level measured by the spectrum analyzer + 3dB
This should work if the RBW of the phase measurement is indeed set to 1Hz.
Another verification option may be to use the phase modulation of a
signal generator. This can not check the effective noise bandwidth of
the FFT but it can check linearity over the whole range.
The output of the mixer is terminated with 50ohm so a factor of 10 in
voltage should give a 20dB power step.
When operation in the linear range the phase noise measurement setup
should measure 20dB less with every factor 10 reduction in phase
modulation depth where 90 degrees is equal to 100% modulation depth so
equal to the signal you get when measuring a beat note.
When measuring with modulation depth of 90,9,0.9,0.09 and 0.009 degrees
the measured level should step from 0,-20,-40, -60 to -80dB
Any feedback?
Erik.
On 26-6-2022 20:52, Magnus Danielson via time-nuts wrote:
Hi Erik!
Great progress! Sure interesting to look at them phase-noise plots,
right? It's a really good tool in addition to the stability of ADEV
and friends.
As I recall it, the ADE-1 is not documented to be isolated, but it is
very obvious when you look down the backside of it. However, it has
capacitive coupling and one should consider both common mode rejection
and common mode loading it down for these to work well.
Word of caution when it comes to levels, as the windowing filter used
causes shifts in noise-levels, so estimation of noise-levels becomes a
little bit tricky as you try to get the nitty gritty right, but
getting the overall shape view you already gained a lot with the
things you achieved.
A technique used to push further down into lower noise-levels is the
cross-correlation technique, where you split the signal into two
channels, each being exactly what you have now, and then rather than
squaring the output of the FFT from each channel, you multiply one
with the completment of the other, then average on those. This allows
you to supress the noise of each reference oscillator. You do not have
to go there from start, as you already make very useful measurements,
but I'm just suggesting what may lie up ahead.
Compared to some of the other sources, the Rigol SG does fairly well,
but then again, things can be even more quiet. For the XO you can see
the 15 dB/Oct slope as expected for flicker frequency. Try to locate
the source of the peaks you see and see if you can clean it up. The XO
seems to be a fairly good DUT for doing that.
Cheers,
Magnus
Magnus, Bob,
When the mixer is operating in the linear region for the DUT input (0dBm
or lower), would it be possible to use a calibrated noise sources to do
an extra verification of the noise level measurement?
Of course with a noise source you get 3dB as both sidebands fold.
Verification steps:
Verify the DUT output level is correctly brought to 0dB (using
attenuators) using a calibrated spectrum analyzer
Connect the DUT to the phase measurement setup and set the reference to
a 500Hz offset to get a beat note and verify the beat note is registered
at 0dB, change the DUT level some dB up and down to confirm its in a
linear region.
Measure the per Hz output power of a noise source using a calibrated
spectrum analyzer and a noise marker set to 10MHz.
Connect the noise source to the phase measurement setup and check if the
noise level is measured at level measured by the spectrum analyzer + 3dB
This should work if the RBW of the phase measurement is indeed set to 1Hz.
Another verification option may be to use the phase modulation of a
signal generator. This can not check the effective noise bandwidth of
the FFT but it can check linearity over the whole range.
The output of the mixer is terminated with 50ohm so a factor of 10 in
voltage should give a 20dB power step.
When operation in the linear range the phase noise measurement setup
should measure 20dB less with every factor 10 reduction in phase
modulation depth where 90 degrees is equal to 100% modulation depth so
equal to the signal you get when measuring a beat note.
When measuring with modulation depth of 90,9,0.9,0.09 and 0.009 degrees
the measured level should step from 0,-20,-40, -60 to -80dB
Any feedback?
Erik.
On 26-6-2022 20:52, Magnus Danielson via time-nuts wrote:
> Hi Erik!
>
> Great progress! Sure interesting to look at them phase-noise plots,
> right? It's a really good tool in addition to the stability of ADEV
> and friends.
>
> As I recall it, the ADE-1 is not documented to be isolated, but it is
> very obvious when you look down the backside of it. However, it has
> capacitive coupling and one should consider both common mode rejection
> and common mode loading it down for these to work well.
>
> Word of caution when it comes to levels, as the windowing filter used
> causes shifts in noise-levels, so estimation of noise-levels becomes a
> little bit tricky as you try to get the nitty gritty right, but
> getting the overall shape view you already gained a lot with the
> things you achieved.
>
> A technique used to push further down into lower noise-levels is the
> cross-correlation technique, where you split the signal into two
> channels, each being exactly what you have now, and then rather than
> squaring the output of the FFT from each channel, you multiply one
> with the completment of the other, then average on those. This allows
> you to supress the noise of each reference oscillator. You do not have
> to go there from start, as you already make very useful measurements,
> but I'm just suggesting what may lie up ahead.
>
> Compared to some of the other sources, the Rigol SG does fairly well,
> but then again, things can be even more quiet. For the XO you can see
> the 15 dB/Oct slope as expected for flicker frequency. Try to locate
> the source of the peaks you see and see if you can clean it up. The XO
> seems to be a fairly good DUT for doing that.
>
> Cheers,
> Magnus
BK
Bob kb8tq
Mon, Jun 27, 2022 4:06 PM
On Jun 27, 2022, at 1:43 AM, Erik Kaashoek via time-nuts time-nuts@lists.febo.com wrote:
Magnus, Bob,
When the mixer is operating in the linear region for the DUT input (0dBm or lower), would it be possible to use a calibrated noise sources to do an extra verification of the noise level measurement?
Of course with a noise source you get 3dB as both sidebands fold.
The “normal” approach is to put the mixer into saturation. This gives you the
best noise floor. It also does a bit better at separating AM noise from PM noise
(since you are trying to measure phase noise …..).
Verification steps:
Verify the DUT output level is correctly brought to 0dB (using attenuators) using a calibrated spectrum analyzer
Connect the DUT to the phase measurement setup and set the reference to a 500Hz offset to get a beat note and verify the beat note is registered at 0dB, change the DUT level some dB up and down to confirm its in a linear region.
Measure the per Hz output power of a noise source using a calibrated spectrum analyzer and a noise marker set to 10MHz.
Connect the noise source to the phase measurement setup and check if the noise level is measured at level measured by the spectrum analyzer + 3dB
This should work if the RBW of the phase measurement is indeed set to 1Hz.
If you do it this way, you still need to do conversion for the “one radian” reference
level that is used with phase modulation ( = the reference is not one cycle ). Yes
that’s a bit weird / obscure.
Another verification option may be to use the phase modulation of a signal generator. This can not check the effective noise bandwidth of the FFT but it can check linearity over the whole range.
The output of the mixer is terminated with 50ohm so a factor of 10 in voltage should give a 20dB power step.
Audio termination at 50 ohms does not do much for isolation. (again a bit of an obscure
point). By terminating in 10X the nominal impedance ( so 500 ohms in this case) you get
another 6 db of gain in the system. Since this is ahead of the preamp, it might improve
your noise floor.
Bob
When operation in the linear range the phase noise measurement setup should measure 20dB less with every factor 10 reduction in phase modulation depth where 90 degrees is equal to 100% modulation depth so equal to the signal you get when measuring a beat note.
When measuring with modulation depth of 90,9,0.9,0.09 and 0.009 degrees the measured level should step from 0,-20,-40, -60 to -80dB
Any feedback?
Erik.
On 26-6-2022 20:52, Magnus Danielson via time-nuts wrote:
Hi Erik!
Great progress! Sure interesting to look at them phase-noise plots, right? It's a really good tool in addition to the stability of ADEV and friends.
As I recall it, the ADE-1 is not documented to be isolated, but it is very obvious when you look down the backside of it. However, it has capacitive coupling and one should consider both common mode rejection and common mode loading it down for these to work well.
Word of caution when it comes to levels, as the windowing filter used causes shifts in noise-levels, so estimation of noise-levels becomes a little bit tricky as you try to get the nitty gritty right, but getting the overall shape view you already gained a lot with the things you achieved.
A technique used to push further down into lower noise-levels is the cross-correlation technique, where you split the signal into two channels, each being exactly what you have now, and then rather than squaring the output of the FFT from each channel, you multiply one with the completment of the other, then average on those. This allows you to supress the noise of each reference oscillator. You do not have to go there from start, as you already make very useful measurements, but I'm just suggesting what may lie up ahead.
Compared to some of the other sources, the Rigol SG does fairly well, but then again, things can be even more quiet. For the XO you can see the 15 dB/Oct slope as expected for flicker frequency. Try to locate the source of the peaks you see and see if you can clean it up. The XO seems to be a fairly good DUT for doing that.
Cheers,
Magnus
Hi
> On Jun 27, 2022, at 1:43 AM, Erik Kaashoek via time-nuts <time-nuts@lists.febo.com> wrote:
>
> Magnus, Bob,
> When the mixer is operating in the linear region for the DUT input (0dBm or lower), would it be possible to use a calibrated noise sources to do an extra verification of the noise level measurement?
> Of course with a noise source you get 3dB as both sidebands fold.
The “normal” approach is to put the mixer into saturation. This gives you the
best noise floor. It also does a bit better at separating AM noise from PM noise
(since you are trying to measure phase noise …..).
> Verification steps:
> Verify the DUT output level is correctly brought to 0dB (using attenuators) using a calibrated spectrum analyzer
> Connect the DUT to the phase measurement setup and set the reference to a 500Hz offset to get a beat note and verify the beat note is registered at 0dB, change the DUT level some dB up and down to confirm its in a linear region.
> Measure the per Hz output power of a noise source using a calibrated spectrum analyzer and a noise marker set to 10MHz.
> Connect the noise source to the phase measurement setup and check if the noise level is measured at level measured by the spectrum analyzer + 3dB
> This should work if the RBW of the phase measurement is indeed set to 1Hz.
If you do it this way, you still need to do conversion for the “one radian” reference
level that is used with phase modulation ( = the reference is *not* one cycle ). Yes
that’s a bit weird / obscure.
>
> Another verification option may be to use the phase modulation of a signal generator. This can not check the effective noise bandwidth of the FFT but it can check linearity over the whole range.
> The output of the mixer is terminated with 50ohm so a factor of 10 in voltage should give a 20dB power step.
Audio termination at 50 ohms does not do much for isolation. (again a bit of an obscure
point). By terminating in 10X the nominal impedance ( so 500 ohms in this case) you get
another 6 db of gain in the system. Since this is ahead of the preamp, it might improve
your noise floor.
Bob
> When operation in the linear range the phase noise measurement setup should measure 20dB less with every factor 10 reduction in phase modulation depth where 90 degrees is equal to 100% modulation depth so equal to the signal you get when measuring a beat note.
> When measuring with modulation depth of 90,9,0.9,0.09 and 0.009 degrees the measured level should step from 0,-20,-40, -60 to -80dB
>
> Any feedback?
> Erik.
>
> On 26-6-2022 20:52, Magnus Danielson via time-nuts wrote:
>> Hi Erik!
>>
>> Great progress! Sure interesting to look at them phase-noise plots, right? It's a really good tool in addition to the stability of ADEV and friends.
>>
>> As I recall it, the ADE-1 is not documented to be isolated, but it is very obvious when you look down the backside of it. However, it has capacitive coupling and one should consider both common mode rejection and common mode loading it down for these to work well.
>>
>> Word of caution when it comes to levels, as the windowing filter used causes shifts in noise-levels, so estimation of noise-levels becomes a little bit tricky as you try to get the nitty gritty right, but getting the overall shape view you already gained a lot with the things you achieved.
>>
>> A technique used to push further down into lower noise-levels is the cross-correlation technique, where you split the signal into two channels, each being exactly what you have now, and then rather than squaring the output of the FFT from each channel, you multiply one with the completment of the other, then average on those. This allows you to supress the noise of each reference oscillator. You do not have to go there from start, as you already make very useful measurements, but I'm just suggesting what may lie up ahead.
>>
>> Compared to some of the other sources, the Rigol SG does fairly well, but then again, things can be even more quiet. For the XO you can see the 15 dB/Oct slope as expected for flicker frequency. Try to locate the source of the peaks you see and see if you can clean it up. The XO seems to be a fairly good DUT for doing that.
>>
>> Cheers,
>> Magnus
> _______________________________________________
> time-nuts mailing list -- time-nuts@lists.febo.com
> To unsubscribe send an email to time-nuts-leave@lists.febo.com
MD
Magnus Danielson
Mon, Jun 27, 2022 4:26 PM
Hi Erik,
On 2022-06-27 11:43, Erik Kaashoek via time-nuts wrote:
Magnus, Bob,
When the mixer is operating in the linear region for the DUT input
(0dBm or lower), would it be possible to use a calibrated noise
sources to do an extra verification of the noise level measurement?
Yes. NIST build such calibrators for exactly that purpose.
Of course with a noise source you get 3dB as both sidebands fold.
Verification steps:
Verify the DUT output level is correctly brought to 0dB (using
attenuators) using a calibrated spectrum analyzer
Connect the DUT to the phase measurement setup and set the reference
to a 500Hz offset to get a beat note and verify the beat note is
registered at 0dB, change the DUT level some dB up and down to confirm
its in a linear region.
Measure the per Hz output power of a noise source using a calibrated
spectrum analyzer and a noise marker set to 10MHz.
Connect the noise source to the phase measurement setup and check if
the noise level is measured at level measured by the spectrum analyzer
- 3dB
This should work if the RBW of the phase measurement is indeed set to
1Hz.
500 Hz is kind of arbitrary number you chose there. At the very least it
should be verified, but I would assume there is an underlying goal which
made you choose 500 Hz and that should be specified.
The RBW is not the noise BW. You need to correct the bin-bandwidth with
the specific noise bandwidth correction for the window-filter you use.
There is a neat article [1] on it which I also contributed into IEEE 1139.
Another verification option may be to use the phase modulation of a
signal generator. This can not check the effective noise bandwidth of
the FFT but it can check linearity over the whole range.
The output of the mixer is terminated with 50ohm so a factor of 10 in
voltage should give a 20dB power step.
When operation in the linear range the phase noise measurement setup
should measure 20dB less with every factor 10 reduction in phase
modulation depth where 90 degrees is equal to 100% modulation depth so
equal to the signal you get when measuring a beat note.
When measuring with modulation depth of 90,9,0.9,0.09 and 0.009
degrees the measured level should step from 0,-20,-40, -60 to -80dB
For higher modulation depth it's very hard to do linear modulation well.
There is a different variant you can use, to inject a signal to form a
side-band signal next to the carrier. By creating a specific amplitude
compared to the carrier, and for a particular offset, it's trivial
exercise to know the AM and PM noise level. A single sideband sine will
divide equally to AM and PM, thus reducing 3 dB. Thus, setting a
side-band at 27 dB bellow the carrier, makes a -30 dBc phase modulation.
[1]
https://pure.mpg.de/pubman/faces/ViewItemOverviewPage.jsp?itemId=item_152164
Cheers,
Magnus
Any feedback?
Erik.
On 26-6-2022 20:52, Magnus Danielson via time-nuts wrote:
Hi Erik!
Great progress! Sure interesting to look at them phase-noise plots,
right? It's a really good tool in addition to the stability of ADEV
and friends.
As I recall it, the ADE-1 is not documented to be isolated, but it is
very obvious when you look down the backside of it. However, it has
capacitive coupling and one should consider both common mode
rejection and common mode loading it down for these to work well.
Word of caution when it comes to levels, as the windowing filter used
causes shifts in noise-levels, so estimation of noise-levels becomes
a little bit tricky as you try to get the nitty gritty right, but
getting the overall shape view you already gained a lot with the
things you achieved.
A technique used to push further down into lower noise-levels is the
cross-correlation technique, where you split the signal into two
channels, each being exactly what you have now, and then rather than
squaring the output of the FFT from each channel, you multiply one
with the completment of the other, then average on those. This allows
you to supress the noise of each reference oscillator. You do not
have to go there from start, as you already make very useful
measurements, but I'm just suggesting what may lie up ahead.
Compared to some of the other sources, the Rigol SG does fairly well,
but then again, things can be even more quiet. For the XO you can see
the 15 dB/Oct slope as expected for flicker frequency. Try to locate
the source of the peaks you see and see if you can clean it up. The
XO seems to be a fairly good DUT for doing that.
Cheers,
Magnus
time-nuts mailing list -- time-nuts@lists.febo.com
To unsubscribe send an email to time-nuts-leave@lists.febo.com
Hi Erik,
On 2022-06-27 11:43, Erik Kaashoek via time-nuts wrote:
> Magnus, Bob,
> When the mixer is operating in the linear region for the DUT input
> (0dBm or lower), would it be possible to use a calibrated noise
> sources to do an extra verification of the noise level measurement?
Yes. NIST build such calibrators for exactly that purpose.
> Of course with a noise source you get 3dB as both sidebands fold.
Naturally.
> Verification steps:
> Verify the DUT output level is correctly brought to 0dB (using
> attenuators) using a calibrated spectrum analyzer
> Connect the DUT to the phase measurement setup and set the reference
> to a 500Hz offset to get a beat note and verify the beat note is
> registered at 0dB, change the DUT level some dB up and down to confirm
> its in a linear region.
> Measure the per Hz output power of a noise source using a calibrated
> spectrum analyzer and a noise marker set to 10MHz.
> Connect the noise source to the phase measurement setup and check if
> the noise level is measured at level measured by the spectrum analyzer
> + 3dB
> This should work if the RBW of the phase measurement is indeed set to
> 1Hz.
500 Hz is kind of arbitrary number you chose there. At the very least it
should be verified, but I would assume there is an underlying goal which
made you choose 500 Hz and that should be specified.
The RBW is not the noise BW. You need to correct the bin-bandwidth with
the specific noise bandwidth correction for the window-filter you use.
There is a neat article [1] on it which I also contributed into IEEE 1139.
>
> Another verification option may be to use the phase modulation of a
> signal generator. This can not check the effective noise bandwidth of
> the FFT but it can check linearity over the whole range.
> The output of the mixer is terminated with 50ohm so a factor of 10 in
> voltage should give a 20dB power step.
> When operation in the linear range the phase noise measurement setup
> should measure 20dB less with every factor 10 reduction in phase
> modulation depth where 90 degrees is equal to 100% modulation depth so
> equal to the signal you get when measuring a beat note.
> When measuring with modulation depth of 90,9,0.9,0.09 and 0.009
> degrees the measured level should step from 0,-20,-40, -60 to -80dB
For higher modulation depth it's very hard to do linear modulation well.
There is a different variant you can use, to inject a signal to form a
side-band signal next to the carrier. By creating a specific amplitude
compared to the carrier, and for a particular offset, it's trivial
exercise to know the AM and PM noise level. A single sideband sine will
divide equally to AM and PM, thus reducing 3 dB. Thus, setting a
side-band at 27 dB bellow the carrier, makes a -30 dBc phase modulation.
[1]
https://pure.mpg.de/pubman/faces/ViewItemOverviewPage.jsp?itemId=item_152164
Cheers,
Magnus
>
> Any feedback?
> Erik.
>
> On 26-6-2022 20:52, Magnus Danielson via time-nuts wrote:
>> Hi Erik!
>>
>> Great progress! Sure interesting to look at them phase-noise plots,
>> right? It's a really good tool in addition to the stability of ADEV
>> and friends.
>>
>> As I recall it, the ADE-1 is not documented to be isolated, but it is
>> very obvious when you look down the backside of it. However, it has
>> capacitive coupling and one should consider both common mode
>> rejection and common mode loading it down for these to work well.
>>
>> Word of caution when it comes to levels, as the windowing filter used
>> causes shifts in noise-levels, so estimation of noise-levels becomes
>> a little bit tricky as you try to get the nitty gritty right, but
>> getting the overall shape view you already gained a lot with the
>> things you achieved.
>>
>> A technique used to push further down into lower noise-levels is the
>> cross-correlation technique, where you split the signal into two
>> channels, each being exactly what you have now, and then rather than
>> squaring the output of the FFT from each channel, you multiply one
>> with the completment of the other, then average on those. This allows
>> you to supress the noise of each reference oscillator. You do not
>> have to go there from start, as you already make very useful
>> measurements, but I'm just suggesting what may lie up ahead.
>>
>> Compared to some of the other sources, the Rigol SG does fairly well,
>> but then again, things can be even more quiet. For the XO you can see
>> the 15 dB/Oct slope as expected for flicker frequency. Try to locate
>> the source of the peaks you see and see if you can clean it up. The
>> XO seems to be a fairly good DUT for doing that.
>>
>> Cheers,
>> Magnus
> _______________________________________________
> time-nuts mailing list -- time-nuts@lists.febo.com
> To unsubscribe send an email to time-nuts-leave@lists.febo.com
MD
Magnus Danielson
Mon, Jun 27, 2022 11:37 PM
Hi,
On 2022-06-27 18:06, Bob kb8tq via time-nuts wrote:
On Jun 27, 2022, at 1:43 AM, Erik Kaashoek via time-nuts time-nuts@lists.febo.com wrote:
Magnus, Bob,
When the mixer is operating in the linear region for the DUT input (0dBm or lower), would it be possible to use a calibrated noise sources to do an extra verification of the noise level measurement?
Of course with a noise source you get 3dB as both sidebands fold.
The “normal” approach is to put the mixer into saturation. This gives you the
best noise floor. It also does a bit better at separating AM noise from PM noise
(since you are trying to measure phase noise …..).
It could be worth mentioning that there are both linear and non-linear
AM->PM conversion as well as PM->AM conversion. They should be
perfectly separate, but real life circuits is unable to maintain perfect
symmetry.
Now, the AM noise tends to be stronger than the PM noise, so leakage
tends to be more severe in that direction, in that it causes more harm.
Running the signal into clipping is a non-linear way to cut away most of
the AM and maintain the PM, and such limiter detectors turns out to be
very good for FM reception (FM is just a variant of PM).
Modern high-speed ADC receivers with arctan CORDIC processing has
supperiour AM and PM separation and avoids much of those problems. Just
for reference, does not help here.
A linear filter provides AM->PM conversion when the upper side-band and
the lower side-band does not have the same gain for the same offset
frequency. This asymmetry converts AM (even) to PM (odd) as well as PM
(odd to AM (even), while AM to AM and PM to PM reduces amplitude. Even a
simple low-pass filter will do this, and to minimize it's effect the
cut-off frequency should be higher than the carrier frequency so it is
essentially flat and equal for both LSB and USB. Similarly should a
resonator be tuned to be on the carrier frequency, or the upper and
lower slopes will not match well.
Anyway, wanted to take the opportunity to explain a little more.
Verification steps:
Verify the DUT output level is correctly brought to 0dB (using attenuators) using a calibrated spectrum analyzer
Connect the DUT to the phase measurement setup and set the reference to a 500Hz offset to get a beat note and verify the beat note is registered at 0dB, change the DUT level some dB up and down to confirm its in a linear region.
Measure the per Hz output power of a noise source using a calibrated spectrum analyzer and a noise marker set to 10MHz.
Connect the noise source to the phase measurement setup and check if the noise level is measured at level measured by the spectrum analyzer + 3dB
This should work if the RBW of the phase measurement is indeed set to 1Hz.
If you do it this way, you still need to do conversion for the “one radian” reference
level that is used with phase modulation ( = the reference is not one cycle ). Yes
that’s a bit weird / obscure.
But then again, it's exactly what the standard says. We even rewrote
that section to clarify it.
Another verification option may be to use the phase modulation of a signal generator. This can not check the effective noise bandwidth of the FFT but it can check linearity over the whole range.
The output of the mixer is terminated with 50ohm so a factor of 10 in voltage should give a 20dB power step.
Audio termination at 50 ohms does not do much for isolation. (again a bit of an obscure
point). By terminating in 10X the nominal impedance ( so 500 ohms in this case) you get
another 6 db of gain in the system. Since this is ahead of the preamp, it might improve
your noise floor.
You may however want to have 50 Ohm termination for RF frequencies
(sum-frequency as well as leakage of carrier frequency). The NIST T&F
archive has illustrations of different mixer loading networks such that
it is high impedance at LF but low impedance at RF. Loading down the LF
which is detected signal makes no sense, as we amplify the voltage and
not the power received. We do want to terminate the RF so that it is not
reflected back into the mixerin a termal unstable fashion. If the RF
termination can be close enough to the mixer, other impedance than 50
Ohm can be considered for even more optimal result, but unless one knows
what one does, I'd say stick with 50 Ohms for that is what the mixer is
mast likely designed for.
Cheers,
Magnus
Bob
When operation in the linear range the phase noise measurement setup should measure 20dB less with every factor 10 reduction in phase modulation depth where 90 degrees is equal to 100% modulation depth so equal to the signal you get when measuring a beat note.
When measuring with modulation depth of 90,9,0.9,0.09 and 0.009 degrees the measured level should step from 0,-20,-40, -60 to -80dB
Any feedback?
Erik.
On 26-6-2022 20:52, Magnus Danielson via time-nuts wrote:
Hi Erik!
Great progress! Sure interesting to look at them phase-noise plots, right? It's a really good tool in addition to the stability of ADEV and friends.
As I recall it, the ADE-1 is not documented to be isolated, but it is very obvious when you look down the backside of it. However, it has capacitive coupling and one should consider both common mode rejection and common mode loading it down for these to work well.
Word of caution when it comes to levels, as the windowing filter used causes shifts in noise-levels, so estimation of noise-levels becomes a little bit tricky as you try to get the nitty gritty right, but getting the overall shape view you already gained a lot with the things you achieved.
A technique used to push further down into lower noise-levels is the cross-correlation technique, where you split the signal into two channels, each being exactly what you have now, and then rather than squaring the output of the FFT from each channel, you multiply one with the completment of the other, then average on those. This allows you to supress the noise of each reference oscillator. You do not have to go there from start, as you already make very useful measurements, but I'm just suggesting what may lie up ahead.
Compared to some of the other sources, the Rigol SG does fairly well, but then again, things can be even more quiet. For the XO you can see the 15 dB/Oct slope as expected for flicker frequency. Try to locate the source of the peaks you see and see if you can clean it up. The XO seems to be a fairly good DUT for doing that.
Cheers,
Magnus
time-nuts mailing list -- time-nuts@lists.febo.com
To unsubscribe send an email to time-nuts-leave@lists.febo.com
time-nuts mailing list -- time-nuts@lists.febo.com
To unsubscribe send an email to time-nuts-leave@lists.febo.com
Hi,
On 2022-06-27 18:06, Bob kb8tq via time-nuts wrote:
> Hi
>
>> On Jun 27, 2022, at 1:43 AM, Erik Kaashoek via time-nuts <time-nuts@lists.febo.com> wrote:
>>
>> Magnus, Bob,
>> When the mixer is operating in the linear region for the DUT input (0dBm or lower), would it be possible to use a calibrated noise sources to do an extra verification of the noise level measurement?
>> Of course with a noise source you get 3dB as both sidebands fold.
> The “normal” approach is to put the mixer into saturation. This gives you the
> best noise floor. It also does a bit better at separating AM noise from PM noise
> (since you are trying to measure phase noise …..).
It could be worth mentioning that there are both linear and non-linear
AM->PM conversion as well as PM->AM conversion. They *should* be
perfectly separate, but real life circuits is unable to maintain perfect
symmetry.
Now, the AM noise tends to be stronger than the PM noise, so leakage
tends to be more severe in that direction, in that it causes more harm.
Running the signal into clipping is a non-linear way to cut away most of
the AM and maintain the PM, and such limiter detectors turns out to be
very good for FM reception (FM is just a variant of PM).
Modern high-speed ADC receivers with arctan CORDIC processing has
supperiour AM and PM separation and avoids much of those problems. Just
for reference, does not help here.
A linear filter provides AM->PM conversion when the upper side-band and
the lower side-band does not have the same gain for the same offset
frequency. This asymmetry converts AM (even) to PM (odd) as well as PM
(odd to AM (even), while AM to AM and PM to PM reduces amplitude. Even a
simple low-pass filter will do this, and to minimize it's effect the
cut-off frequency should be higher than the carrier frequency so it is
essentially flat and equal for both LSB and USB. Similarly should a
resonator be tuned to be on the carrier frequency, or the upper and
lower slopes will not match well.
Anyway, wanted to take the opportunity to explain a little more.
>
>> Verification steps:
>> Verify the DUT output level is correctly brought to 0dB (using attenuators) using a calibrated spectrum analyzer
>> Connect the DUT to the phase measurement setup and set the reference to a 500Hz offset to get a beat note and verify the beat note is registered at 0dB, change the DUT level some dB up and down to confirm its in a linear region.
>> Measure the per Hz output power of a noise source using a calibrated spectrum analyzer and a noise marker set to 10MHz.
>> Connect the noise source to the phase measurement setup and check if the noise level is measured at level measured by the spectrum analyzer + 3dB
>> This should work if the RBW of the phase measurement is indeed set to 1Hz.
> If you do it this way, you still need to do conversion for the “one radian” reference
> level that is used with phase modulation ( = the reference is *not* one cycle ). Yes
> that’s a bit weird / obscure.
But then again, it's exactly what the standard says. We even rewrote
that section to clarify it.
>
>
>> Another verification option may be to use the phase modulation of a signal generator. This can not check the effective noise bandwidth of the FFT but it can check linearity over the whole range.
>> The output of the mixer is terminated with 50ohm so a factor of 10 in voltage should give a 20dB power step.
> Audio termination at 50 ohms does not do much for isolation. (again a bit of an obscure
> point). By terminating in 10X the nominal impedance ( so 500 ohms in this case) you get
> another 6 db of gain in the system. Since this is ahead of the preamp, it might improve
> your noise floor.
You may however want to have 50 Ohm termination for RF frequencies
(sum-frequency as well as leakage of carrier frequency). The NIST T&F
archive has illustrations of different mixer loading networks such that
it is high impedance at LF but low impedance at RF. Loading down the LF
which is detected signal makes no sense, as we amplify the voltage and
not the power received. We do want to terminate the RF so that it is not
reflected back into the mixerin a termal unstable fashion. If the RF
termination can be close enough to the mixer, other impedance than 50
Ohm can be considered for even more optimal result, but unless one knows
what one does, I'd say stick with 50 Ohms for that is what the mixer is
mast likely designed for.
Cheers,
Magnus
>
> Bob
>
>
>> When operation in the linear range the phase noise measurement setup should measure 20dB less with every factor 10 reduction in phase modulation depth where 90 degrees is equal to 100% modulation depth so equal to the signal you get when measuring a beat note.
>> When measuring with modulation depth of 90,9,0.9,0.09 and 0.009 degrees the measured level should step from 0,-20,-40, -60 to -80dB
>>
>> Any feedback?
>> Erik.
>>
>> On 26-6-2022 20:52, Magnus Danielson via time-nuts wrote:
>>> Hi Erik!
>>>
>>> Great progress! Sure interesting to look at them phase-noise plots, right? It's a really good tool in addition to the stability of ADEV and friends.
>>>
>>> As I recall it, the ADE-1 is not documented to be isolated, but it is very obvious when you look down the backside of it. However, it has capacitive coupling and one should consider both common mode rejection and common mode loading it down for these to work well.
>>>
>>> Word of caution when it comes to levels, as the windowing filter used causes shifts in noise-levels, so estimation of noise-levels becomes a little bit tricky as you try to get the nitty gritty right, but getting the overall shape view you already gained a lot with the things you achieved.
>>>
>>> A technique used to push further down into lower noise-levels is the cross-correlation technique, where you split the signal into two channels, each being exactly what you have now, and then rather than squaring the output of the FFT from each channel, you multiply one with the completment of the other, then average on those. This allows you to supress the noise of each reference oscillator. You do not have to go there from start, as you already make very useful measurements, but I'm just suggesting what may lie up ahead.
>>>
>>> Compared to some of the other sources, the Rigol SG does fairly well, but then again, things can be even more quiet. For the XO you can see the 15 dB/Oct slope as expected for flicker frequency. Try to locate the source of the peaks you see and see if you can clean it up. The XO seems to be a fairly good DUT for doing that.
>>>
>>> Cheers,
>>> Magnus
>> _______________________________________________
>> time-nuts mailing list -- time-nuts@lists.febo.com
>> To unsubscribe send an email to time-nuts-leave@lists.febo.com
> _______________________________________________
> time-nuts mailing list -- time-nuts@lists.febo.com
> To unsubscribe send an email to time-nuts-leave@lists.febo.com
