W
WarrenS
Fri, Dec 5, 2008 4:39 AM
Building a Sub Pico Second phase detector.
I was inspired to build this project yesterday after downloading and trying
Ulrich Bangert's 'DF6JB's Plotter 2008-10-10' program with its unbelievable
flexible user Interface capabilities. http://www.ulrich-bangert.de/html/downloads.html
What I needed was a Phase detector to use with the 'Plotter' program.
I decided to see what it takes to build a simple high resolution, sub Pico second,
linear phase logging detector using standard off the self IC's.
How If works:
The 5 or 10 MHz signal to be measured is buffered and toggles a
synchronous divide by two or four FF. This gives a 2.5MHz square wave and its complement.
Each side of the flip-flop connects to two of four XOR gates.
The 10 MHz reference signal goes thru a matching buffer and then to a pair of synchronous
Flip-Flops that provide a zero and a 90 deg phase shifted 2.5MHz square wave.
Each of these goes to two inputs of the XOR gates. The four XOR phase detectors
are connected to give four PWM type XOR phase detectors, each separate by 90 deg.
Each of the four XOR outputs are buffered by a cmos buffer gate
that has been powered by it's own 5 volt reference supply.
The buffer outputs then goes thru a multi-stage passive RC filter set up to
give two differential filtered PASSIVE + - 5 volt outputs, 90 deg apart.
Logging Data:
For the most flexible and best performance, two differential 16 plus bit ADC's
should be used, each connected to one of the dual differential Phase detectors.
After using the appropriate Analog RC filters, oversampling, digital filters, and digital
scaling, you get a file with a single column of data to feed "Plotter" the phase
difference of the two 10 MHz signals.
The Data scaling and processing:
For simple controlled short term or lower resolution data taking a PC Multimeter,
if it is isolated so that you can use it differentially will work. If not you need to add a differential amp.
For best performance, process the phase data from the two differential phase detectors
through two identical digital filter algorithms.
Doing this real time on a PC or after all the data is recorder on a XL spread sheet both work for me.
Besides the filtering, the spread sheet or PC needs to also do the linearizing by
( K1* Phase1_Data) + (K2 * Phase2_Data).
K1 and K2 are the sine value of their respective Phase detectors.
One of the several tricks to why this can provide orders of magnitude better
performance than is generally obtained from similar type phase detectors
is because of the four matched Phase detectors that are added, subtracted
and combined and linerized in such a way as to cancel the type of errors
found in single XOR phase detectors.
Preliminary Performance
The noise floor that I have seen while feeding the same low noise osc, to both inputs,
is around 10 uv peak to peak at low Bandwidths, at zero phase, using a 6 digit DVM
with a slow filter which corresponds to <<1 ps. Test are still underway to see what the
lower limit is, and what the sensitivity to the environment is.
This is just the start of an on going learning project, It is just at the breadboard stage and
needs to be verified, critiqued, cleaned up and packaged up.
Noted that when working with sub ps resolution, extra care needs to be taken.
Although it looks to be a standard digital circuit, It is not. It is a very sensitive Analog circuit
capable of giving 1 part in a million type of resolution. It can resolve path distance changes
in the 1/100 to 1/1000 of an inch, and needs to be built with care and 'respect'.
Another use (beside watching just how noisy your "GOOD " osc is),
It can be used to compare and adjust the freq differences between two osc
very quickly and with more resolution than most can use.
1 E-12 freq difference gave several counts per second change on
the DVM, and with the DVM updating at several times a second,
it made fine freq adjustments much easer than slower monitoring ways.
If you know of other simple high resolution phase detectors,
or see any problems or improvements
with the idea, I'd like to hear from you.
Have fun
WarrenS
Building a Sub Pico Second phase detector.
I was inspired to build this project yesterday after downloading and trying
Ulrich Bangert's 'DF6JB's Plotter 2008-10-10' program with its unbelievable
flexible user Interface capabilities. http://www.ulrich-bangert.de/html/downloads.html
What I needed was a Phase detector to use with the 'Plotter' program.
I decided to see what it takes to build a simple high resolution, sub Pico second,
linear phase logging detector using standard off the self IC's.
How If works:
The 5 or 10 MHz signal to be measured is buffered and toggles a
synchronous divide by two or four FF. This gives a 2.5MHz square wave and its complement.
Each side of the flip-flop connects to two of four XOR gates.
The 10 MHz reference signal goes thru a matching buffer and then to a pair of synchronous
Flip-Flops that provide a zero and a 90 deg phase shifted 2.5MHz square wave.
Each of these goes to two inputs of the XOR gates. The four XOR phase detectors
are connected to give four PWM type XOR phase detectors, each separate by 90 deg.
Each of the four XOR outputs are buffered by a cmos buffer gate
that has been powered by it's own 5 volt reference supply.
The buffer outputs then goes thru a multi-stage passive RC filter set up to
give two differential filtered PASSIVE + - 5 volt outputs, 90 deg apart.
Logging Data:
For the most flexible and best performance, two differential 16 plus bit ADC's
should be used, each connected to one of the dual differential Phase detectors.
After using the appropriate Analog RC filters, oversampling, digital filters, and digital
scaling, you get a file with a single column of data to feed "Plotter" the phase
difference of the two 10 MHz signals.
The Data scaling and processing:
For simple controlled short term or lower resolution data taking a PC Multimeter,
if it is isolated so that you can use it differentially will work. If not you need to add a differential amp.
For best performance, process the phase data from the two differential phase detectors
through two identical digital filter algorithms.
Doing this real time on a PC or after all the data is recorder on a XL spread sheet both work for me.
Besides the filtering, the spread sheet or PC needs to also do the linearizing by
( K1* Phase1_Data) + (K2 * Phase2_Data).
K1 and K2 are the sine value of their respective Phase detectors.
One of the several tricks to why this can provide orders of magnitude better
performance than is generally obtained from similar type phase detectors
is because of the four matched Phase detectors that are added, subtracted
and combined and linerized in such a way as to cancel the type of errors
found in single XOR phase detectors.
Preliminary Performance
The noise floor that I have seen while feeding the same low noise osc, to both inputs,
is around 10 uv peak to peak at low Bandwidths, at zero phase, using a 6 digit DVM
with a slow filter which corresponds to <<1 ps. Test are still underway to see what the
lower limit is, and what the sensitivity to the environment is.
This is just the start of an on going learning project, It is just at the breadboard stage and
needs to be verified, critiqued, cleaned up and packaged up.
Noted that when working with sub ps resolution, extra care needs to be taken.
Although it looks to be a standard digital circuit, It is not. It is a very sensitive Analog circuit
capable of giving 1 part in a million type of resolution. It can resolve path distance changes
in the 1/100 to 1/1000 of an inch, and needs to be built with care and 'respect'.
Another use (beside watching just how noisy your "GOOD " osc is),
It can be used to compare and adjust the freq differences between two osc
very quickly and with more resolution than most can use.
1 E-12 freq difference gave several counts per second change on
the DVM, and with the DVM updating at several times a second,
it made fine freq adjustments much easer than slower monitoring ways.
If you know of other simple high resolution phase detectors,
or see any problems or improvements
with the idea, I'd like to hear from you.
Have fun
WarrenS
BG
Bruce Griffiths
Fri, Dec 5, 2008 5:09 AM
Building a Sub Pico Second phase detector.
I was inspired to build this project yesterday after downloading and trying
Ulrich Bangert's 'DF6JB's Plotter 2008-10-10' program with its unbelievable
flexible user Interface capabilities. http://www.ulrich-bangert.de/html/downloads.html
What I needed was a Phase detector to use with the 'Plotter' program.
I decided to see what it takes to build a simple high resolution, sub Pico second,
linear phase logging detector using standard off the self IC's.
How If works:
The 5 or 10 MHz signal to be measured is buffered and toggles a
synchronous divide by two or four FF. This gives a 2.5MHz square wave and its complement.
Each side of the flip-flop connects to two of four XOR gates.
The 10 MHz reference signal goes thru a matching buffer and then to a pair of synchronous
Flip-Flops that provide a zero and a 90 deg phase shifted 2.5MHz square wave.
Each of these goes to two inputs of the XOR gates. The four XOR phase detectors
are connected to give four PWM type XOR phase detectors, each separate by 90 deg.
Each of the four XOR outputs are buffered by a cmos buffer gate
that has been powered by it's own 5 volt reference supply.
The buffer outputs then goes thru a multi-stage passive RC filter set up to
give two differential filtered PASSIVE + - 5 volt outputs, 90 deg apart.
Logging Data:
For the most flexible and best performance, two differential 16 plus bit ADC's
should be used, each connected to one of the dual differential Phase detectors.
After using the appropriate Analog RC filters, oversampling, digital filters, and digital
scaling, you get a file with a single column of data to feed "Plotter" the phase
difference of the two 10 MHz signals.
The Data scaling and processing:
For simple controlled short term or lower resolution data taking a PC Multimeter,
if it is isolated so that you can use it differentially will work. If not you need to add a differential amp.
For best performance, process the phase data from the two differential phase detectors
through two identical digital filter algorithms.
Doing this real time on a PC or after all the data is recorder on a XL spread sheet both work for me.
Besides the filtering, the spread sheet or PC needs to also do the linearizing by
( K1* Phase1_Data) + (K2 * Phase2_Data).
K1 and K2 are the sine value of their respective Phase detectors.
One of the several tricks to why this can provide orders of magnitude better
performance than is generally obtained from similar type phase detectors
is because of the four matched Phase detectors that are added, subtracted
and combined and linerized in such a way as to cancel the type of errors
found in single XOR phase detectors.
Preliminary Performance
The noise floor that I have seen while feeding the same low noise osc, to both inputs,
is around 10 uv peak to peak at low Bandwidths, at zero phase, using a 6 digit DVM
with a slow filter which corresponds to <<1 ps. Test are still underway to see what the
lower limit is, and what the sensitivity to the environment is.
This is just the start of an on going learning project, It is just at the breadboard stage and
needs to be verified, critiqued, cleaned up and packaged up.
Noted that when working with sub ps resolution, extra care needs to be taken.
Although it looks to be a standard digital circuit, It is not. It is a very sensitive Analog circuit
capable of giving 1 part in a million type of resolution. It can resolve path distance changes
in the 1/100 to 1/1000 of an inch, and needs to be built with care and 'respect'.
Another use (beside watching just how noisy your "GOOD " osc is),
It can be used to compare and adjust the freq differences between two osc
very quickly and with more resolution than most can use.
1 E-12 freq difference gave several counts per second change on
the DVM, and with the DVM updating at several times a second,
it made fine freq adjustments much easer than slower monitoring ways.
If you know of other simple high resolution phase detectors,
or see any problems or improvements
with the idea, I'd like to hear from you.
Have fun
WarrenS
Warren
Since HCMOS buffers typically have about 4ps of random propagation delay
jitter and ACMOS devices typically have about 1ps of RJ this isnt too
surprising.
Newer logic families may have even lower random jitter.
Doesn't this phase detector, like all digital phase detectors, have
significant non linearity at the ends of its range?
In the case of an XOR gate phase detector this is caused by the finite
slew rate of the gate output.
With the quadrature phased outputs at least 2 of the phase detectors
will be operating in the linear part of their range.
The particular pair that are linear depends on the relative phase of the
2 inputs.
One or more of the ubiquitous 24 bit resolution sigma delta ADCs with
differential inputs and a reference derived from the XOR power supply,
will for CMOS XOR gates probably be a relatively inexpensive replacement
for the DVM int he final system.
If one used an FPGA or CPLD for this as the internal crosstalk may limit
performance to a few tens of picosec noise for 1 sec averaging unless
differential I/O logic such as LDVS, ECL etc are used.
Although the circuit is simple enough not to warrant an FPGA it would be
useful to have programmable dividers for each input to allow comparison
of input frequencies that arent either nominally equal or have a
frequency ratio of 2:1. Using external retiming flipflops should cure
the crosstalk problem with such a divider.
In practice such a divider should perhaps be an external device with its
own power supply and enclosure.
Such a divider can be used to increase the effective range of the phase
detector at the expense of its resolution.
Bruce
WarrenS wrote:
> Building a Sub Pico Second phase detector.
>
> I was inspired to build this project yesterday after downloading and trying
> Ulrich Bangert's 'DF6JB's Plotter 2008-10-10' program with its unbelievable
> flexible user Interface capabilities. http://www.ulrich-bangert.de/html/downloads.html
> What I needed was a Phase detector to use with the 'Plotter' program.
> I decided to see what it takes to build a simple high resolution, sub Pico second,
> linear phase logging detector using standard off the self IC's.
>
> How If works:
> The 5 or 10 MHz signal to be measured is buffered and toggles a
> synchronous divide by two or four FF. This gives a 2.5MHz square wave and its complement.
> Each side of the flip-flop connects to two of four XOR gates.
>
> The 10 MHz reference signal goes thru a matching buffer and then to a pair of synchronous
> Flip-Flops that provide a zero and a 90 deg phase shifted 2.5MHz square wave.
> Each of these goes to two inputs of the XOR gates. The four XOR phase detectors
> are connected to give four PWM type XOR phase detectors, each separate by 90 deg.
>
> Each of the four XOR outputs are buffered by a cmos buffer gate
> that has been powered by it's own 5 volt reference supply.
> The buffer outputs then goes thru a multi-stage passive RC filter set up to
> give two differential filtered PASSIVE + - 5 volt outputs, 90 deg apart.
>
> Logging Data:
> For the most flexible and best performance, two differential 16 plus bit ADC's
> should be used, each connected to one of the dual differential Phase detectors.
> After using the appropriate Analog RC filters, oversampling, digital filters, and digital
> scaling, you get a file with a single column of data to feed "Plotter" the phase
> difference of the two 10 MHz signals.
>
> The Data scaling and processing:
> For simple controlled short term or lower resolution data taking a PC Multimeter,
> if it is isolated so that you can use it differentially will work. If not you need to add a differential amp.
> For best performance, process the phase data from the two differential phase detectors
> through two identical digital filter algorithms.
> Doing this real time on a PC or after all the data is recorder on a XL spread sheet both work for me.
> Besides the filtering, the spread sheet or PC needs to also do the linearizing by
> ( K1* Phase1_Data) + (K2 * Phase2_Data).
> K1 and K2 are the sine value of their respective Phase detectors.
>
> One of the several tricks to why this can provide orders of magnitude better
> performance than is generally obtained from similar type phase detectors
> is because of the four matched Phase detectors that are added, subtracted
> and combined and linerized in such a way as to cancel the type of errors
> found in single XOR phase detectors.
>
> Preliminary Performance
> The noise floor that I have seen while feeding the same low noise osc, to both inputs,
> is around 10 uv peak to peak at low Bandwidths, at zero phase, using a 6 digit DVM
> with a slow filter which corresponds to <<1 ps. Test are still underway to see what the
> lower limit is, and what the sensitivity to the environment is.
>
> This is just the start of an on going learning project, It is just at the breadboard stage and
> needs to be verified, critiqued, cleaned up and packaged up.
> Noted that when working with sub ps resolution, extra care needs to be taken.
> Although it looks to be a standard digital circuit, It is not. It is a very sensitive Analog circuit
> capable of giving 1 part in a million type of resolution. It can resolve path distance changes
> in the 1/100 to 1/1000 of an inch, and needs to be built with care and 'respect'.
>
> Another use (beside watching just how noisy your "GOOD " osc is),
> It can be used to compare and adjust the freq differences between two osc
> very quickly and with more resolution than most can use.
> 1 E-12 freq difference gave several counts per second change on
> the DVM, and with the DVM updating at several times a second,
> it made fine freq adjustments much easer than slower monitoring ways.
>
>
> If you know of other simple high resolution phase detectors,
> or see any problems or improvements
> with the idea, I'd like to hear from you.
>
> Have fun
> WarrenS
>
Warren
Since HCMOS buffers typically have about 4ps of random propagation delay
jitter and ACMOS devices typically have about 1ps of RJ this isnt too
surprising.
Newer logic families may have even lower random jitter.
Doesn't this phase detector, like all digital phase detectors, have
significant non linearity at the ends of its range?
In the case of an XOR gate phase detector this is caused by the finite
slew rate of the gate output.
With the quadrature phased outputs at least 2 of the phase detectors
will be operating in the linear part of their range.
The particular pair that are linear depends on the relative phase of the
2 inputs.
One or more of the ubiquitous 24 bit resolution sigma delta ADCs with
differential inputs and a reference derived from the XOR power supply,
will for CMOS XOR gates probably be a relatively inexpensive replacement
for the DVM int he final system.
If one used an FPGA or CPLD for this as the internal crosstalk may limit
performance to a few tens of picosec noise for 1 sec averaging unless
differential I/O logic such as LDVS, ECL etc are used.
Although the circuit is simple enough not to warrant an FPGA it would be
useful to have programmable dividers for each input to allow comparison
of input frequencies that arent either nominally equal or have a
frequency ratio of 2:1. Using external retiming flipflops should cure
the crosstalk problem with such a divider.
In practice such a divider should perhaps be an external device with its
own power supply and enclosure.
Such a divider can be used to increase the effective range of the phase
detector at the expense of its resolution.
Bruce
JM
Joseph M Gwinn
Mon, Dec 8, 2008 10:01 PM
People used passive mixers driving electromechanical stripchart recorders
to compare high-stability oscillators in the good old days.
One assumes that there is a purely analog approach to measurement of
picosecond changes in delay at 10 MHz using a single oscillator, but I
have not seen any methods described, probably because the relevant
articles appeared many decades ago.
Can anyone suggest some articles to read?
Thanks,
Joe Gwinn
time-nuts-bounces@febo.com wrote on 12/04/2008 11:39:55 PM:
Building a Sub Pico Second phase detector.
I was inspired to build this project yesterday after
downloading and trying
Ulrich Bangert's 'DF6JB's Plotter 2008-10-10' program with its
unbelievable
flexible user Interface capabilities. http://www.ulrich-
bangert.de/html/downloads.html
What I needed was a Phase detector to use with the 'Plotter' program.
I decided to see what it takes to build a simple high
resolution, sub Pico second,
linear phase logging detector using standard off the self IC's.
How If works:
The 5 or 10 MHz signal to be measured is buffered and toggles a
synchronous divide by two or four FF. This gives a 2.5MHz
square wave and its complement.
Each side of the flip-flop connects to two of four XOR gates.
The 10 MHz reference signal goes thru a matching buffer and
then to a pair of synchronous
Flip-Flops that provide a zero and a 90 deg phase shifted 2.
5MHz square wave.
Each of these goes to two inputs of the XOR gates. The four XOR
phase detectors
are connected to give four PWM type XOR phase detectors, each
separate by 90 deg.
Each of the four XOR outputs are buffered by a cmos buffer gate
that has been powered by it's own 5 volt reference supply.
The buffer outputs then goes thru a multi-stage passive RC
filter set up to
give two differential filtered PASSIVE + - 5 volt outputs, 90 deg apart.
Logging Data:
For the most flexible and best performance, two differential
16 plus bit ADC's
should be used, each connected to one of the dual differential
Phase detectors.
After using the appropriate Analog RC filters, oversampling,
digital filters, and digital
scaling, you get a file with a single column of data to feed
"Plotter" the phase
difference of the two 10 MHz signals.
The Data scaling and processing:
For simple controlled short term or lower resolution data
taking a PC Multimeter,
if it is isolated so that you can use it differentially will
work. If not you need to add a differential amp.
For best performance, process the phase data from the two
differential phase detectors
through two identical digital filter algorithms.
Doing this real time on a PC or after all the data is recorder
on a XL spread sheet both work for me.
Besides the filtering, the spread sheet or PC needs to also do
the linearizing by
( K1* Phase1_Data) + (K2 * Phase2_Data).
K1 and K2 are the sine value of their respective Phase detectors.
One of the several tricks to why this can provide orders of
magnitude better
performance than is generally obtained from similar type phase detectors
is because of the four matched Phase detectors that are added,
subtracted
and combined and linerized in such a way as to cancel the type of errors
found in single XOR phase detectors.
Preliminary Performance
The noise floor that I have seen while feeding the same low
noise osc, to both inputs,
is around 10 uv peak to peak at low Bandwidths, at zero phase,
using a 6 digit DVM
with a slow filter which corresponds to <<1 ps. Test are still
underway to see what the
lower limit is, and what the sensitivity to the environment is.
This is just the start of an on going learning project, It is
just at the breadboard stage and
needs to be verified, critiqued, cleaned up and packaged up.
Noted that when working with sub ps resolution, extra care
needs to be taken.
Although it looks to be a standard digital circuit, It is not.
It is a very sensitive Analog circuit
capable of giving 1 part in a million type of resolution. It
can resolve path distance changes
in the 1/100 to 1/1000 of an inch, and needs to be built with
care and 'respect'.
Another use (beside watching just how noisy your "GOOD " osc is),
It can be used to compare and adjust the freq differences
between two osc
very quickly and with more resolution than most can use.
1 E-12 freq difference gave several counts per second change on
the DVM, and with the DVM updating at several times a second,
it made fine freq adjustments much easer than slower monitoring ways.
If you know of other simple high resolution phase detectors,
or see any problems or improvements
with the idea, I'd like to hear from you.
Have fun
WarrenS
time-nuts mailing list -- time-nuts@febo.com
To unsubscribe, go to https://www.febo.com/cgi-
bin/mailman/listinfo/time-nuts
and follow the instructions there.
People used passive mixers driving electromechanical stripchart recorders
to compare high-stability oscillators in the good old days.
One assumes that there is a purely analog approach to measurement of
picosecond changes in delay at 10 MHz using a single oscillator, but I
have not seen any methods described, probably because the relevant
articles appeared many decades ago.
Can anyone suggest some articles to read?
Thanks,
Joe Gwinn
time-nuts-bounces@febo.com wrote on 12/04/2008 11:39:55 PM:
>
> Building a Sub Pico Second phase detector.
>
> I was inspired to build this project yesterday after
> downloading and trying
> Ulrich Bangert's 'DF6JB's Plotter 2008-10-10' program with its
> unbelievable
> flexible user Interface capabilities. http://www.ulrich-
> bangert.de/html/downloads.html
> What I needed was a Phase detector to use with the 'Plotter' program.
> I decided to see what it takes to build a simple high
> resolution, sub Pico second,
> linear phase logging detector using standard off the self IC's.
>
> How If works:
> The 5 or 10 MHz signal to be measured is buffered and toggles a
> synchronous divide by two or four FF. This gives a 2.5MHz
> square wave and its complement.
> Each side of the flip-flop connects to two of four XOR gates.
>
> The 10 MHz reference signal goes thru a matching buffer and
> then to a pair of synchronous
> Flip-Flops that provide a zero and a 90 deg phase shifted 2.
> 5MHz square wave.
> Each of these goes to two inputs of the XOR gates. The four XOR
> phase detectors
> are connected to give four PWM type XOR phase detectors, each
> separate by 90 deg.
>
> Each of the four XOR outputs are buffered by a cmos buffer gate
> that has been powered by it's own 5 volt reference supply.
> The buffer outputs then goes thru a multi-stage passive RC
> filter set up to
> give two differential filtered PASSIVE + - 5 volt outputs, 90 deg apart.
>
> Logging Data:
> For the most flexible and best performance, two differential
> 16 plus bit ADC's
> should be used, each connected to one of the dual differential
> Phase detectors.
> After using the appropriate Analog RC filters, oversampling,
> digital filters, and digital
> scaling, you get a file with a single column of data to feed
> "Plotter" the phase
> difference of the two 10 MHz signals.
>
> The Data scaling and processing:
> For simple controlled short term or lower resolution data
> taking a PC Multimeter,
> if it is isolated so that you can use it differentially will
> work. If not you need to add a differential amp.
> For best performance, process the phase data from the two
> differential phase detectors
> through two identical digital filter algorithms.
> Doing this real time on a PC or after all the data is recorder
> on a XL spread sheet both work for me.
> Besides the filtering, the spread sheet or PC needs to also do
> the linearizing by
> ( K1* Phase1_Data) + (K2 * Phase2_Data).
> K1 and K2 are the sine value of their respective Phase detectors.
>
> One of the several tricks to why this can provide orders of
> magnitude better
> performance than is generally obtained from similar type phase detectors
> is because of the four matched Phase detectors that are added,
> subtracted
> and combined and linerized in such a way as to cancel the type of errors
> found in single XOR phase detectors.
>
> Preliminary Performance
> The noise floor that I have seen while feeding the same low
> noise osc, to both inputs,
> is around 10 uv peak to peak at low Bandwidths, at zero phase,
> using a 6 digit DVM
> with a slow filter which corresponds to <<1 ps. Test are still
> underway to see what the
> lower limit is, and what the sensitivity to the environment is.
>
> This is just the start of an on going learning project, It is
> just at the breadboard stage and
> needs to be verified, critiqued, cleaned up and packaged up.
> Noted that when working with sub ps resolution, extra care
> needs to be taken.
> Although it looks to be a standard digital circuit, It is not.
> It is a very sensitive Analog circuit
> capable of giving 1 part in a million type of resolution. It
> can resolve path distance changes
> in the 1/100 to 1/1000 of an inch, and needs to be built with
> care and 'respect'.
>
> Another use (beside watching just how noisy your "GOOD " osc is),
> It can be used to compare and adjust the freq differences
> between two osc
> very quickly and with more resolution than most can use.
> 1 E-12 freq difference gave several counts per second change on
> the DVM, and with the DVM updating at several times a second,
> it made fine freq adjustments much easer than slower monitoring ways.
>
>
> If you know of other simple high resolution phase detectors,
> or see any problems or improvements
> with the idea, I'd like to hear from you.
>
> Have fun
> WarrenS
> _______________________________________________
> time-nuts mailing list -- time-nuts@febo.com
> To unsubscribe, go to https://www.febo.com/cgi-
> bin/mailman/listinfo/time-nuts
> and follow the instructions there.
BG
Bruce Griffiths
Mon, Dec 8, 2008 10:53 PM
People used passive mixers driving electromechanical stripchart recorders
to compare high-stability oscillators in the good old days.
One assumes that there is a purely analog approach to measurement of
picosecond changes in delay at 10 MHz using a single oscillator, but I
have not seen any methods described, probably because the relevant
articles appeared many decades ago.
Can anyone suggest some articles to read?
Thanks,
Joe Gwinn
Joe
Although one could in principle do this with a single diode double
balanced mixer used as a phase detector all one may end up measuring is
the effect of ambient temperature changes on the mixer phase shift.
Lower mixer phase shift tempcos are possible if the RF port is unsaturated.
A classical dual mixer system is probably better in that with matched
tempco mixers maintained at the same temperature the differential phase
shift tempco should (with careful matching) be lower.
Other than the numerous classical papers on dual mixer systems and the
occasionl NIST paper that have some mixer phase shift tempco data
(albeit sparse), I am not aware of any specific papers.
A purely analog approach to phase shift measurement has to be more
difficult than a hybrid one using a pair of low frequency ADCs (eg high
end sound card).
Bruce
Joseph M Gwinn wrote:
> People used passive mixers driving electromechanical stripchart recorders
> to compare high-stability oscillators in the good old days.
>
> One assumes that there is a purely analog approach to measurement of
> picosecond changes in delay at 10 MHz using a single oscillator, but I
> have not seen any methods described, probably because the relevant
> articles appeared many decades ago.
>
> Can anyone suggest some articles to read?
>
> Thanks,
>
> Joe Gwinn
>
Joe
Although one could in principle do this with a single diode double
balanced mixer used as a phase detector all one may end up measuring is
the effect of ambient temperature changes on the mixer phase shift.
Lower mixer phase shift tempcos are possible if the RF port is unsaturated.
A classical dual mixer system is probably better in that with matched
tempco mixers maintained at the same temperature the differential phase
shift tempco should (with careful matching) be lower.
Other than the numerous classical papers on dual mixer systems and the
occasionl NIST paper that have some mixer phase shift tempco data
(albeit sparse), I am not aware of any specific papers.
A purely analog approach to phase shift measurement has to be more
difficult than a hybrid one using a pair of low frequency ADCs (eg high
end sound card).
Bruce
JM
Joseph M Gwinn
Mon, Dec 8, 2008 11:37 PM
People used passive mixers driving electromechanical stripchart
to compare high-stability oscillators in the good old days.
One assumes that there is a purely analog approach to measurement of
picosecond changes in delay at 10 MHz using a single oscillator, but I
have not seen any methods described, probably because the relevant
articles appeared many decades ago.
Can anyone suggest some articles to read?
Thanks,
Joe Gwinn
Joe
Although one could in principle do this with a single diode double
balanced mixer used as a phase detector all one may end up measuring is
the effect of ambient temperature changes on the mixer phase shift.
Lower mixer phase shift tempcos are possible if the RF port is
unsaturated.
Single diode? Why wouldn't one use a standard (MiniCircuits or the like)
four-diode two-transformer double-balanced mixer as the phase detector?
Many mixers have IF response down to DC.
A classical dual mixer system is probably better in that with matched
tempco mixers maintained at the same temperature the differential phase
shift tempco should (with careful matching) be lower.
Dual mixer as in DMTD (dual mixer time difference) would certainly work,
but is pretty complex and temperature sensitive.
I did use a loaner Symmetricom 5120A (a full digital DMTD implementation)
to make some measurements six months ago, and after a few days of
continuous operation it had settled to the point that one could see 0.01
pS changes. (And touching one of the BNC connectors caused a 1-3 pS
jump.) This instrument costs about $30K, and is intended more for
measuring phase noise and allan variance than delay changes.
Anyway, I have to wonder what people did before DMTD was invented.
Other than the numerous classical papers on dual mixer systems and the
occasionl NIST paper that have some mixer phase shift tempco data
(albeit sparse), I am not aware of any specific papers.
I've read many or most of the classical DMTD papers, and have seen various
passing estimates that diode-ring mixers have a temperature sensitivity of
8 to 10 pS per degree C. (I recall your figure was 10 pS/K.) I assume
that the DC offset also varies with themerature and drive signal
amplitude.
A purely analog approach to phase shift measurement has to be more
difficult than a hybrid one using a pair of low frequency ADCs (eg high
end sound card).
Is the sound-card approach workable at the millidegree to microdegree
level, if the change is spread out over an hour? One picosecond at 10 MHz
is 3.6 millidegrees of phase.
Joe
Bruce,
time-nuts-bounces@febo.com wrote on 12/08/2008 05:53:08 PM:
> Joseph M Gwinn wrote:
> > People used passive mixers driving electromechanical stripchart
recorders
> > to compare high-stability oscillators in the good old days.
> >
> > One assumes that there is a purely analog approach to measurement of
> > picosecond changes in delay at 10 MHz using a single oscillator, but I
> > have not seen any methods described, probably because the relevant
> > articles appeared many decades ago.
> >
> > Can anyone suggest some articles to read?
> >
> > Thanks,
> >
> > Joe Gwinn
> >
> Joe
>
> Although one could in principle do this with a single diode double
> balanced mixer used as a phase detector all one may end up measuring is
> the effect of ambient temperature changes on the mixer phase shift.
> Lower mixer phase shift tempcos are possible if the RF port is
> unsaturated.
Single diode? Why wouldn't one use a standard (MiniCircuits or the like)
four-diode two-transformer double-balanced mixer as the phase detector?
Many mixers have IF response down to DC.
> A classical dual mixer system is probably better in that with matched
> tempco mixers maintained at the same temperature the differential phase
> shift tempco should (with careful matching) be lower.
Dual mixer as in DMTD (dual mixer time difference) would certainly work,
but is pretty complex and temperature sensitive.
I did use a loaner Symmetricom 5120A (a full digital DMTD implementation)
to make some measurements six months ago, and after a few days of
continuous operation it had settled to the point that one could see 0.01
pS changes. (And touching one of the BNC connectors caused a 1-3 pS
jump.) This instrument costs about $30K, and is intended more for
measuring phase noise and allan variance than delay changes.
Anyway, I have to wonder what people did before DMTD was invented.
> Other than the numerous classical papers on dual mixer systems and the
> occasionl NIST paper that have some mixer phase shift tempco data
> (albeit sparse), I am not aware of any specific papers.
I've read many or most of the classical DMTD papers, and have seen various
passing estimates that diode-ring mixers have a temperature sensitivity of
8 to 10 pS per degree C. (I recall your figure was 10 pS/K.) I assume
that the DC offset also varies with themerature and drive signal
amplitude.
> A purely analog approach to phase shift measurement has to be more
> difficult than a hybrid one using a pair of low frequency ADCs (eg high
> end sound card).
Is the sound-card approach workable at the millidegree to microdegree
level, if the change is spread out over an hour? One picosecond at 10 MHz
is 3.6 millidegrees of phase.
Joe
BG
Bruce Griffiths
Tue, Dec 9, 2008 12:12 AM
People used passive mixers driving electromechanical stripchart
to compare high-stability oscillators in the good old days.
One assumes that there is a purely analog approach to measurement of
picosecond changes in delay at 10 MHz using a single oscillator, but I
have not seen any methods described, probably because the relevant
articles appeared many decades ago.
Can anyone suggest some articles to read?
Thanks,
Joe Gwinn
Joe
Although one could in principle do this with a single diode double
balanced mixer used as a phase detector all one may end up measuring is
the effect of ambient temperature changes on the mixer phase shift.
Lower mixer phase shift tempcos are possible if the RF port is
unsaturated.
Single diode? Why wouldn't one use a standard (MiniCircuits or the like)
four-diode two-transformer double-balanced mixer as the phase detector?
Many mixers have IF response down to DC.
Oops, I meant "single diode type double balanced mixer style phase
detector".
A classical dual mixer system is probably better in that with matched
tempco mixers maintained at the same temperature the differential phase
shift tempco should (with careful matching) be lower.
Dual mixer as in DMTD (dual mixer time difference) would certainly work,
but is pretty complex and temperature sensitive.
I did use a loaner Symmetricom 5120A (a full digital DMTD implementation)
to make some measurements six months ago, and after a few days of
continuous operation it had settled to the point that one could see 0.01
pS changes. (And touching one of the BNC connectors caused a 1-3 pS
jump.) This instrument costs about $30K, and is intended more for
measuring phase noise and allan variance than delay changes.
Anyway, I have to wonder what people did before DMTD was invented.
Other than the numerous classical papers on dual mixer systems and the
occasionl NIST paper that have some mixer phase shift tempco data
(albeit sparse), I am not aware of any specific papers.
I've read many or most of the classical DMTD papers, and have seen various
passing estimates that diode-ring mixers have a temperature sensitivity of
8 to 10 pS per degree C. (I recall your figure was 10 pS/K.) I assume
that the DC offset also varies with themerature and drive signal
amplitude.
The only reference I have on the offset tempco is a miniciruits
application note from which one can deduce that the equivalent phase
shift tempco associated with the offset tempco is a few hundred
femtosec/C (@ 10MHz +7dBm) at some temperatures for the particular mixer
used. The graph also indicated (if you are lucky) that the offset tempco
may be zero at around 20C.
A NIST paper indicated that mixer phase shift tempco was around 10x
lower if the Rf port was unsaturated. It also indicated that the mixer
phase shift tempco is much lower if the input frequency is 100MHz rather
than 10MHz. This was one reason given for shifting to 100MHz DMTD systems.
A purely analog approach to phase shift measurement has to be more
difficult than a hybrid one using a pair of low frequency ADCs (eg high
end sound card).
Is the sound-card approach workable at the millidegree to microdegree
level, if the change is spread out over an hour? One picosecond at 10 MHz
is 3.6 millidegrees of phase.
Joe
Preliminary (non optimum) tests by Ulrich indicate that picosecond
stability for times up to 100sec is very easy to achieve.
Beyond that mixer phase shift tempco mismatch may be significant.
ADEV noise level of around 2E-14/Tau (1s < tau <100s).
Haven't yet seen have data for longer tau.
With identical beat frequency outputs crosstalk between channels within
the sound card shouldn't be a great problem.
In any case its very easy to measure the crosstalk transfer function.
One concern particularly for low beat frequencies is the phase shift in
the sound card input coupling capacitors (usually electrolytics).
It should be easy to test the sound card phase shift stability for this
application by driving both inputs from the same signal source.
Bruce
Joseph M Gwinn wrote:
> Bruce,
>
>
> time-nuts-bounces@febo.com wrote on 12/08/2008 05:53:08 PM:
>
>
>> Joseph M Gwinn wrote:
>>
>>> People used passive mixers driving electromechanical stripchart
>>>
> recorders
>
>>> to compare high-stability oscillators in the good old days.
>>>
>>> One assumes that there is a purely analog approach to measurement of
>>> picosecond changes in delay at 10 MHz using a single oscillator, but I
>>>
>
>
>>> have not seen any methods described, probably because the relevant
>>> articles appeared many decades ago.
>>>
>>> Can anyone suggest some articles to read?
>>>
>>> Thanks,
>>>
>>> Joe Gwinn
>>>
>>>
>> Joe
>>
>> Although one could in principle do this with a single diode double
>> balanced mixer used as a phase detector all one may end up measuring is
>> the effect of ambient temperature changes on the mixer phase shift.
>> Lower mixer phase shift tempcos are possible if the RF port is
>> unsaturated.
>>
>
> Single diode? Why wouldn't one use a standard (MiniCircuits or the like)
> four-diode two-transformer double-balanced mixer as the phase detector?
> Many mixers have IF response down to DC.
>
>
Oops, I meant "single diode type double balanced mixer style phase
detector".
>
>> A classical dual mixer system is probably better in that with matched
>> tempco mixers maintained at the same temperature the differential phase
>> shift tempco should (with careful matching) be lower.
>>
>
> Dual mixer as in DMTD (dual mixer time difference) would certainly work,
> but is pretty complex and temperature sensitive.
>
> I did use a loaner Symmetricom 5120A (a full digital DMTD implementation)
> to make some measurements six months ago, and after a few days of
> continuous operation it had settled to the point that one could see 0.01
> pS changes. (And touching one of the BNC connectors caused a 1-3 pS
> jump.) This instrument costs about $30K, and is intended more for
> measuring phase noise and allan variance than delay changes.
>
> Anyway, I have to wonder what people did before DMTD was invented.
>
>
>
>> Other than the numerous classical papers on dual mixer systems and the
>> occasionl NIST paper that have some mixer phase shift tempco data
>> (albeit sparse), I am not aware of any specific papers.
>>
>
> I've read many or most of the classical DMTD papers, and have seen various
> passing estimates that diode-ring mixers have a temperature sensitivity of
> 8 to 10 pS per degree C. (I recall your figure was 10 pS/K.) I assume
> that the DC offset also varies with themerature and drive signal
> amplitude.
>
The only reference I have on the offset tempco is a miniciruits
application note from which one can deduce that the equivalent phase
shift tempco associated with the offset tempco is a few hundred
femtosec/C (@ 10MHz +7dBm) at some temperatures for the particular mixer
used. The graph also indicated (if you are lucky) that the offset tempco
may be zero at around 20C.
A NIST paper indicated that mixer phase shift tempco was around 10x
lower if the Rf port was unsaturated. It also indicated that the mixer
phase shift tempco is much lower if the input frequency is 100MHz rather
than 10MHz. This was one reason given for shifting to 100MHz DMTD systems.
>
>
>> A purely analog approach to phase shift measurement has to be more
>> difficult than a hybrid one using a pair of low frequency ADCs (eg high
>> end sound card).
>>
>
> Is the sound-card approach workable at the millidegree to microdegree
> level, if the change is spread out over an hour? One picosecond at 10 MHz
> is 3.6 millidegrees of phase.
>
> Joe
>
>
Preliminary (non optimum) tests by Ulrich indicate that picosecond
stability for times up to 100sec is very easy to achieve.
Beyond that mixer phase shift tempco mismatch may be significant.
ADEV noise level of around 2E-14/Tau (1s < tau <100s).
Haven't yet seen have data for longer tau.
With identical beat frequency outputs crosstalk between channels within
the sound card shouldn't be a great problem.
In any case its very easy to measure the crosstalk transfer function.
One concern particularly for low beat frequencies is the phase shift in
the sound card input coupling capacitors (usually electrolytics).
It should be easy to test the sound card phase shift stability for this
application by driving both inputs from the same signal source.
Bruce
JM
Joseph M Gwinn
Tue, Dec 9, 2008 9:56 PM
People used passive mixers driving electromechanical stripchart
to compare high-stability oscillators in the good old days.
One assumes that there is a purely analog approach to measurement of
picosecond changes in delay at 10 MHz using a single oscillator, but
have not seen any methods described, probably because the relevant
articles appeared many decades ago.
Can anyone suggest some articles to read?
Thanks,
Joe Gwinn
Joe
Although one could in principle do this with a single diode double
balanced mixer used as a phase detector all one may end up measuring
the effect of ambient temperature changes on the mixer phase shift.
Lower mixer phase shift tempcos are possible if the RF port is
unsaturated.
Single diode? Why wouldn't one use a standard (MiniCircuits or the
four-diode two-transformer double-balanced mixer as the phasedetector?
Many mixers have IF response down to DC.
Oops, I meant "single diode type double balanced mixer style phase
detector".
Ah. Four single diodes in a ratrace ring. Max drive +13 dBm or so.
Called Class I or Type I.
MiniCircuits ZRPD-1 being one example.
By the way, despite the circuit diagram in the datasheet, the
corresponding phase-detector module MPD-1 can be wired to have the IF
output ground isolated from the common RF, LO and case ground. A little
work with an ohmmeter will tell the tale. This can help to contain the
low frequency beatnote.
A classical dual mixer system is probably better in that with matched
tempco mixers maintained at the same temperature the differential
shift tempco should (with careful matching) be lower.
Dual mixer as in DMTD (dual mixer time difference) would certainly
but is pretty complex and temperature sensitive.
I did use a loaner Symmetricom 5120A (a full digital DMTD
to make some measurements six months ago, and after a few days of
continuous operation it had settled to the point that one could see
pS changes. (And touching one of the BNC connectors caused a 1-3 pS
jump.) This instrument costs about $30K, and is intended more for
measuring phase noise and allan variance than delay changes.
Anyway, I have to wonder what people did before DMTD was invented.
Other than the numerous classical papers on dual mixer systems and
occasionl NIST paper that have some mixer phase shift tempco data
(albeit sparse), I am not aware of any specific papers.
I've read many or most of the classical DMTD papers, and have seen
passing estimates that diode-ring mixers have a temperature
8 to 10 pS per degree C. (I recall your figure was 10 pS/K.) I
that the DC offset also varies with temperature and drive signal
amplitude.
The only reference I have on the offset tempco is a miniciruits
application note from which one can deduce that the equivalent phase
shift tempco associated with the offset tempco is a few hundred
femtosec/C (@ 10MHz +7dBm) at some temperatures for the particular mixer
used. The graph also indicated (if you are lucky) that the offset tempco
may be zero at around 20C.
Do you recall the part number?
A NIST paper indicated that mixer phase shift tempco was around 10x
lower if the Rf port was unsaturated. It also indicated that the mixer
phase shift tempco is much lower if the input frequency is 100MHz rather
than 10MHz. This was one reason given for shifting to 100MHz
DMTD systems.
Do you recall which paper?
What I've seen that seems useful is the Watkins-Johnson application note
from 1978 on use of mixers as phase detectors: "Mixers as Phase
Detectors", Stephan R. Kurtz, 8 pages. This may be the source of the NIST
article's information. The electrons are available on the web from WJ
Communications (now owned by TriQuint), filename "
http://www.wj.com/archive/documents/Tech_Notes_Archived/Mixers_phase_detectors.pdf
". Don't know how long this URL will work, as WJ is assimilated into
TriQuint.
A purely analog approach to phase shift measurement has to be more
difficult than a hybrid one using a pair of low frequency ADCs (eg
Is the sound-card approach workable at the millidegree to microdegree
level, if the change is spread out over an hour? One picosecond at 10
is 3.6 millidegrees of phase.
Joe
Preliminary (non optimum) tests by Ulrich indicate that picosecond
stability for times up to 100sec is very easy to achieve.
Beyond that mixer phase shift tempco mismatch may be significant.
It would not be that hard to make an oven for the mixer, as the level of
control needed is far less stringent than for a crystal.
ADEV noise level of around 2E-14/Tau (1s < tau <100s).
Haven't yet seen [or] have data for longer tau.
Yes. Need at least 10^4 seconds.
With identical beat frequency outputs, crosstalk between channels within
the sound card shouldn't be a great problem.
I'm not sure I believe this, as there is likely ground coupling within the
soundcard and the ear is famously insensitive to phase. Channel isolation
of 60 dB isn't enough to prevent phase shifts.
In any case it's very easy to measure the crosstalk transfer function.
One concern particularly for low beat frequencies is the phase shift in
the sound card input coupling capacitors (usually electrolytics).
It should be easy to test the sound card phase shift stability for this
application by driving both inputs from the same signal source.
I assume that the beatnote must be ~100 Hz for the soundcard to handle
with low phase shift. One might get to 10 Hz, but 1 Hz is likely
hopeless.
One thing that will be very useful is a list of sound cards by make and
model, annotated with their advantages and disadvantages for time-nut use.
"High-end" may not be a sufficient description.
By the way, I looked at the operating and service manual for the HP
K34-59991A Broadband Linear Phase Comparator. Very interesting little
gadget, but little performance data was given. Does anyone know how well
phase change can be measured? It would be easy to duplicate this with
modern ICs. Also, about when was this unit made? The manual has no date.
Joe
Bruce
time-nuts-bounces@febo.com wrote on 12/08/2008 07:12:22 PM:
> Joseph M Gwinn wrote:
> > Bruce,
> >
> >
> > time-nuts-bounces@febo.com wrote on 12/08/2008 05:53:08 PM:
> >
> >
> >> Joseph M Gwinn wrote:
> >>
> >>> People used passive mixers driving electromechanical stripchart
recorders
> >>> to compare high-stability oscillators in the good old days.
> >>>
> >>> One assumes that there is a purely analog approach to measurement of
> >>> picosecond changes in delay at 10 MHz using a single oscillator, but
I
> >>> have not seen any methods described, probably because the relevant
> >>> articles appeared many decades ago.
> >>>
> >>> Can anyone suggest some articles to read?
> >>>
> >>> Thanks,
> >>>
> >>> Joe Gwinn
> >>>
> >>>
> >> Joe
> >>
> >> Although one could in principle do this with a single diode double
> >> balanced mixer used as a phase detector all one may end up measuring
is
> >> the effect of ambient temperature changes on the mixer phase shift.
> >> Lower mixer phase shift tempcos are possible if the RF port is
> >> unsaturated.
> >>
> >
> > Single diode? Why wouldn't one use a standard (MiniCircuits or the
like)
> > four-diode two-transformer double-balanced mixer as the phasedetector?
> > Many mixers have IF response down to DC.
> >
> Oops, I meant "single diode type double balanced mixer style phase
> detector".
Ah. Four single diodes in a ratrace ring. Max drive +13 dBm or so.
Called Class I or Type I.
MiniCircuits ZRPD-1 being one example.
By the way, despite the circuit diagram in the datasheet, the
corresponding phase-detector module MPD-1 can be wired to have the IF
output ground isolated from the common RF, LO and case ground. A little
work with an ohmmeter will tell the tale. This can help to contain the
low frequency beatnote.
> >> A classical dual mixer system is probably better in that with matched
> >> tempco mixers maintained at the same temperature the differential
phase
> >> shift tempco should (with careful matching) be lower.
> >
> > Dual mixer as in DMTD (dual mixer time difference) would certainly
work,
> > but is pretty complex and temperature sensitive.
> >
> > I did use a loaner Symmetricom 5120A (a full digital DMTD
implementation)
> > to make some measurements six months ago, and after a few days of
> > continuous operation it had settled to the point that one could see
0.01
> > pS changes. (And touching one of the BNC connectors caused a 1-3 pS
> > jump.) This instrument costs about $30K, and is intended more for
> > measuring phase noise and allan variance than delay changes.
> >
> > Anyway, I have to wonder what people did before DMTD was invented.
> >
> >
> >
> >> Other than the numerous classical papers on dual mixer systems and
the
> >> occasionl NIST paper that have some mixer phase shift tempco data
> >> (albeit sparse), I am not aware of any specific papers.
> >
> > I've read many or most of the classical DMTD papers, and have seen
various
> > passing estimates that diode-ring mixers have a temperature
sensitivity of
> > 8 to 10 pS per degree C. (I recall your figure was 10 pS/K.) I
assume
> > that the DC offset also varies with temperature and drive signal
> > amplitude.
> >
> The only reference I have on the offset tempco is a miniciruits
> application note from which one can deduce that the equivalent phase
> shift tempco associated with the offset tempco is a few hundred
> femtosec/C (@ 10MHz +7dBm) at some temperatures for the particular mixer
> used. The graph also indicated (if you are lucky) that the offset tempco
> may be zero at around 20C.
Do you recall the part number?
> A NIST paper indicated that mixer phase shift tempco was around 10x
> lower if the Rf port was unsaturated. It also indicated that the mixer
> phase shift tempco is much lower if the input frequency is 100MHz rather
> than 10MHz. This was one reason given for shifting to 100MHz
> DMTD systems.
Do you recall which paper?
What I've seen that seems useful is the Watkins-Johnson application note
from 1978 on use of mixers as phase detectors: "Mixers as Phase
Detectors", Stephan R. Kurtz, 8 pages. This may be the source of the NIST
article's information. The electrons are available on the web from WJ
Communications (now owned by TriQuint), filename "
http://www.wj.com/archive/documents/Tech_Notes_Archived/Mixers_phase_detectors.pdf
". Don't know how long this URL will work, as WJ is assimilated into
TriQuint.
> >> A purely analog approach to phase shift measurement has to be more
> >> difficult than a hybrid one using a pair of low frequency ADCs (eg
high
> >> end sound card).
> >>
> >
> > Is the sound-card approach workable at the millidegree to microdegree
> > level, if the change is spread out over an hour? One picosecond at 10
MHz
> > is 3.6 millidegrees of phase.
> >
> > Joe
> >
> >
> Preliminary (non optimum) tests by Ulrich indicate that picosecond
> stability for times up to 100sec is very easy to achieve.
> Beyond that mixer phase shift tempco mismatch may be significant.
It would not be that hard to make an oven for the mixer, as the level of
control needed is far less stringent than for a crystal.
> ADEV noise level of around 2E-14/Tau (1s < tau <100s).
> Haven't yet seen [or] have data for longer tau.
Yes. Need at least 10^4 seconds.
> With identical beat frequency outputs, crosstalk between channels within
> the sound card shouldn't be a great problem.
I'm not sure I believe this, as there is likely ground coupling within the
soundcard and the ear is famously insensitive to phase. Channel isolation
of 60 dB isn't enough to prevent phase shifts.
> In any case it's very easy to measure the crosstalk transfer function.
Yes.
> One concern particularly for low beat frequencies is the phase shift in
> the sound card input coupling capacitors (usually electrolytics).
>
> It should be easy to test the sound card phase shift stability for this
> application by driving both inputs from the same signal source.
I assume that the beatnote must be ~100 Hz for the soundcard to handle
with low phase shift. One might get to 10 Hz, but 1 Hz is likely
hopeless.
One thing that will be very useful is a list of sound cards by make and
model, annotated with their advantages and disadvantages for time-nut use.
"High-end" may not be a sufficient description.
By the way, I looked at the operating and service manual for the HP
K34-59991A Broadband Linear Phase Comparator. Very interesting little
gadget, but little performance data was given. Does anyone know how well
phase change can be measured? It would be easy to duplicate this with
modern ICs. Also, about when was this unit made? The manual has no date.
Joe
BG
Bruce Griffiths
Tue, Dec 9, 2008 11:24 PM
People used passive mixers driving electromechanical stripchart
to compare high-stability oscillators in the good old days.
One assumes that there is a purely analog approach to measurement of
picosecond changes in delay at 10 MHz using a single oscillator, but
have not seen any methods described, probably because the relevant
articles appeared many decades ago.
Can anyone suggest some articles to read?
Thanks,
Joe Gwinn
Joe
Although one could in principle do this with a single diode double
balanced mixer used as a phase detector all one may end up measuring
the effect of ambient temperature changes on the mixer phase shift.
Lower mixer phase shift tempcos are possible if the RF port is
unsaturated.
Single diode? Why wouldn't one use a standard (MiniCircuits or the
four-diode two-transformer double-balanced mixer as the phasedetector?
Many mixers have IF response down to DC.
Oops, I meant "single diode type double balanced mixer style phase
detector".
Ah. Four single diodes in a ratrace ring. Max drive +13 dBm or so.
Called Class I or Type I.
MiniCircuits ZRPD-1 being one example.
By the way, despite the circuit diagram in the datasheet, the
corresponding phase-detector module MPD-1 can be wired to have the IF
output ground isolated from the common RF, LO and case ground. A little
work with an ohmmeter will tell the tale. This can help to contain the
low frequency beatnote.
Yes, thats usually the case for the Minicircuits PCB mount phase
detectors and mixers except for some surface mount versions (usually the
very high frequency models).
A PCB mount mixer package is also preferable as its then much easier to
use a capacitive IF port termination (for lower noise) in conjunction
with series resistors at the RF and LO ports (for lower VSWR) than if a
mixer with SMA or other coax connectors were used.
A classical dual mixer system is probably better in that with matched
tempco mixers maintained at the same temperature the differential
shift tempco should (with careful matching) be lower.
Dual mixer as in DMTD (dual mixer time difference) would certainly
but is pretty complex and temperature sensitive.
I did use a loaner Symmetricom 5120A (a full digital DMTD
to make some measurements six months ago, and after a few days of
continuous operation it had settled to the point that one could see
pS changes. (And touching one of the BNC connectors caused a 1-3 pS
jump.) This instrument costs about $30K, and is intended more for
measuring phase noise and allan variance than delay changes.
Anyway, I have to wonder what people did before DMTD was invented.
Other than the numerous classical papers on dual mixer systems and
occasionl NIST paper that have some mixer phase shift tempco data
(albeit sparse), I am not aware of any specific papers.
I've read many or most of the classical DMTD papers, and have seen
passing estimates that diode-ring mixers have a temperature
8 to 10 pS per degree C. (I recall your figure was 10 pS/K.) I
that the DC offset also varies with temperature and drive signal
amplitude.
The only reference I have on the offset tempco is a miniciruits
application note from which one can deduce that the equivalent phase
shift tempco associated with the offset tempco is a few hundred
femtosec/C (@ 10MHz +7dBm) at some temperatures for the particular mixer
used. The graph also indicated (if you are lucky) that the offset tempco
may be zero at around 20C.
Do you recall the part number?
Supposedly an SRA-1, but some caution is in order as some statements as
to the effect of the input offset of an opamp based IF preamp in the
same application note were of dubious veracity unless one were to use an
inverting opamp input stage.
A NIST paper indicated that mixer phase shift tempco was around 10x
lower if the Rf port was unsaturated. It also indicated that the mixer
phase shift tempco is much lower if the input frequency is 100MHz rather
than 10MHz. This was one reason given for shifting to 100MHz
DMTD systems.
Do you recall which paper?
What I've seen that seems useful is the Watkins-Johnson application note
from 1978 on use of mixers as phase detectors: "Mixers as Phase
Detectors", Stephan R. Kurtz, 8 pages. This may be the source of the NIST
article's information. The electrons are available on the web from WJ
Communications (now owned by TriQuint), filename "
http://www.wj.com/archive/documents/Tech_Notes_Archived/Mixers_phase_detectors.pdf
". Don't know how long this URL will work, as WJ is assimilated into
TriQuint.
A purely analog approach to phase shift measurement has to be more
difficult than a hybrid one using a pair of low frequency ADCs (eg
Is the sound-card approach workable at the millidegree to microdegree
level, if the change is spread out over an hour? One picosecond at 10
is 3.6 millidegrees of phase.
Joe
Preliminary (non optimum) tests by Ulrich indicate that picosecond
stability for times up to 100sec is very easy to achieve.
Beyond that mixer phase shift tempco mismatch may be significant.
It would not be that hard to make an oven for the mixer, as the level of
control needed is far less stringent than for a crystal.
ADEV noise level of around 2E-14/Tau (1s < tau <100s).
Haven't yet seen [or] have data for longer tau.
Yes. Need at least 10^4 seconds.
With identical beat frequency outputs, crosstalk between channels within
the sound card shouldn't be a great problem.
I'm not sure I believe this, as there is likely ground coupling within the
soundcard and the ear is famously insensitive to phase. Channel isolation
of 60 dB isn't enough to prevent phase shifts.
It will be present but its effect in some cases (when the phase shift
between channels is such that the crosstalk phase is at 90 degrees to
the signal of interest) will be negligible, in other cases it is easily
measured and compensated for.
In any case it's very easy to measure the crosstalk transfer function.
One concern particularly for low beat frequencies is the phase shift in
the sound card input coupling capacitors (usually electrolytics).
It should be easy to test the sound card phase shift stability for this
application by driving both inputs from the same signal source.
I assume that the beatnote must be ~100 Hz for the soundcard to handle
with low phase shift. One might get to 10 Hz, but 1 Hz is likely
hopeless.
One thing that will be very useful is a list of sound cards by make and
model, annotated with their advantages and disadvantages for time-nut use.
"High-end" may not be a sufficient description.
By the way, I looked at the operating and service manual for the HP
K34-59991A Broadband Linear Phase Comparator. Very interesting little
gadget, but little performance data was given. Does anyone know how well
phase change can be measured? It would be easy to duplicate this with
modern ICs. Also, about when was this unit made? The manual has no date.
Joe
Joe
I suspect that slow phase changes much less than 1ns or so are hard to
distinguish from gain drift given the gain tempco of the ECL phase detector.
A beat note near 1kHz appears to be even better if one is using
something like an enhanced Costas receiver or even using WKS
interpolation to locate and time stamp zero crossings.
So far only the M-Audio AP192 has been used.
Tests with an embedded motherboard 16 bit sound system show
significantly increased noise.
I've found that the noise level of motherboard sound systems varies
enormously from one motherboard model (sample of 2) to another.
Any 24 bit sound card with a performance close to or better than that of
the AP192 should suffice.
Other cards using AKM 24 bit ADCs should also be suitable.
Ideally an external sound card with balanced XLR inputs would be best.
HP produced a number of different phase comparators each with a
different type of phase detector.
The K34-5991A design can't be older than the early 1970's because the
MECLIII devices used weren't available until then.
Warren built a similar phase detector (differential XOR or XOR + XNOR)
using CMOS ICs and for a common 10MHz input with a phase difference near
zero found short term output noise of of around 10uV or so (10V phase
detector FSR) using a passive low pass filter.
In principle an ADC like the LTC2484 could be used with a 2.5V CMOS
OR/XNOR phase detector and passive low pass filters.
The ratiometric conversion capability will significantly reduce the
sensitivity to the XOR gate supply if the XOR gate supply is also used
for the ADC reference voltage.
If one used 5V logic a resistive output attenuator would be needed which
reduce the gain stability somewhat.
All such phase detectors suffer from substantial nonlinearity near the
ends of the range due to gate output slew rate limitations.
However if operated near the centre of the range subpicosecond
sensitivity/short term stability may be possible.
Since the circuit volume is small and the tempco relatively low (few
ps/C at most) regulating the circuit temperature should be relatively
easy to do.
Bruce
Joseph M Gwinn wrote:
> Bruce
>
>
> time-nuts-bounces@febo.com wrote on 12/08/2008 07:12:22 PM:
>
>
>> Joseph M Gwinn wrote:
>>
>>> Bruce,
>>>
>>>
>>> time-nuts-bounces@febo.com wrote on 12/08/2008 05:53:08 PM:
>>>
>>>
>>>
>>>> Joseph M Gwinn wrote:
>>>>
>>>>
>>>>> People used passive mixers driving electromechanical stripchart
>>>>>
> recorders
>
>>>>> to compare high-stability oscillators in the good old days.
>>>>>
>>>>> One assumes that there is a purely analog approach to measurement of
>>>>>
>
>
>>>>> picosecond changes in delay at 10 MHz using a single oscillator, but
>>>>>
> I
>
>>>>> have not seen any methods described, probably because the relevant
>>>>> articles appeared many decades ago.
>>>>>
>>>>> Can anyone suggest some articles to read?
>>>>>
>>>>> Thanks,
>>>>>
>>>>> Joe Gwinn
>>>>>
>>>>>
>>>>>
>>>> Joe
>>>>
>>>> Although one could in principle do this with a single diode double
>>>> balanced mixer used as a phase detector all one may end up measuring
>>>>
> is
>
>>>> the effect of ambient temperature changes on the mixer phase shift.
>>>> Lower mixer phase shift tempcos are possible if the RF port is
>>>> unsaturated.
>>>>
>>>>
>>> Single diode? Why wouldn't one use a standard (MiniCircuits or the
>>>
> like)
>
>>> four-diode two-transformer double-balanced mixer as the phasedetector?
>>>
>
>
>>> Many mixers have IF response down to DC.
>>>
>>>
>> Oops, I meant "single diode type double balanced mixer style phase
>> detector".
>>
>
> Ah. Four single diodes in a ratrace ring. Max drive +13 dBm or so.
> Called Class I or Type I.
>
> MiniCircuits ZRPD-1 being one example.
>
> By the way, despite the circuit diagram in the datasheet, the
> corresponding phase-detector module MPD-1 can be wired to have the IF
> output ground isolated from the common RF, LO and case ground. A little
> work with an ohmmeter will tell the tale. This can help to contain the
> low frequency beatnote.
>
>
Yes, thats usually the case for the Minicircuits PCB mount phase
detectors and mixers except for some surface mount versions (usually the
very high frequency models).
A PCB mount mixer package is also preferable as its then much easier to
use a capacitive IF port termination (for lower noise) in conjunction
with series resistors at the RF and LO ports (for lower VSWR) than if a
mixer with SMA or other coax connectors were used.
>
>
>>>> A classical dual mixer system is probably better in that with matched
>>>> tempco mixers maintained at the same temperature the differential
>>>>
> phase
>
>>>> shift tempco should (with careful matching) be lower.
>>>>
>>> Dual mixer as in DMTD (dual mixer time difference) would certainly
>>>
> work,
>
>>> but is pretty complex and temperature sensitive.
>>>
>>> I did use a loaner Symmetricom 5120A (a full digital DMTD
>>>
> implementation)
>
>>> to make some measurements six months ago, and after a few days of
>>> continuous operation it had settled to the point that one could see
>>>
> 0.01
>
>>> pS changes. (And touching one of the BNC connectors caused a 1-3 pS
>>> jump.) This instrument costs about $30K, and is intended more for
>>> measuring phase noise and allan variance than delay changes.
>>>
>>> Anyway, I have to wonder what people did before DMTD was invented.
>>>
>>>
>>>
>>>
>>>> Other than the numerous classical papers on dual mixer systems and
>>>>
> the
>
>>>> occasionl NIST paper that have some mixer phase shift tempco data
>>>> (albeit sparse), I am not aware of any specific papers.
>>>>
>>> I've read many or most of the classical DMTD papers, and have seen
>>>
> various
>
>>> passing estimates that diode-ring mixers have a temperature
>>>
> sensitivity of
>
>>> 8 to 10 pS per degree C. (I recall your figure was 10 pS/K.) I
>>>
> assume
>
>>> that the DC offset also varies with temperature and drive signal
>>> amplitude.
>>>
>>>
>> The only reference I have on the offset tempco is a miniciruits
>> application note from which one can deduce that the equivalent phase
>> shift tempco associated with the offset tempco is a few hundred
>> femtosec/C (@ 10MHz +7dBm) at some temperatures for the particular mixer
>> used. The graph also indicated (if you are lucky) that the offset tempco
>> may be zero at around 20C.
>>
>
> Do you recall the part number?
>
>
>
Supposedly an SRA-1, but some caution is in order as some statements as
to the effect of the input offset of an opamp based IF preamp in the
same application note were of dubious veracity unless one were to use an
inverting opamp input stage.
>> A NIST paper indicated that mixer phase shift tempco was around 10x
>> lower if the Rf port was unsaturated. It also indicated that the mixer
>> phase shift tempco is much lower if the input frequency is 100MHz rather
>> than 10MHz. This was one reason given for shifting to 100MHz
>> DMTD systems.
>>
>
> Do you recall which paper?
>
>
>
http://tf.nist.gov/timefreq/general/pdf/971.pdf
<http://tf.nist.gov/timefreq/general/pdf/971.pdf>
Has some measurement data on mixer phase shift tempco and power
sensitivity and their frequency dependence etc.
I'll search for the paper that stated that the phase shift tempco was
lower if the RF port was unsaturated.
AFAIK there was no accompanying measurement data
> What I've seen that seems useful is the Watkins-Johnson application note
> from 1978 on use of mixers as phase detectors: "Mixers as Phase
> Detectors", Stephan R. Kurtz, 8 pages. This may be the source of the NIST
> article's information. The electrons are available on the web from WJ
> Communications (now owned by TriQuint), filename "
> http://www.wj.com/archive/documents/Tech_Notes_Archived/Mixers_phase_detectors.pdf
> ". Don't know how long this URL will work, as WJ is assimilated into
> TriQuint.
>
>
>
>>>> A purely analog approach to phase shift measurement has to be more
>>>> difficult than a hybrid one using a pair of low frequency ADCs (eg
>>>>
> high
>
>>>> end sound card).
>>>>
>>>>
>>> Is the sound-card approach workable at the millidegree to microdegree
>>> level, if the change is spread out over an hour? One picosecond at 10
>>>
> MHz
>
>>> is 3.6 millidegrees of phase.
>>>
>>> Joe
>>>
>>>
>>>
>> Preliminary (non optimum) tests by Ulrich indicate that picosecond
>> stability for times up to 100sec is very easy to achieve.
>> Beyond that mixer phase shift tempco mismatch may be significant.
>>
>
> It would not be that hard to make an oven for the mixer, as the level of
> control needed is far less stringent than for a crystal.
>
>
>
>> ADEV noise level of around 2E-14/Tau (1s < tau <100s).
>> Haven't yet seen [or] have data for longer tau.
>>
>
> Yes. Need at least 10^4 seconds.
>
>
>
>> With identical beat frequency outputs, crosstalk between channels within
>> the sound card shouldn't be a great problem.
>>
>
> I'm not sure I believe this, as there is likely ground coupling within the
> soundcard and the ear is famously insensitive to phase. Channel isolation
> of 60 dB isn't enough to prevent phase shifts.
>
>
It will be present but its effect in some cases (when the phase shift
between channels is such that the crosstalk phase is at 90 degrees to
the signal of interest) will be negligible, in other cases it is easily
measured and compensated for.
>
>> In any case it's very easy to measure the crosstalk transfer function.
>>
>
> Yes.
>
>
>
>> One concern particularly for low beat frequencies is the phase shift in
>> the sound card input coupling capacitors (usually electrolytics).
>>
>> It should be easy to test the sound card phase shift stability for this
>> application by driving both inputs from the same signal source.
>>
>
> I assume that the beatnote must be ~100 Hz for the soundcard to handle
> with low phase shift. One might get to 10 Hz, but 1 Hz is likely
> hopeless.
>
> One thing that will be very useful is a list of sound cards by make and
> model, annotated with their advantages and disadvantages for time-nut use.
>
> "High-end" may not be a sufficient description.
>
>
> By the way, I looked at the operating and service manual for the HP
> K34-59991A Broadband Linear Phase Comparator. Very interesting little
> gadget, but little performance data was given. Does anyone know how well
> phase change can be measured? It would be easy to duplicate this with
> modern ICs. Also, about when was this unit made? The manual has no date.
>
>
> Joe
>
>
Joe
I suspect that slow phase changes much less than 1ns or so are hard to
distinguish from gain drift given the gain tempco of the ECL phase detector.
A beat note near 1kHz appears to be even better if one is using
something like an enhanced Costas receiver or even using WKS
interpolation to locate and time stamp zero crossings.
So far only the M-Audio AP192 has been used.
Tests with an embedded motherboard 16 bit sound system show
significantly increased noise.
I've found that the noise level of motherboard sound systems varies
enormously from one motherboard model (sample of 2) to another.
Any 24 bit sound card with a performance close to or better than that of
the AP192 should suffice.
Other cards using AKM 24 bit ADCs should also be suitable.
Ideally an external sound card with balanced XLR inputs would be best.
HP produced a number of different phase comparators each with a
different type of phase detector.
The K34-5991A design can't be older than the early 1970's because the
MECLIII devices used weren't available until then.
Warren built a similar phase detector (differential XOR or XOR + XNOR)
using CMOS ICs and for a common 10MHz input with a phase difference near
zero found short term output noise of of around 10uV or so (10V phase
detector FSR) using a passive low pass filter.
In principle an ADC like the LTC2484 could be used with a 2.5V CMOS
OR/XNOR phase detector and passive low pass filters.
The ratiometric conversion capability will significantly reduce the
sensitivity to the XOR gate supply if the XOR gate supply is also used
for the ADC reference voltage.
If one used 5V logic a resistive output attenuator would be needed which
reduce the gain stability somewhat.
All such phase detectors suffer from substantial nonlinearity near the
ends of the range due to gate output slew rate limitations.
However if operated near the centre of the range subpicosecond
sensitivity/short term stability may be possible.
Since the circuit volume is small and the tempco relatively low (few
ps/C at most) regulating the circuit temperature should be relatively
easy to do.
Bruce
JM
Joseph M Gwinn
Wed, Dec 10, 2008 12:34 AM
This one is new to me, and interesting. More later, when I've read it.
I had found this one, and is why I used a 3 dB attenuator on the LO port
and a 8 dB attenuator on the RF port. But the pieces are coming together.
Thanks,
Joe
Bruce,
time-nuts-bounces@febo.com wrote on 12/09/2008 07:10:55 PM:
> Paper with capacitive IF port termination data:
>
> http://tf.nist.gov/timefreq/general/pdf/112.pdf
This one is new to me, and interesting. More later, when I've read it.
> Phase detector sensitivity to distortion:
>
> http://tf.nist.gov/timefreq/general/pdf/1437.pdf
I had found this one, and is why I used a 3 dB attenuator on the LO port
and a 8 dB attenuator on the RF port. But the pieces are coming together.
Thanks,
Joe
BG
Bruce Griffiths
Wed, Dec 10, 2008 12:55 AM
This one is new to me, and interesting. More later, when I've read it.
Joseph M Gwinn wrote:
> Bruce,
>
>
> time-nuts-bounces@febo.com wrote on 12/09/2008 07:10:55 PM:
>
>
>> Paper with capacitive IF port termination data:
>>
>> http://tf.nist.gov/timefreq/general/pdf/112.pdf
>>
>
> This one is new to me, and interesting. More later, when I've read it.
>
>
>
>> Phase detector sensitivity to distortion:
>>
>> http://tf.nist.gov/timefreq/general/pdf/1437.pdf
>>
>
> I had found this one, and is why I used a 3 dB attenuator on the LO port
> and a 8 dB attenuator on the RF port. But the pieces are coming together.
>
>
> Thanks,
>
> Joe
>
> _______________________________________________
> time-nuts mailing list -- time-nuts@febo.com
> To unsubscribe, go to https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
> and follow the instructions there.
>
>
Joe
Did you also look at?:
http://tf.nist.gov/timefreq/general/pdf/971.pdf
Which indicates how one can tune the phase shift tempco to some extent.
Bruce
JM
Joseph M Gwinn
Wed, Dec 10, 2008 1:18 AM
Many mixers have IF response down to DC.
Oops, I meant "single diode type double balanced mixer style phase
detector".
Ah. Four single diodes in a ratrace ring. Max drive +13 dBm or so.
Called Class I or Type I.
MiniCircuits ZRPD-1 being one example.
By the way, despite the circuit diagram in the datasheet, the
corresponding phase-detector module MPD-1 can be wired to have the IF
output ground isolated from the common RF, LO and case ground. A
little work with an ohmmeter will tell the tale. This can help to
Yes, that's usually the case for the Minicircuits PCB mount phase
detectors and mixers except for some surface mount versions (usually the
very high frequency models).
So it was already known. It looks to me that MiniCircuit's intent is to
support automated testing of modules.
A PCB mount mixer package is also preferable as its then much easier to
use a capacitive IF port termination (for lower noise) in conjunction
with series resistors at the RF and LO ports (for lower VSWR) than if a
mixer with SMA or other coax connectors were used.
I've been using 3 and 8 dB coaxial attenuators at the LO and RF inputs
respectively, and it makes a big difference.
But I don't understand the part about capacitive loading of the IF port. I
would think that the low pass filter would need to present a matched
impedance at the sum frequency, so the emerging high-level 20 MHz signal
is not reflected back into the mixer.
MiniCircuits AN-41-001 "FAQ about Phase Detectors" has on page 2 a 500 ohm
resistor to ground and a 5000 ohm resistor to the first filter capacitor,
so the capacitor is isolated from the IF port by the resistors.
I just got your posting about paper 112, so more later.
[snip]
I've read many or most of the classical DMTD papers, and have seen
various passing estimates that diode-ring mixers have a temperature
sensitivity of 8 to 10 pS per degree C. (I recall your figure was
I assume that the DC offset also varies with temperature and drive
The only reference I have on the offset tempco is a miniciruits
application note from which one can deduce that the equivalent phase
shift tempco associated with the offset tempco is a few hundred
femtosec/C (@ 10MHz +7dBm) at some temperatures for the particular
used. The graph also indicated (if you are lucky) that the offset
may be zero at around 20C.
Do you recall the part number?
Supposedly an SRA-1, but some caution is in order as some statements as
to the effect of the input offset of an opamp based IF preamp in the
same application note were of dubious veracity unless one were to use an
inverting opamp input stage.
This issue was mentioned in another app note, but their main issue
appeared to be that the opamp bias currents could cause an offset.
A NIST paper indicated that mixer phase shift tempco was around 10x
lower if the RF port was unsaturated. It also indicated that the
phase shift tempco is much lower if the input frequency is 100MHz
than 10MHz. This was one reason given for shifting to 100MHz
DMTD systems.
Do you recall which paper?
I do know this paper. At the bottom of page 834, to the right, is the
estimate 3.5 pS/K.
Another reason to go to 100 MHz is that the temperature coefficient of
electrical length of polymer-insulated coax is far lower at 100 MHz
compared to 10 MHz.
I'll search for the paper that stated that the phase shift tempco was
lower if the RF port was unsaturated.
I think I have seen this too, but don't recall where. But it's why I use
an 8 dB pad on the RF input.
AFAIK there was no accompanying measurement data
What I've seen that seems useful is the Watkins-Johnson application
from 1978 on use of mixers as phase detectors: "Mixers as Phase
Detectors", Stephan R. Kurtz, 8 pages. This may be the source of the
article's information. The electrons are available on the web from WJ
Communications (now owned by TriQuint), filename
Don't know how long this URL will work, as WJ is assimilated into
TriQuint.
This app note is reference 7 of paper 971 above.
[Soundcards]
Preliminary (non optimum) tests by Ulrich indicate that picosecond
stability for times up to 100sec is very easy to achieve.
Beyond that mixer phase shift tempco mismatch may be significant.
It would not be that hard to make an oven for the mixer, as
control needed is far less stringent than for a crystal.
ADEV noise level of around 2E-14/Tau (1s < tau <100s).
Haven't yet seen [or] have data for longer tau.
Yes. Need at least 10^4 seconds.
With identical beat frequency outputs, crosstalk between
channels within the sound card shouldn't be a great problem.
I'm not sure I believe this, as there is likely ground coupling within
soundcard and the ear is famously insensitive to phase. Channel
of 60 dB isn't enough to prevent phase shifts.
It will be present but its effect in some cases (when the phase shift
between channels is such that the crosstalk phase is at 90 degrees to
the signal of interest) will be negligible, in other cases it is easily
measured and compensated for.
Isn't 90 degrees (quadrature) the worst case for causing phase shifts?
To get a one picosecond change at 10 MHz by injection of an attenuated
quadrature copy of the main signal requires a relative voltage ratio of
Tan[(10^-12)(10^7)(360)] = Tan[0.0036] = 0.0000628 of the main signal, or
20 Log[0.0000628]= -84 dBc. This is well exceeds the interchannel
isolation of many sound cards.
Cancellation by mathematical means could be possible, but will require a
dynamic range well exceeding 84 dB. This ought to be easy to arrange.
One concern particularly for low beat frequencies is the phase shift
the sound card input coupling capacitors (usually electrolytics).
It should be easy to test the sound card phase shift stability for
application by driving both inputs from the same signal source.
Or terminate one channel input and drive the other, and measure the
amplitude and phase of whatever comes out of the terminated channel,
compared to the driven channel. Then swap channels and repeat. The phase
and amplitude will depend on frequency, so a sweep will be required, and
some frequencies may need to be avoided.
I assume that the beatnote must be ~100 Hz for the soundcard to handle
with low phase shift. One might get to 10 Hz, but 1 Hz is likely
hopeless.
One thing that will be very useful is a list of sound cards by make
model, annotated with their advantages and disadvantages for time-nut
"High-end" may not be a sufficient description.
By the way, I looked at the operating and service manual for the HP
K34-59991A Broadband Linear Phase Comparator. Very interesting little
gadget, but little performance data was given. Does anyone know how
phase change can be measured? It would be easy to duplicate this with
modern ICs. Also, about when was this unit made? The manual has no
Joe
I suspect that slow phase changes much less than 1ns or so are hard to
distinguish from gain drift given the gain tempco of the ECL
phase detector.
A beat note near 1kHz appears to be even better if one is using
something like an enhanced Costas receiver or even using WKS
interpolation to locate and time stamp zero crossings.
But it limits the phase slope gain. I suppose there is an optimum
somewhere.
So far only the M-Audio AP192 has been used.
Tests with an embedded motherboard 16 bit sound system show
significantly increased noise.
I've found that the noise level of motherboard sound systems varies
enormously from one motherboard model (sample of 2) to another.
Any 24 bit sound card with a performance close to or better than that of
the AP192 should suffice.
In general, firewire connected sound cards should be better, because the
soundcard maker has complete control of what's inside the box. Unlike
inside a PC.
Other cards using AKM 24 bit ADCs should also be suitable.
Who is AKM?
20 Log[ 2^24 ] = 144 dB, so something else will be the limit.
Ideally an external sound card with balanced XLR inputs would be best.
HP produced a number of different phase comparators each with a
different type of phase detector.
The K34-5991A design can't be older than the early 1970's because the
MECLIII devices used weren't available until then.
OK. I recall MECL. RIP. But we have PECL now.
Warren built a similar phase detector (differential XOR or XOR + XNOR)
using CMOS ICs and for a common 10MHz input with a phase difference near
zero found short term output noise of of around 10uV or so (10V phase
detector FSR) using a passive low pass filter.
(10uV)/(10v)= 1 ppm. (100 nS)(10^-6)= 0.1 pS.
In principle an ADC like the LTC2484 could be used with a 2.5V CMOS
OR/XNOR phase detector and passive low pass filters.
The ratiometric conversion capability will significantly reduce the
sensitivity to the XOR gate supply if the XOR gate supply is also used
for the ADC reference voltage.
If one used 5V logic a resistive output attenuator would be needed which
reduce the gain stability somewhat.
All such phase detectors suffer from substantial nonlinearity near the
ends of the range due to gate output slew rate limitations.
If one is tracking through multiple phase cycles (as did the HP unit),
this would matter.
As for slew rate, one can buy faster logic, but probably at the expense of
jitter (as bandwidth must be larger).
The HP unit used balanced logic all the way up to the flipflops. This
doubles the relative crossing rate for a given slew rate. But doubles the
noise, but it must have been worth it, or HP wouldn't have done that.
In beatnote zero crossing detectors, differential signals help keep ground
loops and bounce down.
However if operated near the centre of the range subpicosecond
sensitivity/short term stability may be possible.
Since the circuit volume is small and the tempco relatively low (few
ps/C at most) regulating the circuit temperature should be relatively
easy to do.
We ought to be able to achieve less than 0.1 K change.
Joe
Bruce,
time-nuts-bounces@febo.com wrote on 12/09/2008 06:24:22 PM:
> Joseph M Gwinn wrote:
> > Bruce
> >
> >
> > time-nuts-bounces@febo.com wrote on 12/08/2008 07:12:22 PM:
> >
> >
[snip]
> >
> >
> >>> Many mixers have IF response down to DC.
> >>>
> >>>
> >> Oops, I meant "single diode type double balanced mixer style phase
> >> detector".
> >>
> >
> > Ah. Four single diodes in a ratrace ring. Max drive +13 dBm or so.
> > Called Class I or Type I.
> >
> > MiniCircuits ZRPD-1 being one example.
> >
> > By the way, despite the circuit diagram in the datasheet, the
> > corresponding phase-detector module MPD-1 can be wired to have the IF
> > output ground isolated from the common RF, LO and case ground. A
> > little work with an ohmmeter will tell the tale. This can help to
contain the
> > low frequency beatnote.
>
> Yes, that's usually the case for the Minicircuits PCB mount phase
> detectors and mixers except for some surface mount versions (usually the
> very high frequency models).
So it was already known. It looks to me that MiniCircuit's intent is to
support automated testing of modules.
> A PCB mount mixer package is also preferable as its then much easier to
> use a capacitive IF port termination (for lower noise) in conjunction
> with series resistors at the RF and LO ports (for lower VSWR) than if a
> mixer with SMA or other coax connectors were used.
I've been using 3 and 8 dB coaxial attenuators at the LO and RF inputs
respectively, and it makes a big difference.
But I don't understand the part about capacitive loading of the IF port. I
would think that the low pass filter would need to present a matched
impedance at the sum frequency, so the emerging high-level 20 MHz signal
is not reflected back into the mixer.
MiniCircuits AN-41-001 "FAQ about Phase Detectors" has on page 2 a 500 ohm
resistor to ground and a 5000 ohm resistor to the first filter capacitor,
so the capacitor is isolated from the IF port by the resistors.
I just got your posting about paper 112, so more later.
[snip]
> >>>>
> >>> I've read many or most of the classical DMTD papers, and have seen
> >>> various passing estimates that diode-ring mixers have a temperature
> >>> sensitivity of 8 to 10 pS per degree C. (I recall your figure was
10 pS/K.)
> >>> I assume that the DC offset also varies with temperature and drive
signal
> >>> amplitude.
> >>>
> >>>
> >> The only reference I have on the offset tempco is a miniciruits
> >> application note from which one can deduce that the equivalent phase
> >> shift tempco associated with the offset tempco is a few hundred
> >> femtosec/C (@ 10MHz +7dBm) at some temperatures for the particular
mixer
> >> used. The graph also indicated (if you are lucky) that the offset
tempco
> >> may be zero at around 20C.
> >>
> >
> > Do you recall the part number?
> >
> >
> >
> Supposedly an SRA-1, but some caution is in order as some statements as
> to the effect of the input offset of an opamp based IF preamp in the
> same application note were of dubious veracity unless one were to use an
> inverting opamp input stage.
This issue was mentioned in another app note, but their main issue
appeared to be that the opamp bias currents could cause an offset.
> >> A NIST paper indicated that mixer phase shift tempco was around 10x
> >> lower if the RF port was unsaturated. It also indicated that the
mixer
> >> phase shift tempco is much lower if the input frequency is 100MHz
rather
> >> than 10MHz. This was one reason given for shifting to 100MHz
> >> DMTD systems.
> >>
> >
> > Do you recall which paper?
> >
> >
> >
> http://tf.nist.gov/timefreq/general/pdf/971.pdf
> <http://tf.nist.gov/timefreq/general/pdf/971.pdf>
> Has some measurement data on mixer phase shift tempco and power
> sensitivity and their frequency dependence etc.
I do know this paper. At the bottom of page 834, to the right, is the
estimate 3.5 pS/K.
Another reason to go to 100 MHz is that the temperature coefficient of
electrical length of polymer-insulated coax is far lower at 100 MHz
compared to 10 MHz.
> I'll search for the paper that stated that the phase shift tempco was
> lower if the RF port was unsaturated.
I think I have seen this too, but don't recall where. But it's why I use
an 8 dB pad on the RF input.
> AFAIK there was no accompanying measurement data
>
> > What I've seen that seems useful is the Watkins-Johnson application
note
> > from 1978 on use of mixers as phase detectors: "Mixers as Phase
> > Detectors", Stephan R. Kurtz, 8 pages. This may be the source of the
NIST
> > article's information. The electrons are available on the web from WJ
> > Communications (now owned by TriQuint), filename
<
http://www.wj.com/archive/documents/Tech_Notes_Archived/Mixers_phase_detectors.pdf
>.
> > Don't know how long this URL will work, as WJ is assimilated into
> > TriQuint.
This app note is reference 7 of paper 971 above.
[Soundcards]
> >>>
> >> Preliminary (non optimum) tests by Ulrich indicate that picosecond
> >> stability for times up to 100sec is very easy to achieve.
> >> Beyond that mixer phase shift tempco mismatch may be significant.
> >>
> >
> > It would not be that hard to make an oven for the mixer, as
> the level of
> > control needed is far less stringent than for a crystal.
> >
> >
> >
> >> ADEV noise level of around 2E-14/Tau (1s < tau <100s).
> >> Haven't yet seen [or] have data for longer tau.
> >>
> >
> > Yes. Need at least 10^4 seconds.
> >
> >
> >
> >> With identical beat frequency outputs, crosstalk between
> >> channels within the sound card shouldn't be a great problem.
> >>
> >
> > I'm not sure I believe this, as there is likely ground coupling within
the
> > soundcard and the ear is famously insensitive to phase. Channel
isolation
> > of 60 dB isn't enough to prevent phase shifts.
> >
> >
> It will be present but its effect in some cases (when the phase shift
> between channels is such that the crosstalk phase is at 90 degrees to
> the signal of interest) will be negligible, in other cases it is easily
> measured and compensated for.
Isn't 90 degrees (quadrature) the worst case for causing phase shifts?
To get a one picosecond change at 10 MHz by injection of an attenuated
quadrature copy of the main signal requires a relative voltage ratio of
Tan[(10^-12)(10^7)(360)] = Tan[0.0036] = 0.0000628 of the main signal, or
20 Log[0.0000628]= -84 dBc. This is well exceeds the interchannel
isolation of many sound cards.
Cancellation by mathematical means could be possible, but will require a
dynamic range well exceeding 84 dB. This ought to be easy to arrange.
> >> One concern particularly for low beat frequencies is the phase shift
in
> >> the sound card input coupling capacitors (usually electrolytics).
> >>
> >> It should be easy to test the sound card phase shift stability for
this
> >> application by driving both inputs from the same signal source.
Or terminate one channel input and drive the other, and measure the
amplitude and phase of whatever comes out of the terminated channel,
compared to the driven channel. Then swap channels and repeat. The phase
and amplitude will depend on frequency, so a sweep will be required, and
some frequencies may need to be avoided.
> > I assume that the beatnote must be ~100 Hz for the soundcard to handle
> > with low phase shift. One might get to 10 Hz, but 1 Hz is likely
> > hopeless.
> >
> > One thing that will be very useful is a list of sound cards by make
and
> > model, annotated with their advantages and disadvantages for time-nut
use.
> >
> > "High-end" may not be a sufficient description.
> >
> >
> > By the way, I looked at the operating and service manual for the HP
> > K34-59991A Broadband Linear Phase Comparator. Very interesting little
> > gadget, but little performance data was given. Does anyone know how
well
> > phase change can be measured? It would be easy to duplicate this with
> > modern ICs. Also, about when was this unit made? The manual has no
date.
> >
> >
> > Joe
> >
> >
> Joe
>
> I suspect that slow phase changes much less than 1ns or so are hard to
> distinguish from gain drift given the gain tempco of the ECL
> phase detector.
>
> A beat note near 1kHz appears to be even better if one is using
> something like an enhanced Costas receiver or even using WKS
> interpolation to locate and time stamp zero crossings.
But it limits the phase slope gain. I suppose there is an optimum
somewhere.
> So far only the M-Audio AP192 has been used.
> Tests with an embedded motherboard 16 bit sound system show
> significantly increased noise.
> I've found that the noise level of motherboard sound systems varies
> enormously from one motherboard model (sample of 2) to another.
>
> Any 24 bit sound card with a performance close to or better than that of
> the AP192 should suffice.
In general, firewire connected sound cards should be better, because the
soundcard maker has complete control of what's inside the box. Unlike
inside a PC.
> Other cards using AKM 24 bit ADCs should also be suitable.
Who is AKM?
20 Log[ 2^24 ] = 144 dB, so something else will be the limit.
> Ideally an external sound card with balanced XLR inputs would be best.
>
> HP produced a number of different phase comparators each with a
> different type of phase detector.
> The K34-5991A design can't be older than the early 1970's because the
> MECLIII devices used weren't available until then.
OK. I recall MECL. RIP. But we have PECL now.
> Warren built a similar phase detector (differential XOR or XOR + XNOR)
> using CMOS ICs and for a common 10MHz input with a phase difference near
> zero found short term output noise of of around 10uV or so (10V phase
> detector FSR) using a passive low pass filter.
(10uV)/(10v)= 1 ppm. (100 nS)(10^-6)= 0.1 pS.
> In principle an ADC like the LTC2484 could be used with a 2.5V CMOS
> OR/XNOR phase detector and passive low pass filters.
> The ratiometric conversion capability will significantly reduce the
> sensitivity to the XOR gate supply if the XOR gate supply is also used
> for the ADC reference voltage.
> If one used 5V logic a resistive output attenuator would be needed which
> reduce the gain stability somewhat.
>
> All such phase detectors suffer from substantial nonlinearity near the
> ends of the range due to gate output slew rate limitations.
If one is tracking through multiple phase cycles (as did the HP unit),
this would matter.
As for slew rate, one can buy faster logic, but probably at the expense of
jitter (as bandwidth must be larger).
The HP unit used balanced logic all the way up to the flipflops. This
doubles the relative crossing rate for a given slew rate. But doubles the
noise, but it must have been worth it, or HP wouldn't have done that.
In beatnote zero crossing detectors, differential signals help keep ground
loops and bounce down.
> However if operated near the centre of the range subpicosecond
> sensitivity/short term stability may be possible.
Yes, 0.1 pS.
> Since the circuit volume is small and the tempco relatively low (few
> ps/C at most) regulating the circuit temperature should be relatively
> easy to do.
We ought to be able to achieve less than 0.1 K change.
Joe
JM
Joseph M Gwinn
Wed, Dec 10, 2008 1:21 AM
This one is new to me, and interesting. More later, when I've read
I had found this one, and is why I used a 3 dB attenuator on the LO
and a 8 dB attenuator on the RF port. But the pieces are coming
bin/mailman/listinfo/time-nuts
and follow the instructions there.
Yes, I did know of that one, but thanks. It seems largely based on prior
papers, a summation of sorts.
Joe
Bruce,
time-nuts-bounces@febo.com wrote on 12/09/2008 07:55:50 PM:
> Joseph M Gwinn wrote:
> > Bruce,
> >
> >
> > time-nuts-bounces@febo.com wrote on 12/09/2008 07:10:55 PM:
> >
> >
> >> Paper with capacitive IF port termination data:
> >>
> >> http://tf.nist.gov/timefreq/general/pdf/112.pdf
> >>
> >
> > This one is new to me, and interesting. More later, when I've read
it.
> >
> >
> >
> >> Phase detector sensitivity to distortion:
> >>
> >> http://tf.nist.gov/timefreq/general/pdf/1437.pdf
> >>
> >
> > I had found this one, and is why I used a 3 dB attenuator on the LO
port
> > and a 8 dB attenuator on the RF port. But the pieces are coming
together.
> >
> >
> > Thanks,
> >
> > Joe
> >
> > _______________________________________________
> > time-nuts mailing list -- time-nuts@febo.com
> > To unsubscribe, go to https://www.febo.com/cgi-
> bin/mailman/listinfo/time-nuts
> > and follow the instructions there.
> >
> >
> Joe
>
> Did you also look at?:
>
> http://tf.nist.gov/timefreq/general/pdf/971.pdf
>
> Which indicates how one can tune the phase shift tempco to some extent.
Yes, I did know of that one, but thanks. It seems largely based on prior
papers, a summation of sorts.
Joe
BG
Bruce Griffiths
Wed, Dec 10, 2008 2:20 AM
Joe
Joseph M Gwinn wrote:
By the way, despite the circuit diagram in the datasheet, the
corresponding phase-detector module MPD-1 can be wired to have the IF
output ground isolated from the common RF, LO and case ground. A
little work with an ohmmeter will tell the tale. This can help to
Yes, that's usually the case for the Minicircuits PCB mount phase
detectors and mixers except for some surface mount versions (usually the
very high frequency models).
So it was already known. It looks to me that MiniCircuit's intent is to
support automated testing of modules.
A PCB mount mixer package is also preferable as its then much easier to
use a capacitive IF port termination (for lower noise) in conjunction
with series resistors at the RF and LO ports (for lower VSWR) than if a
mixer with SMA or other coax connectors were used.
I've been using 3 and 8 dB coaxial attenuators at the LO and RF inputs
respectively, and it makes a big difference.
But I don't understand the part about capacitive loading of the IF port. I
would think that the low pass filter would need to present a matched
impedance at the sum frequency, so the emerging high-level 20 MHz signal
is not reflected back into the mixer.
Reflecting the sum frequency back into the mixer is actually necessary
to reduce the noise at the IF port.
I believe that one of Agilent's simulation application notes mentions
this effect but I don't recall the actual application note number.
This will affect the mixer RF and IF port impedance so adding a series
resistor may be required to improve the SWR.
MiniCircuits AN-41-001 "FAQ about Phase Detectors" has on page 2 a 500 ohm
resistor to ground and a 5000 ohm resistor to the first filter capacitor,
so the capacitor is isolated from the IF port by the resistors.
I wouldn't take too much notice of that recommendation as I have little
confidence in the author's experience/knowledge.
Supposedly an SRA-1, but some caution is in order as some statements as
to the effect of the input offset of an opamp based IF preamp in the
same application note were of dubious veracity unless one were to use an
inverting opamp input stage.
This issue was mentioned in another app note, but their main issue
appeared to be that the opamp bias currents could cause an offset.
But the circuit they suggest has no effect on bias current induced
offset, the same current flows into the mixer and termination impedance
independent of the series resistance.
A NIST paper indicated that mixer phase shift tempco was around 10x
lower if the RF port was unsaturated. It also indicated that the
phase shift tempco is much lower if the input frequency is 100MHz
than 10MHz. This was one reason given for shifting to 100MHz
DMTD systems.
Do you recall which paper?
I do know this paper. At the bottom of page 834, to the right, is the
estimate 3.5 pS/K.
Another reason to go to 100 MHz is that the temperature coefficient of
electrical length of polymer-insulated coax is far lower at 100 MHz
compared to 10 MHz.
I'll search for the paper that stated that the phase shift tempco was
lower if the RF port was unsaturated.
Don't know how long this URL will work, as WJ is assimilated into
TriQuint.
This app note is reference 7 of paper 971 above.
[Soundcards]
With identical beat frequency outputs, crosstalk between
channels within the sound card shouldn't be a great problem.
I'm not sure I believe this, as there is likely ground coupling within
soundcard and the ear is famously insensitive to phase. Channel
of 60 dB isn't enough to prevent phase shifts.
It will be present but its effect in some cases (when the phase shift
between channels is such that the crosstalk phase is at 90 degrees to
the signal of interest) will be negligible, in other cases it is easily
measured and compensated for.
Isn't 90 degrees (quadrature) the worst case for causing phase shifts?
Yes, I should have said that when the 2 input signals are in quadrature,
any capacitive crosstalk will have little effect on the phase shift.
To get a one picosecond change at 10 MHz by injection of an attenuated
quadrature copy of the main signal requires a relative voltage ratio of
Tan[(10^-12)(10^7)(360)] = Tan[0.0036] = 0.0000628 of the main signal, or
20 Log[0.0000628]= -84 dBc. This is well exceeds the interchannel
isolation of many sound cards.
Cancellation by mathematical means could be possible, but will require a
dynamic range well exceeding 84 dB. This ought to be easy to arrange.
The AP192 has a somewhat higher interchannel isolation than that, the
interchannel crosstalk spec is about -120dB.
With a sufficiently large number of samples the its easy to see
artifacts as low as -140dBFS.
One concern particularly for low beat frequencies is the phase shift
the sound card input coupling capacitors (usually electrolytics).
It should be easy to test the sound card phase shift stability for
application by driving both inputs from the same signal source.
Or terminate one channel input and drive the other, and measure the
amplitude and phase of whatever comes out of the terminated channel,
compared to the driven channel. Then swap channels and repeat. The phase
and amplitude will depend on frequency, so a sweep will be required, and
some frequencies may need to be avoided.
Joe
I suspect that slow phase changes much less than 1ns or so are hard to
distinguish from gain drift given the gain tempco of the ECL
phase detector.
A beat note near 1kHz appears to be even better if one is using
something like an enhanced Costas receiver or even using WKS
interpolation to locate and time stamp zero crossings.
But it limits the phase slope gain. I suppose there is an optimum
somewhere.
So far only the M-Audio AP192 has been used.
Tests with an embedded motherboard 16 bit sound system show
significantly increased noise.
I've found that the noise level of motherboard sound systems varies
enormously from one motherboard model (sample of 2) to another.
Any 24 bit sound card with a performance close to or better than that of
the AP192 should suffice.
In general, firewire connected sound cards should be better, because the
soundcard maker has complete control of what's inside the box. Unlike
inside a PC.
Its hard to find such Firewire systems without such unnecessary frills
(for this application) as high gain preamps.
The gain tempco and linearity of some variable gain audio preamps is
somewhat suspect.
Other cards using AKM 24 bit ADCs should also be suitable.
20 Log[ 2^24 ] = 144 dB, so something else will be the limit.
Actual ENOB ~ 19 to 20 bits.
Ideally an external sound card with balanced XLR inputs would be best.
HP produced a number of different phase comparators each with a
different type of phase detector.
The K34-5991A design can't be older than the early 1970's because the
MECLIII devices used weren't available until then.
OK. I recall MECL. RIP. But we have PECL now.
Same thing, different supplies.
PECL when Vcc is +ve and Vee is GND.
NECL when Vcc is GND and Vee is -ve.
Warren built a similar phase detector (differential XOR or XOR + XNOR)
using CMOS ICs and for a common 10MHz input with a phase difference near
zero found short term output noise of of around 10uV or so (10V phase
detector FSR) using a passive low pass filter.
(10uV)/(10v)= 1 ppm. (100 nS)(10^-6)= 0.1 pS.
In principle an ADC like the LTC2484 could be used with a 2.5V CMOS
OR/XNOR phase detector and passive low pass filters.
The ratiometric conversion capability will significantly reduce the
sensitivity to the XOR gate supply if the XOR gate supply is also used
for the ADC reference voltage.
If one used 5V logic a resistive output attenuator would be needed which
reduce the gain stability somewhat.
All such phase detectors suffer from substantial nonlinearity near the
ends of the range due to gate output slew rate limitations.
If one is tracking through multiple phase cycles (as did the HP unit),
this would matter.
Can alleviate it to some extent by driving a pair of such phase
detectors so that their outputs are in quadrature.
One just selects the phase detector output that is in the linear range.
The quadrature outputs also allow unambiguous assignment of the sign of
any phase change.
Joe
Joseph M Gwinn wrote:
> Bruce,
>
>>> By the way, despite the circuit diagram in the datasheet, the
>>> corresponding phase-detector module MPD-1 can be wired to have the IF
>>> output ground isolated from the common RF, LO and case ground. A
>>> little work with an ohmmeter will tell the tale. This can help to
>>>
> contain the
>
>>> low frequency beatnote.
>>>
>> Yes, that's usually the case for the Minicircuits PCB mount phase
>> detectors and mixers except for some surface mount versions (usually the
>> very high frequency models).
>>
>
> So it was already known. It looks to me that MiniCircuit's intent is to
> support automated testing of modules.
>
>
>
>> A PCB mount mixer package is also preferable as its then much easier to
>> use a capacitive IF port termination (for lower noise) in conjunction
>> with series resistors at the RF and LO ports (for lower VSWR) than if a
>> mixer with SMA or other coax connectors were used.
>>
>
> I've been using 3 and 8 dB coaxial attenuators at the LO and RF inputs
> respectively, and it makes a big difference.
>
> But I don't understand the part about capacitive loading of the IF port. I
> would think that the low pass filter would need to present a matched
> impedance at the sum frequency, so the emerging high-level 20 MHz signal
> is not reflected back into the mixer.
>
Reflecting the sum frequency back into the mixer is actually necessary
to reduce the noise at the IF port.
I believe that one of Agilent's simulation application notes mentions
this effect but I don't recall the actual application note number.
This will affect the mixer RF and IF port impedance so adding a series
resistor may be required to improve the SWR.
> MiniCircuits AN-41-001 "FAQ about Phase Detectors" has on page 2 a 500 ohm
> resistor to ground and a 5000 ohm resistor to the first filter capacitor,
> so the capacitor is isolated from the IF port by the resistors.
>
>
I wouldn't take too much notice of that recommendation as I have little
confidence in the author's experience/knowledge.
>> Supposedly an SRA-1, but some caution is in order as some statements as
>> to the effect of the input offset of an opamp based IF preamp in the
>> same application note were of dubious veracity unless one were to use an
>> inverting opamp input stage.
>>
>
> This issue was mentioned in another app note, but their main issue
> appeared to be that the opamp bias currents could cause an offset.
>
>
>
But the circuit they suggest has no effect on bias current induced
offset, the same current flows into the mixer and termination impedance
independent of the series resistance.
>>>> A NIST paper indicated that mixer phase shift tempco was around 10x
>>>> lower if the RF port was unsaturated. It also indicated that the
>>>>
> mixer
>
>>>> phase shift tempco is much lower if the input frequency is 100MHz
>>>>
> rather
>
>>>> than 10MHz. This was one reason given for shifting to 100MHz
>>>> DMTD systems.
>>>>
>>>>
>>> Do you recall which paper?
>>>
>>>
>>>
>>>
>> http://tf.nist.gov/timefreq/general/pdf/971.pdf
>> <http://tf.nist.gov/timefreq/general/pdf/971.pdf>
>> Has some measurement data on mixer phase shift tempco and power
>> sensitivity and their frequency dependence etc.
>>
>
> I do know this paper. At the bottom of page 834, to the right, is the
> estimate 3.5 pS/K.
>
> Another reason to go to 100 MHz is that the temperature coefficient of
> electrical length of polymer-insulated coax is far lower at 100 MHz
> compared to 10 MHz.
>
>
>
>> I'll search for the paper that stated that the phase shift tempco was
>> lower if the RF port was unsaturated.
>>
>
> I think I have seen this too, but don't recall where. But it's why I use
> an 8 dB pad on the RF input.
>
>
> http://www.wj.com/archive/documents/Tech_Notes_Archived/Mixers_phase_detectors.pdf
>
>> .
>>
>>> Don't know how long this URL will work, as WJ is assimilated into
>>> TriQuint.
>>>
>
> This app note is reference 7 of paper 971 above.
>
>
> [Soundcards]
>>>
>>>> With identical beat frequency outputs, crosstalk between
>>>> channels within the sound card shouldn't be a great problem.
>>>>
>>>>
>>> I'm not sure I believe this, as there is likely ground coupling within
>>>
> the
>
>>> soundcard and the ear is famously insensitive to phase. Channel
>>>
> isolation
>
>>> of 60 dB isn't enough to prevent phase shifts.
>>>
>>>
>>>
>> It will be present but its effect in some cases (when the phase shift
>> between channels is such that the crosstalk phase is at 90 degrees to
>> the signal of interest) will be negligible, in other cases it is easily
>> measured and compensated for.
>>
>
> Isn't 90 degrees (quadrature) the worst case for causing phase shifts?
>
>
Yes, I should have said that when the 2 input signals are in quadrature,
any capacitive crosstalk will have little effect on the phase shift.
> To get a one picosecond change at 10 MHz by injection of an attenuated
> quadrature copy of the main signal requires a relative voltage ratio of
> Tan[(10^-12)(10^7)(360)] = Tan[0.0036] = 0.0000628 of the main signal, or
> 20 Log[0.0000628]= -84 dBc. This is well exceeds the interchannel
> isolation of many sound cards.
>
> Cancellation by mathematical means could be possible, but will require a
> dynamic range well exceeding 84 dB. This ought to be easy to arrange.
>
>
The AP192 has a somewhat higher interchannel isolation than that, the
interchannel crosstalk spec is about -120dB.
With a sufficiently large number of samples the its easy to see
artifacts as low as -140dBFS.
>
>>>> One concern particularly for low beat frequencies is the phase shift
>>>>
> in
>
>>>> the sound card input coupling capacitors (usually electrolytics).
>>>>
>>>> It should be easy to test the sound card phase shift stability for
>>>>
> this
>
>>>> application by driving both inputs from the same signal source.
>>>>
>
> Or terminate one channel input and drive the other, and measure the
> amplitude and phase of whatever comes out of the terminated channel,
> compared to the driven channel. Then swap channels and repeat. The phase
> and amplitude will depend on frequency, so a sweep will be required, and
> some frequencies may need to be avoided.
>
>
>
>>>
>> Joe
>>
>> I suspect that slow phase changes much less than 1ns or so are hard to
>> distinguish from gain drift given the gain tempco of the ECL
>> phase detector.
>>
>> A beat note near 1kHz appears to be even better if one is using
>> something like an enhanced Costas receiver or even using WKS
>> interpolation to locate and time stamp zero crossings.
>>
>
> But it limits the phase slope gain. I suppose there is an optimum
> somewhere.
>
>
>
>> So far only the M-Audio AP192 has been used.
>> Tests with an embedded motherboard 16 bit sound system show
>> significantly increased noise.
>> I've found that the noise level of motherboard sound systems varies
>> enormously from one motherboard model (sample of 2) to another.
>>
>> Any 24 bit sound card with a performance close to or better than that of
>> the AP192 should suffice.
>>
>
> In general, firewire connected sound cards should be better, because the
> soundcard maker has complete control of what's inside the box. Unlike
> inside a PC.
>
>
Its hard to find such Firewire systems without such unnecessary frills
(for this application) as high gain preamps.
The gain tempco and linearity of some variable gain audio preamps is
somewhat suspect.
>
>> Other cards using AKM 24 bit ADCs should also be suitable.
>>
>
> Who is AKM?
>
>
Asahi Kasei EKM
http://www.asahi-kasei.co.jp/akm/en/
http://www.asahi-kasei.co.jp/akm/en/product/proaudio.html
> 20 Log[ 2^24 ] = 144 dB, so something else will be the limit.
>
>
>
Actual ENOB ~ 19 to 20 bits.
>> Ideally an external sound card with balanced XLR inputs would be best.
>>
>> HP produced a number of different phase comparators each with a
>> different type of phase detector.
>> The K34-5991A design can't be older than the early 1970's because the
>> MECLIII devices used weren't available until then.
>>
>
> OK. I recall MECL. RIP. But we have PECL now.
>
>
Same thing, different supplies.
PECL when Vcc is +ve and Vee is GND.
NECL when Vcc is GND and Vee is -ve.
>
>> Warren built a similar phase detector (differential XOR or XOR + XNOR)
>> using CMOS ICs and for a common 10MHz input with a phase difference near
>> zero found short term output noise of of around 10uV or so (10V phase
>> detector FSR) using a passive low pass filter.
>>
>
> (10uV)/(10v)= 1 ppm. (100 nS)(10^-6)= 0.1 pS.
>
>
>
>> In principle an ADC like the LTC2484 could be used with a 2.5V CMOS
>> OR/XNOR phase detector and passive low pass filters.
>> The ratiometric conversion capability will significantly reduce the
>> sensitivity to the XOR gate supply if the XOR gate supply is also used
>> for the ADC reference voltage.
>> If one used 5V logic a resistive output attenuator would be needed which
>> reduce the gain stability somewhat.
>>
>> All such phase detectors suffer from substantial nonlinearity near the
>> ends of the range due to gate output slew rate limitations.
>>
>
> If one is tracking through multiple phase cycles (as did the HP unit),
> this would matter.
>
>
Can alleviate it to some extent by driving a pair of such phase
detectors so that their outputs are in quadrature.
One just selects the phase detector output that is in the linear range.
The quadrature outputs also allow unambiguous assignment of the sign of
any phase change.
> Joe
>
>
Bruce
BG
Bruce Griffiths
Wed, Dec 10, 2008 9:46 PM
Joe
Another limitation on using too low a beat frequency is imposed by the
increasing equivalent input noise spectral density of the sound card as
the frequency decreases.
Bruce
Joe
Another limitation on using too low a beat frequency is imposed by the
increasing equivalent input noise spectral density of the sound card as
the frequency decreases.
Bruce
JM
Joseph M Gwinn
Thu, Dec 11, 2008 12:48 AM
Joe
Joseph M Gwinn wrote:
A PCB mount mixer package is also preferable as its then much easier
use a capacitive IF port termination (for lower noise) in conjunction
with series resistors at the RF and LO ports (for lower VSWR) than if
mixer with SMA or other coax connectors were used.
I've been using 3 and 8 dB coaxial attenuators at the LO and RF inputs
respectively, and it makes a big difference.
But I don't understand the part about capacitive loading of the IF
would think that the low pass filter would need to present a matched
impedance at the sum frequency, so the emerging high-level 20 MHz
is not reflected back into the mixer.
Reflecting the sum frequency back into the mixer is actually necessary
to reduce the noise at the IF port.
I believe that one of Agilent's simulation application notes mentions
this effect but I don't recall the actual application note number.
This will affect the mixer RF and IF port impedance so adding a series
resistor may be required to improve the SWR.
How big an effect is this? Is the absolute noise decreased, or does it
remain the same while the signal increase?
If I'm understanding Walls and Stein (paper 112) correctly, the advantage
is because with the capacitor load the beatnote waveform approaches
square, thus increasing the zero-crossing speed and therefor the phase
sensitivity. This is no doubt true, but the question was if this also
caused a small everything-dependent phase shift, something that would not
have mattered in the measurement of phase noise. The object of paper 112
was to remedy a 10 to 20 dB error in phase noise measurements. The
critical words are in the lower left column of page 337, in the paragraph
beginning "If the mixer is terminated ...".
MiniCircuits AN-41-001 "FAQ about Phase Detectors" has on page 2 a 500
resistor to ground and a 5000 ohm resistor to the first filter
so the capacitor is isolated from the IF port by the resistors.
I wouldn't take too much notice of that recommendation as I have little
confidence in the author's experience/knowledge.
Well, OK, but:
Stephen Kurtz says the same thing on the third column of the third page, a
bit above Figure 6.
Nelson and Walls (paper 971), Figure 4, also shows the low pass filter
arranged to absorb the sum signal, not allowing it to be reflected back
into the mixer.
Supposedly an SRA-1, but some caution is in order as some
to the effect of the input offset of an opamp based IF preamp in the
same application note were of dubious veracity unless one
inverting opamp input stage.
This issue was mentioned in another app note, but their main issue
appeared to be that the opamp bias currents could cause an offset.
But the circuit they suggest has no effect on bias current induced
offset, the same current flows into the mixer and termination impedance
independent of the series resistance.
You're right that the proposed remedy didn't make sense. I don't know
that this is a big problem with modern opamps, especially FET input ones
(if needed).
[snip]
With identical beat frequency outputs, crosstalk between
channels within the sound card shouldn't be a great problem.
I'm not sure I believe this, as there is likely ground coupling
soundcard and the ear is famously insensitive to phase. Channel
isolation of 60 dB isn't enough to prevent phase shifts.
It will be present but its effect in some cases (when the phase shift
between channels is such that the crosstalk phase is at 90 degrees to
the signal of interest) will be negligible, in other cases it is
measured and compensated for.
Isn't 90 degrees (quadrature) the worst case for causing phase shifts?
Yes, I should have said that when the 2 input signals are in quadrature,
any capacitive crosstalk will have little effect on the phase shift.
Ah. Because the capacitor coupling adds a second 90 degree shift,
bringing the total to 180 degrees.
But crosstalk by ground coupling will be unaffected. As will crosstalk by
transformer action. Those boards are pretty crowded.
To get a one picosecond change at 10 MHz by injection of an attenuated
quadrature copy of the main signal requires a relative voltage ratio
Tan[(10^-12)(10^7)(360)] = Tan[0.0036] = 0.0000628 of the main signal,
20 Log[0.0000628]= -84 dBc. This is well exceeds the interchannel
isolation of many sound cards.
Cancellation by mathematical means could be possible, but will require
dynamic range well exceeding 84 dB. This ought to be easy to arrange.
The AP192 has a somewhat higher interchannel isolation than that, the
interchannel crosstalk spec is about -120dB.
With a sufficiently large number of samples the its easy to see
artifacts as low as -140dBFS.
Yep. Seems like a very good card.
One concern particularly for low beat frequencies is the phase
in the sound card input coupling capacitors (usually
It should be easy to test the sound card phase shift stability for
this application by driving both inputs from the same signal
Or terminate one channel input and drive the other, and measure the
amplitude and phase of whatever comes out of the terminated channel,
compared to the driven channel. Then swap channels and repeat. The
and amplitude will depend on frequency, so a sweep will be required,
some frequencies may need to be avoided.
Joe
I suspect that slow phase changes much less than 1ns or so are hard
distinguish from gain drift given the gain tempco of the ECL
phase detector.
A beat note near 1kHz appears to be even better if one is using
something like an enhanced Costas receiver or even using WKS
interpolation to locate and time stamp zero crossings.
But it limits the phase slope gain. I suppose there is an optimum
somewhere.
So far only the M-Audio AP192 has been used.
Tests with an embedded motherboard 16 bit sound system show
significantly increased noise.
I've found that the noise level of motherboard sound systems varies
enormously from one motherboard model (sample of 2) to another.
Any 24 bit sound card with a performance close to or better than that
the AP192 should suffice.
In general, firewire connected sound cards should be better, because
soundcard maker has complete control of what's inside the box. Unlike
It's hard to find such Firewire systems without such unnecessary frills
(for this application) as high gain preamps.
The AP192 has high-level inputs, but I don't know if this bypasses the
preamps, or attenuates. Given their target market, I'd bet it bypasses.
The gain tempco and linearity of some variable gain audio preamps is
somewhat suspect.
I would think that none of these cards has a good tempco of anything,
given the lack of necessity in their market.
I would think that linearity would be quite good, given the horsepower
competitions on linearity.
Other cards using AKM 24 bit ADCs should also be suitable.
Thanks. I'll look into their data.
20 Log[ 2^24 ] = 144 dB, so something else will be the limit.
Actual ENOB ~ 19 to 20 bits.
Makes sense. 20 Log [ 2^19 ] = 114 dB. Still plenty good enough.
Ideally an external sound card with balanced XLR inputs would be
HP produced a number of different phase comparators each with a
different type of phase detector.
OK. And the PLL folk must have a million designs.
The K34-5991A design can't be older than the early 1970's because the
MECLIII devices used weren't available until then.
OK. I recall MECL. RIP. But we have PECL now.
Same thing, different supplies.
PECL when Vcc is +ve and Vee is GND.
NECL when Vcc is GND and Vee is -ve.
Warren built a similar phase detector (differential XOR or XOR +
using CMOS ICs and for a common 10MHz input with a phase difference
zero found short term output noise of of around 10uV or so (10V phase
detector FSR) using a passive low pass filter.
(10uV)/(10v)= 1 ppm. (100 nS)(10^-6)= 0.1 pS.
In principle an ADC like the LTC2484 could be used with a 2.5V CMOS
OR/XNOR phase detector and passive low pass filters.
The ratiometric conversion capability will significantly reduce the
sensitivity to the XOR gate supply if the XOR gate supply isalso used
for the ADC reference voltage.
If one used 5V logic a resistive output attenuator would be needed
reduce the gain stability somewhat.
All such phase detectors suffer from substantial nonlinearity near
ends of the range due to gate output slew rate limitations.
If one is tracking through multiple phase cycles (as did the HP unit),
Can alleviate it to some extent by driving a pair of such phase
detectors so that their outputs are in quadrature.
One just selects the phase detector output that is in the linear range.
The quadrature outputs also allow unambiguous assignment of the sign of
any phase change.
The Symmetricom 5120A does something very clever to alleviate this
problem. Explained in US patent 7,227,346 and "Direct-Digital Phase-Noise
Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger, W. Solbrig, and
S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics, and
Frequency Control Joint 50th Anniversary Conference, pages 287-291.
Joe
Bruce,
time-nuts-bounces@febo.com wrote on 12/09/2008 09:20:08 PM:
> Joe
>
> Joseph M Gwinn wrote:
> > Bruce,
> >
> >> A PCB mount mixer package is also preferable as its then much easier
to
> >> use a capacitive IF port termination (for lower noise) in conjunction
> >> with series resistors at the RF and LO ports (for lower VSWR) than if
a
> >> mixer with SMA or other coax connectors were used.
> >>
> >
> > I've been using 3 and 8 dB coaxial attenuators at the LO and RF inputs
> > respectively, and it makes a big difference.
> >
> > But I don't understand the part about capacitive loading of the IF
port. I
> > would think that the low pass filter would need to present a matched
> > impedance at the sum frequency, so the emerging high-level 20 MHz
signal
> > is not reflected back into the mixer.
> >
> Reflecting the sum frequency back into the mixer is actually necessary
> to reduce the noise at the IF port.
> I believe that one of Agilent's simulation application notes mentions
> this effect but I don't recall the actual application note number.
> This will affect the mixer RF and IF port impedance so adding a series
> resistor may be required to improve the SWR.
How big an effect is this? Is the absolute noise decreased, or does it
remain the same while the signal increase?
If I'm understanding Walls and Stein (paper 112) correctly, the advantage
is because with the capacitor load the beatnote waveform approaches
square, thus increasing the zero-crossing speed and therefor the phase
sensitivity. This is no doubt true, but the question was if this also
caused a small everything-dependent phase shift, something that would not
have mattered in the measurement of phase noise. The object of paper 112
was to remedy a 10 to 20 dB error in phase noise measurements. The
critical words are in the lower left column of page 337, in the paragraph
beginning "If the mixer is terminated ...".
> > MiniCircuits AN-41-001 "FAQ about Phase Detectors" has on page 2 a 500
ohm
> > resistor to ground and a 5000 ohm resistor to the first filter
capacitor,
> > so the capacitor is isolated from the IF port by the resistors.
> >
> >
> I wouldn't take too much notice of that recommendation as I have little
> confidence in the author's experience/knowledge.
Well, OK, but:
Stephen Kurtz says the same thing on the third column of the third page, a
bit above Figure 6.
Nelson and Walls (paper 971), Figure 4, also shows the low pass filter
arranged to absorb the sum signal, not allowing it to be reflected back
into the mixer.
> >> Supposedly an SRA-1, but some caution is in order as some
> statements as
> >> to the effect of the input offset of an opamp based IF preamp in the
> >> same application note were of dubious veracity unless one
> were to use an
> >> inverting opamp input stage.
> >>
> >
> > This issue was mentioned in another app note, but their main issue
> > appeared to be that the opamp bias currents could cause an offset.
> >
> >
> >
> But the circuit they suggest has no effect on bias current induced
> offset, the same current flows into the mixer and termination impedance
> independent of the series resistance.
You're right that the proposed remedy didn't make sense. I don't know
that this is a big problem with modern opamps, especially FET input ones
(if needed).
[snip]
> >
> > [Soundcards]
> >>>
> >>>> With identical beat frequency outputs, crosstalk between
> >>>> channels within the sound card shouldn't be a great problem.
> >>>>
> >>>>
> >>> I'm not sure I believe this, as there is likely ground coupling
within the
> >>> soundcard and the ear is famously insensitive to phase. Channel
> >>> isolation of 60 dB isn't enough to prevent phase shifts.
> >>>
> >>>
> >>>
> >> It will be present but its effect in some cases (when the phase shift
> >> between channels is such that the crosstalk phase is at 90 degrees to
> >> the signal of interest) will be negligible, in other cases it is
easily
> >> measured and compensated for.
> >>
> >
> > Isn't 90 degrees (quadrature) the worst case for causing phase shifts?
> >
> >
> Yes, I should have said that when the 2 input signals are in quadrature,
> any capacitive crosstalk will have little effect on the phase shift.
Ah. Because the capacitor coupling adds a second 90 degree shift,
bringing the total to 180 degrees.
But crosstalk by ground coupling will be unaffected. As will crosstalk by
transformer action. Those boards are pretty crowded.
> > To get a one picosecond change at 10 MHz by injection of an attenuated
> > quadrature copy of the main signal requires a relative voltage ratio
of
> > Tan[(10^-12)(10^7)(360)] = Tan[0.0036] = 0.0000628 of the main signal,
or
> > 20 Log[0.0000628]= -84 dBc. This is well exceeds the interchannel
> > isolation of many sound cards.
> >
> > Cancellation by mathematical means could be possible, but will require
a
> > dynamic range well exceeding 84 dB. This ought to be easy to arrange.
> >
> >
> The AP192 has a somewhat higher interchannel isolation than that, the
> interchannel crosstalk spec is about -120dB.
> With a sufficiently large number of samples the its easy to see
> artifacts as low as -140dBFS.
Yep. Seems like a very good card.
> >>>> One concern particularly for low beat frequencies is the phase
shift
> >>>> in the sound card input coupling capacitors (usually
electrolytics).
> >>>>
> >>>> It should be easy to test the sound card phase shift stability for
> >>>> this application by driving both inputs from the same signal
source.
> >>>>
> >
> > Or terminate one channel input and drive the other, and measure the
> > amplitude and phase of whatever comes out of the terminated channel,
> > compared to the driven channel. Then swap channels and repeat. The
phase
> > and amplitude will depend on frequency, so a sweep will be required,
and
> > some frequencies may need to be avoided.
> >
> >
> >
> >>>
> >> Joe
> >>
> >> I suspect that slow phase changes much less than 1ns or so are hard
to
> >> distinguish from gain drift given the gain tempco of the ECL
> >> phase detector.
> >>
> >> A beat note near 1kHz appears to be even better if one is using
> >> something like an enhanced Costas receiver or even using WKS
> >> interpolation to locate and time stamp zero crossings.
> >>
> >
> > But it limits the phase slope gain. I suppose there is an optimum
> > somewhere.
> >
> >
> >
> >> So far only the M-Audio AP192 has been used.
> >> Tests with an embedded motherboard 16 bit sound system show
> >> significantly increased noise.
> >> I've found that the noise level of motherboard sound systems varies
> >> enormously from one motherboard model (sample of 2) to another.
> >>
> >> Any 24 bit sound card with a performance close to or better than that
of
> >> the AP192 should suffice.
> >>
> >
> > In general, firewire connected sound cards should be better, because
the
> > soundcard maker has complete control of what's inside the box. Unlike
> > inside a PC.
> >
> >
> It's hard to find such Firewire systems without such unnecessary frills
> (for this application) as high gain preamps.
The AP192 has high-level inputs, but I don't know if this bypasses the
preamps, or attenuates. Given their target market, I'd bet it bypasses.
> The gain tempco and linearity of some variable gain audio preamps is
> somewhat suspect.
I would think that none of these cards has a good tempco of anything,
given the lack of necessity in their market.
I would think that linearity would be quite good, given the horsepower
competitions on linearity.
> >> Other cards using AKM 24 bit ADCs should also be suitable.
> >>
> >
> > Who is AKM?
> >
> >
> Asahi Kasei EKM
>
> http://www.asahi-kasei.co.jp/akm/en/
>
> http://www.asahi-kasei.co.jp/akm/en/product/proaudio.html
>
Thanks. I'll look into their data.
> > 20 Log[ 2^24 ] = 144 dB, so something else will be the limit.
> >
> >
> >
> Actual ENOB ~ 19 to 20 bits.
Makes sense. 20 Log [ 2^19 ] = 114 dB. Still plenty good enough.
> >> Ideally an external sound card with balanced XLR inputs would be
best.
Yes.
> >> HP produced a number of different phase comparators each with a
> >> different type of phase detector.
OK. And the PLL folk must have a million designs.
> >> The K34-5991A design can't be older than the early 1970's because the
> >> MECLIII devices used weren't available until then.
> >>
> >
> > OK. I recall MECL. RIP. But we have PECL now.
> >
> >
> Same thing, different supplies.
> PECL when Vcc is +ve and Vee is GND.
> NECL when Vcc is GND and Vee is -ve.
> >
> >> Warren built a similar phase detector (differential XOR or XOR +
XNOR)
> >> using CMOS ICs and for a common 10MHz input with a phase difference
near
> >> zero found short term output noise of of around 10uV or so (10V phase
> >> detector FSR) using a passive low pass filter.
> >>
> >
> > (10uV)/(10v)= 1 ppm. (100 nS)(10^-6)= 0.1 pS.
> >
> >
> >
> >> In principle an ADC like the LTC2484 could be used with a 2.5V CMOS
> >> OR/XNOR phase detector and passive low pass filters.
> >> The ratiometric conversion capability will significantly reduce the
> >> sensitivity to the XOR gate supply if the XOR gate supply isalso used
> >> for the ADC reference voltage.
> >> If one used 5V logic a resistive output attenuator would be needed
which
> >> reduce the gain stability somewhat.
> >>
> >> All such phase detectors suffer from substantial nonlinearity near
the
> >> ends of the range due to gate output slew rate limitations.
> >>
> >
> > If one is tracking through multiple phase cycles (as did the HP unit),
> > this would matter.
> >
> >
> Can alleviate it to some extent by driving a pair of such phase
> detectors so that their outputs are in quadrature.
> One just selects the phase detector output that is in the linear range.
> The quadrature outputs also allow unambiguous assignment of the sign of
> any phase change.
The Symmetricom 5120A does something very clever to alleviate this
problem. Explained in US patent 7,227,346 and "Direct-Digital Phase-Noise
Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger, W. Solbrig, and
S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics, and
Frequency Control Joint 50th Anniversary Conference, pages 287-291.
Joe
JM
Joseph M Gwinn
Thu, Dec 11, 2008 12:52 AM
Joe
Another limitation on using too low a beat frequency is imposed by the
increasing equivalent input noise spectral density of the sound card as
the frequency decreases.
Yes, although people made good use of 1 Hz in DMTD instruments despite the
junk near DC.
Joe
Bruce,
time-nuts-bounces@febo.com wrote on 12/10/2008 04:46:50 PM:
> Joe
>
> Another limitation on using too low a beat frequency is imposed by the
> increasing equivalent input noise spectral density of the sound card as
> the frequency decreases.
Yes, although people made good use of 1 Hz in DMTD instruments despite the
junk near DC.
Joe
BG
Bruce Griffiths
Thu, Dec 11, 2008 12:58 AM
Joe
Another limitation on using too low a beat frequency is imposed by the
increasing equivalent input noise spectral density of the sound card as
the frequency decreases.
Yes, although people made good use of 1 Hz in DMTD instruments despite the
junk near DC.
Joe
Joe
The low frequency noise of the components used in their unoptimised
slope amplifiers is significantly smaller than that of a sound card.
A Collin's style optimised slope amplifier limiter may be useful for use
with a sound card (particularly the lower resolution ones) if one is
timetagging the beat frequency zero crossings.
Less slope gain will be required than when driving a counter input.
In the flicker noise region choice of beat frequency is relatively non
critical.
Bruce
Joseph M Gwinn wrote:
> Bruce,
>
>
> time-nuts-bounces@febo.com wrote on 12/10/2008 04:46:50 PM:
>
>
>> Joe
>>
>> Another limitation on using too low a beat frequency is imposed by the
>> increasing equivalent input noise spectral density of the sound card as
>> the frequency decreases.
>>
>
> Yes, although people made good use of 1 Hz in DMTD instruments despite the
> junk near DC.
>
>
> Joe
>
Joe
The low frequency noise of the components used in their unoptimised
slope amplifiers is significantly smaller than that of a sound card.
A Collin's style optimised slope amplifier limiter may be useful for use
with a sound card (particularly the lower resolution ones) if one is
timetagging the beat frequency zero crossings.
Less slope gain will be required than when driving a counter input.
In the flicker noise region choice of beat frequency is relatively non
critical.
Bruce
JM
Joseph M Gwinn
Thu, Dec 11, 2008 1:15 AM
Joe
Another limitation on using too low a beat frequency is imposed by
increasing equivalent input noise spectral density of the sound card
Yes, although people made good use of 1 Hz in DMTD instruments despite
Joe
The low frequency noise of the components used in their unoptimised
slope amplifiers is significantly smaller than that of a sound card.
Who is "their"? But I would hazard that people designing for operation at
1 Hz chose components with low flicker noise, to the extent then possible.
A Collin's style optimised slope amplifier limiter may be useful for use
with a sound card (particularly the lower resolution ones) if one is
timetagging the beat frequency zero crossings.
Less slope gain will be required than when driving a counter input.
In the flicker noise region choice of beat frequency is relatively non
critical.
With the vast amount of data available from a soundcard, I'd be tempted to
do a least squares three-parameter fit of a sine wave to the (decimated)
measured data and use only the resulting fitted parameters in subsequent
computations. Like estimating where the zero crossings are. Then the
noise averaging is over the entire fitting window, not just near zero
crossings. (I have been following the talk of using a sin x over x
fitting function, but I don't know if this really works any better than a
simple fit to a sine, given this much data.)
Joe
Bruce,
time-nuts-bounces@febo.com wrote on 12/10/2008 07:58:49 PM:
> Joseph M Gwinn wrote:
> > Bruce,
> >
> >
> > time-nuts-bounces@febo.com wrote on 12/10/2008 04:46:50 PM:
> >
> >
> >> Joe
> >>
> >> Another limitation on using too low a beat frequency is imposed by
the
> >> increasing equivalent input noise spectral density of the sound card
as
> >> the frequency decreases.
> >>
> >
> > Yes, although people made good use of 1 Hz in DMTD instruments despite
the
> > junk near DC.
> >
> >
> > Joe
> >
> Joe
>
> The low frequency noise of the components used in their unoptimised
> slope amplifiers is significantly smaller than that of a sound card.
Who is "their"? But I would hazard that people designing for operation at
1 Hz chose components with low flicker noise, to the extent then possible.
> A Collin's style optimised slope amplifier limiter may be useful for use
> with a sound card (particularly the lower resolution ones) if one is
> timetagging the beat frequency zero crossings.
> Less slope gain will be required than when driving a counter input.
> In the flicker noise region choice of beat frequency is relatively non
> critical.
With the vast amount of data available from a soundcard, I'd be tempted to
do a least squares three-parameter fit of a sine wave to the (decimated)
measured data and use only the resulting fitted parameters in subsequent
computations. Like estimating where the zero crossings are. Then the
noise averaging is over the entire fitting window, not just near zero
crossings. (I have been following the talk of using a sin x over x
fitting function, but I don't know if this really works any better than a
simple fit to a sine, given this much data.)
Joe
BG
Bruce Griffiths
Thu, Dec 11, 2008 1:38 AM
Joe
Joseph M Gwinn wrote:
Reflecting the sum frequency back into the mixer is actually necessary
to reduce the noise at the IF port.
I believe that one of Agilent's simulation application notes mentions
this effect but I don't recall the actual application note number.
This will affect the mixer RF and IF port impedance so adding a series
resistor may be required to improve the SWR.
How big an effect is this? Is the absolute noise decreased, or does it
remain the same while the signal increase?
With the same difference frequency IF port termination impedance, noise
is actually decreased along with the mixer conversion loss.
However if the sound card input noise dominates reducing the mixer
effective output noise wont help.
If I'm understanding Walls and Stein (paper 112) correctly, the advantage
is because with the capacitor load the beatnote waveform approaches
square, thus increasing the zero-crossing speed and therefor the phase
sensitivity. This is no doubt true, but the question was if this also
caused a small everything-dependent phase shift, something that would not
have mattered in the measurement of phase noise. The object of paper 112
was to remedy a 10 to 20 dB error in phase noise measurements. The
critical words are in the lower left column of page 337, in the paragraph
beginning "If the mixer is terminated ...".
Saturating the RF port has a similar effect.
If one is time stamping the zero crossings an increased zero crossing
slope is an advantage.
For relative phase measurements a trapezoidal beat frequency waveform
may be less useful.
MiniCircuits AN-41-001 "FAQ about Phase Detectors" has on page 2 a 500
resistor to ground and a 5000 ohm resistor to the first filter
so the capacitor is isolated from the IF port by the resistors.
I wouldn't take too much notice of that recommendation as I have little
confidence in the author's experience/knowledge.
Well, OK, but:
Stephen Kurtz says the same thing on the third column of the third page, a
bit above Figure 6.
Off course with a capacitive IF port termination matching the RF and LO
ports becomes more critical as does the reverse isolation of the various
amplifiers driving the RF and LO ports.
It may be simpler in fact to use a level 17 mixer with high LO to RF and
LO to IF isolation with the RF port unsaturated as it relaxes the
reverse isolation specs for the isolation amplifiers.
Nelson and Walls (paper 971), Figure 4, also shows the low pass filter
arranged to absorb the sum signal, not allowing it to be reflected back
into the mixer.
Supposedly an SRA-1, but some caution is in order as some
to the effect of the input offset of an opamp based IF preamp in the
same application note were of dubious veracity unless one
inverting opamp input stage.
This issue was mentioned in another app note, but their main issue
appeared to be that the opamp bias currents could cause an offset.
But the circuit they suggest has no effect on bias current induced
offset, the same current flows into the mixer and termination impedance
independent of the series resistance.
You're right that the proposed remedy didn't make sense. I don't know
that this is a big problem with modern opamps, especially FET input ones
(if needed).
The only configuration for which it makes any sense is an inverting
input amplifier with a finite input voltage offset.
Yes, I should have said that when the 2 input signals are in quadrature,
any capacitive crosstalk will have little effect on the phase shift.
Ah. Because the capacitor coupling adds a second 90 degree shift,
bringing the total to 180 degrees.
But crosstalk by ground coupling will be unaffected. As will crosstalk by
transformer action. Those boards are pretty crowded.
Yes its better to measure it rather than relying too much on conjecture.
The AP192 has a somewhat higher interchannel isolation than that, the
interchannel crosstalk spec is about -120dB.
With a sufficiently large number of samples the its easy to see
artifacts as low as -140dBFS.
Yep. Seems like a very good card.
It's hard to find such Firewire systems without such unnecessary frills
(for this application) as high gain preamps.
The AP192 has high-level inputs, but I don't know if this bypasses the
preamps, or attenuates. Given their target market, I'd bet it bypasses.
There are no preamps other than an external differential input amplifier
that translates the 4 Vrms FS inputs at the input connector to a level
that the ADC can handle.
The ADC chip itself has no preamps built in.
There have been numerous complaint about this by some audio nuts,
however for this application not having such amplifiers is ideal.
The gain tempco and linearity of some variable gain audio preamps is
somewhat suspect.
I would think that none of these cards has a good tempco of anything,
given the lack of necessity in their market.
I would think that linearity would be quite good, given the horsepower
competitions on linearity.
Since the 2 ADCs share the same reference their gain tracking tempco
should be quite good given that they use capacitors rather than
resistors within the ADCs.
Other cards using AKM 24 bit ADCs should also be suitable.
Thanks. I'll look into their data.
20 Log[ 2^24 ] = 144 dB, so something else will be the limit.
Actual ENOB ~ 19 to 20 bits.
Makes sense. 20 Log [ 2^19 ] = 114 dB. Still plenty good enough.
Ideally an external sound card with balanced XLR inputs would be
HP produced a number of different phase comparators each with a
different type of phase detector.
OK. And the PLL folk must have a million designs.
Can alleviate it to some extent by driving a pair of such phase
detectors so that their outputs are in quadrature.
One just selects the phase detector output that is in the linear range.
The quadrature outputs also allow unambiguous assignment of the sign of
any phase change.
The Symmetricom 5120A does something very clever to alleviate this
problem. Explained in US patent 7,227,346 and "Direct-Digital Phase-Noise
Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger, W. Solbrig, and
S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics, and
Frequency Control Joint 50th Anniversary Conference, pages 287-291.
Joe
I've read the patent.
Bruce
Joe
Joseph M Gwinn wrote:
> Bruce,
>
>
>> Reflecting the sum frequency back into the mixer is actually necessary
>> to reduce the noise at the IF port.
>> I believe that one of Agilent's simulation application notes mentions
>> this effect but I don't recall the actual application note number.
>> This will affect the mixer RF and IF port impedance so adding a series
>> resistor may be required to improve the SWR.
>>
>
> How big an effect is this? Is the absolute noise decreased, or does it
> remain the same while the signal increase?
>
>
With the same difference frequency IF port termination impedance, noise
is actually decreased along with the mixer conversion loss.
However if the sound card input noise dominates reducing the mixer
effective output noise wont help.
> If I'm understanding Walls and Stein (paper 112) correctly, the advantage
> is because with the capacitor load the beatnote waveform approaches
> square, thus increasing the zero-crossing speed and therefor the phase
> sensitivity. This is no doubt true, but the question was if this also
> caused a small everything-dependent phase shift, something that would not
> have mattered in the measurement of phase noise. The object of paper 112
> was to remedy a 10 to 20 dB error in phase noise measurements. The
> critical words are in the lower left column of page 337, in the paragraph
> beginning "If the mixer is terminated ...".
>
>
>
Saturating the RF port has a similar effect.
If one is time stamping the zero crossings an increased zero crossing
slope is an advantage.
For relative phase measurements a trapezoidal beat frequency waveform
may be less useful.
>>> MiniCircuits AN-41-001 "FAQ about Phase Detectors" has on page 2 a 500
>>>
> ohm
>
>>> resistor to ground and a 5000 ohm resistor to the first filter
>>>
> capacitor,
>
>>> so the capacitor is isolated from the IF port by the resistors.
>>>
>>>
>>>
>> I wouldn't take too much notice of that recommendation as I have little
>> confidence in the author's experience/knowledge.
>>
>
> Well, OK, but:
>
> Stephen Kurtz says the same thing on the third column of the third page, a
> bit above Figure 6.
>
Off course with a capacitive IF port termination matching the RF and LO
ports becomes more critical as does the reverse isolation of the various
amplifiers driving the RF and LO ports.
It may be simpler in fact to use a level 17 mixer with high LO to RF and
LO to IF isolation with the RF port unsaturated as it relaxes the
reverse isolation specs for the isolation amplifiers.
> Nelson and Walls (paper 971), Figure 4, also shows the low pass filter
> arranged to absorb the sum signal, not allowing it to be reflected back
> into the mixer.
>
>
>
>>>> Supposedly an SRA-1, but some caution is in order as some
>>>>
>> statements as
>>
>>>> to the effect of the input offset of an opamp based IF preamp in the
>>>> same application note were of dubious veracity unless one
>>>>
>> were to use an
>>
>>>> inverting opamp input stage.
>>>>
>>>>
>>> This issue was mentioned in another app note, but their main issue
>>> appeared to be that the opamp bias currents could cause an offset.
>>>
>>>
>>>
>>>
>> But the circuit they suggest has no effect on bias current induced
>> offset, the same current flows into the mixer and termination impedance
>> independent of the series resistance.
>>
>
> You're right that the proposed remedy didn't make sense. I don't know
> that this is a big problem with modern opamps, especially FET input ones
> (if needed).
>
>
The only configuration for which it makes any sense is an inverting
input amplifier with a finite input voltage offset.
>> Yes, I should have said that when the 2 input signals are in quadrature,
>> any capacitive crosstalk will have little effect on the phase shift.
>>
>
> Ah. Because the capacitor coupling adds a second 90 degree shift,
> bringing the total to 180 degrees.
>
> But crosstalk by ground coupling will be unaffected. As will crosstalk by
> transformer action. Those boards are pretty crowded.
>
>
>
Yes its better to measure it rather than relying too much on conjecture.
>> The AP192 has a somewhat higher interchannel isolation than that, the
>> interchannel crosstalk spec is about -120dB.
>> With a sufficiently large number of samples the its easy to see
>> artifacts as low as -140dBFS.
>>
>
> Yep. Seems like a very good card.
>
>
>
>
>
>> It's hard to find such Firewire systems without such unnecessary frills
>> (for this application) as high gain preamps.
>>
>
> The AP192 has high-level inputs, but I don't know if this bypasses the
> preamps, or attenuates. Given their target market, I'd bet it bypasses.
>
>
There are no preamps other than an external differential input amplifier
that translates the 4 Vrms FS inputs at the input connector to a level
that the ADC can handle.
The ADC chip itself has no preamps built in.
There have been numerous complaint about this by some audio nuts,
however for this application not having such amplifiers is ideal.
>
>> The gain tempco and linearity of some variable gain audio preamps is
>> somewhat suspect.
>>
>
> I would think that none of these cards has a good tempco of anything,
> given the lack of necessity in their market.
>
> I would think that linearity would be quite good, given the horsepower
> competitions on linearity.
>
>
Since the 2 ADCs share the same reference their gain tracking tempco
should be quite good given that they use capacitors rather than
resistors within the ADCs.
>
>>>> Other cards using AKM 24 bit ADCs should also be suitable.
>>>>
>>>>
>>> Who is AKM?
>>>
>>>
>>>
>> Asahi Kasei EKM
>>
>> http://www.asahi-kasei.co.jp/akm/en/
>>
>> http://www.asahi-kasei.co.jp/akm/en/product/proaudio.html
>>
>>
>
> Thanks. I'll look into their data.
>
>
>
>>> 20 Log[ 2^24 ] = 144 dB, so something else will be the limit.
>>>
>>>
>>>
>>>
>> Actual ENOB ~ 19 to 20 bits.
>>
>
> Makes sense. 20 Log [ 2^19 ] = 114 dB. Still plenty good enough.
>
>
>
>>>> Ideally an external sound card with balanced XLR inputs would be
>>>>
> best.
>
> Yes.
>
>
>
>
>>>> HP produced a number of different phase comparators each with a
>>>> different type of phase detector.
>>>>
>
> OK. And the PLL folk must have a million designs.
>
>
>
>>
>> Can alleviate it to some extent by driving a pair of such phase
>> detectors so that their outputs are in quadrature.
>> One just selects the phase detector output that is in the linear range.
>> The quadrature outputs also allow unambiguous assignment of the sign of
>> any phase change.
>>
>
> The Symmetricom 5120A does something very clever to alleviate this
> problem. Explained in US patent 7,227,346 and "Direct-Digital Phase-Noise
> Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger, W. Solbrig, and
> S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics, and
> Frequency Control Joint 50th Anniversary Conference, pages 287-291.
>
> Joe
>
>
I've read the patent.
Bruce
BG
Bruce Griffiths
Thu, Dec 11, 2008 1:50 AM
Bruce,
The low frequency noise of the components used in their unoptimised
slope amplifiers is significantly smaller than that of a sound card.
Who is "their"? But I would hazard that people designing for operation at
1 Hz chose components with low flicker noise, to the extent then possible.
JPL and others who repeated the same assumptions..
The amplifier choices were reasonably appropriate, however the
distribution of gain and filter time constants is not optimum.
At least they weren't as bad as their early isolation amplifier designs,
but no one seemed to know how to design low phase noise amplifiers at
that time, although at the time of the more recent and somewhat better
isolation amplifier design, they ought to have.
A Collin's style optimised slope amplifier limiter may be useful for use
with a sound card (particularly the lower resolution ones) if one is
timetagging the beat frequency zero crossings.
Less slope gain will be required than when driving a counter input.
In the flicker noise region choice of beat frequency is relatively non
critical.
With the vast amount of data available from a soundcard, I'd be tempted to
do a least squares three-parameter fit of a sine wave to the (decimated)
measured data and use only the resulting fitted parameters in subsequent
computations. Like estimating where the zero crossings are. Then the
noise averaging is over the entire fitting window, not just near zero
crossings. (I have been following the talk of using a sin x over x
fitting function, but I don't know if this really works any better than a
simple fit to a sine, given this much data.)
Joe
Even a crude linear interpolation to locate the zero crossing shouldnt
be too bad.
The phase or zero crossing location of the fitted sine wave will be
relatively insensitive to noise near the beat frequency signal peaks.
If you save the sound card samples in a file, you (and others) can then
analyse the same data using different methods.
Bruce
Joseph M Gwinn wrote:
> Bruce,
>
>
>
> The low frequency noise of the components used in their unoptimised
> slope amplifiers is significantly smaller than that of a sound card.
>
>
> Who is "their"? But I would hazard that people designing for operation at
> 1 Hz chose components with low flicker noise, to the extent then possible.
>
>
JPL and others who repeated the same assumptions..
The amplifier choices were reasonably appropriate, however the
distribution of gain and filter time constants is not optimum.
At least they weren't as bad as their early isolation amplifier designs,
but no one seemed to know how to design low phase noise amplifiers at
that time, although at the time of the more recent and somewhat better
isolation amplifier design, they ought to have.
>
>> A Collin's style optimised slope amplifier limiter may be useful for use
>> with a sound card (particularly the lower resolution ones) if one is
>> timetagging the beat frequency zero crossings.
>> Less slope gain will be required than when driving a counter input.
>> In the flicker noise region choice of beat frequency is relatively non
>> critical.
>>
>
> With the vast amount of data available from a soundcard, I'd be tempted to
> do a least squares three-parameter fit of a sine wave to the (decimated)
> measured data and use only the resulting fitted parameters in subsequent
> computations. Like estimating where the zero crossings are. Then the
> noise averaging is over the entire fitting window, not just near zero
> crossings. (I have been following the talk of using a sin x over x
> fitting function, but I don't know if this really works any better than a
> simple fit to a sine, given this much data.)
>
> Joe
>
>
Even a crude linear interpolation to locate the zero crossing shouldnt
be too bad.
The phase or zero crossing location of the fitted sine wave will be
relatively insensitive to noise near the beat frequency signal peaks.
If you save the sound card samples in a file, you (and others) can then
analyse the same data using different methods.
Bruce
MF
Mike Feher
Thu, Dec 11, 2008 2:05 AM
I am now, and actually have been, at the point where I just do not read
bottom line post/replies. Bruce has a lot of good information to share,
but, now, if I click on a post, and do not immediately see a response it
is just deleted. Maybe it will be my loss, but, technology as well as
the internet is evolving, and bottom line replies totally suck. - Mike
Mike B. Feher
EOZ Inc.
89 Arnold Blvd.
Howell, NJ, 07731
732-886-5960
908-902-3831 - cell
-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com] On
Behalf Of Bruce Griffiths
Sent: Wednesday, December 10, 2008 8:38 PM
To: Discussion of precise time and frequency measurement
Subject: Re: [time-nuts] Sub Pico Second Phase logger
Joe
Joseph M Gwinn wrote:
Reflecting the sum frequency back into the mixer is actually
to reduce the noise at the IF port.
I believe that one of Agilent's simulation application notes mentions
this effect but I don't recall the actual application note number.
This will affect the mixer RF and IF port impedance so adding a
resistor may be required to improve the SWR.
How big an effect is this? Is the absolute noise decreased, or does
remain the same while the signal increase?
With the same difference frequency IF port termination impedance, noise
is actually decreased along with the mixer conversion loss.
However if the sound card input noise dominates reducing the mixer
effective output noise wont help.
If I'm understanding Walls and Stein (paper 112) correctly, the
is because with the capacitor load the beatnote waveform approaches
square, thus increasing the zero-crossing speed and therefor the phase
sensitivity. This is no doubt true, but the question was if this also
caused a small everything-dependent phase shift, something that would
have mattered in the measurement of phase noise. The object of paper
was to remedy a 10 to 20 dB error in phase noise measurements. The
critical words are in the lower left column of page 337, in the
beginning "If the mixer is terminated ...".
Saturating the RF port has a similar effect.
If one is time stamping the zero crossings an increased zero crossing
slope is an advantage.
For relative phase measurements a trapezoidal beat frequency waveform
may be less useful.
MiniCircuits AN-41-001 "FAQ about Phase Detectors" has on page 2 a
resistor to ground and a 5000 ohm resistor to the first filter
so the capacitor is isolated from the IF port by the resistors.
I wouldn't take too much notice of that recommendation as I have
confidence in the author's experience/knowledge.
Well, OK, but:
Stephen Kurtz says the same thing on the third column of the third
Off course with a capacitive IF port termination matching the RF and LO
ports becomes more critical as does the reverse isolation of the various
amplifiers driving the RF and LO ports.
It may be simpler in fact to use a level 17 mixer with high LO to RF and
LO to IF isolation with the RF port unsaturated as it relaxes the
reverse isolation specs for the isolation amplifiers.
Nelson and Walls (paper 971), Figure 4, also shows the low pass filter
arranged to absorb the sum signal, not allowing it to be reflected
Supposedly an SRA-1, but some caution is in order as some
to the effect of the input offset of an opamp based IF preamp in
same application note were of dubious veracity unless one
inverting opamp input stage.
This issue was mentioned in another app note, but their main issue
appeared to be that the opamp bias currents could cause an offset.
But the circuit they suggest has no effect on bias current induced
offset, the same current flows into the mixer and termination
independent of the series resistance.
You're right that the proposed remedy didn't make sense. I don't know
that this is a big problem with modern opamps, especially FET input
The only configuration for which it makes any sense is an inverting
input amplifier with a finite input voltage offset.
Yes, I should have said that when the 2 input signals are in
any capacitive crosstalk will have little effect on the phase shift.
Ah. Because the capacitor coupling adds a second 90 degree shift,
bringing the total to 180 degrees.
But crosstalk by ground coupling will be unaffected. As will
transformer action. Those boards are pretty crowded.
Yes its better to measure it rather than relying too much on conjecture.
The AP192 has a somewhat higher interchannel isolation than that, the
interchannel crosstalk spec is about -120dB.
With a sufficiently large number of samples the its easy to see
artifacts as low as -140dBFS.
Yep. Seems like a very good card.
It's hard to find such Firewire systems without such unnecessary
(for this application) as high gain preamps.
The AP192 has high-level inputs, but I don't know if this bypasses the
preamps, or attenuates. Given their target market, I'd bet it
There are no preamps other than an external differential input amplifier
that translates the 4 Vrms FS inputs at the input connector to a level
that the ADC can handle.
The ADC chip itself has no preamps built in.
There have been numerous complaint about this by some audio nuts,
however for this application not having such amplifiers is ideal.
The gain tempco and linearity of some variable gain audio preamps is
somewhat suspect.
I would think that none of these cards has a good tempco of anything,
given the lack of necessity in their market.
I would think that linearity would be quite good, given the horsepower
competitions on linearity.
Since the 2 ADCs share the same reference their gain tracking tempco
should be quite good given that they use capacitors rather than
resistors within the ADCs.
Other cards using AKM 24 bit ADCs should also be suitable.
Thanks. I'll look into their data.
20 Log[ 2^24 ] = 144 dB, so something else will be the limit.
Actual ENOB ~ 19 to 20 bits.
Makes sense. 20 Log [ 2^19 ] = 114 dB. Still plenty good enough.
Ideally an external sound card with balanced XLR inputs would be
HP produced a number of different phase comparators each with a
different type of phase detector.
OK. And the PLL folk must have a million designs.
Can alleviate it to some extent by driving a pair of such phase
detectors so that their outputs are in quadrature.
One just selects the phase detector output that is in the linear
The quadrature outputs also allow unambiguous assignment of the sign
The Symmetricom 5120A does something very clever to alleviate this
problem. Explained in US patent 7,227,346 and "Direct-Digital
Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger, W. Solbrig,
S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics, and
Frequency Control Joint 50th Anniversary Conference, pages 287-291.
Joe
I am now, and actually have been, at the point where I just do not read
bottom line post/replies. Bruce has a lot of good information to share,
but, now, if I click on a post, and do not immediately see a response it
is just deleted. Maybe it will be my loss, but, technology as well as
the internet is evolving, and bottom line replies totally suck. - Mike
Mike B. Feher
EOZ Inc.
89 Arnold Blvd.
Howell, NJ, 07731
732-886-5960
908-902-3831 - cell
-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com] On
Behalf Of Bruce Griffiths
Sent: Wednesday, December 10, 2008 8:38 PM
To: Discussion of precise time and frequency measurement
Subject: Re: [time-nuts] Sub Pico Second Phase logger
Joe
Joseph M Gwinn wrote:
> Bruce,
>
>
>> Reflecting the sum frequency back into the mixer is actually
necessary
>> to reduce the noise at the IF port.
>> I believe that one of Agilent's simulation application notes mentions
>> this effect but I don't recall the actual application note number.
>> This will affect the mixer RF and IF port impedance so adding a
series
>> resistor may be required to improve the SWR.
>>
>
> How big an effect is this? Is the absolute noise decreased, or does
it
> remain the same while the signal increase?
>
>
With the same difference frequency IF port termination impedance, noise
is actually decreased along with the mixer conversion loss.
However if the sound card input noise dominates reducing the mixer
effective output noise wont help.
> If I'm understanding Walls and Stein (paper 112) correctly, the
advantage
> is because with the capacitor load the beatnote waveform approaches
> square, thus increasing the zero-crossing speed and therefor the phase
> sensitivity. This is no doubt true, but the question was if this also
> caused a small everything-dependent phase shift, something that would
not
> have mattered in the measurement of phase noise. The object of paper
112
> was to remedy a 10 to 20 dB error in phase noise measurements. The
> critical words are in the lower left column of page 337, in the
paragraph
> beginning "If the mixer is terminated ...".
>
>
>
Saturating the RF port has a similar effect.
If one is time stamping the zero crossings an increased zero crossing
slope is an advantage.
For relative phase measurements a trapezoidal beat frequency waveform
may be less useful.
>>> MiniCircuits AN-41-001 "FAQ about Phase Detectors" has on page 2 a
500
>>>
> ohm
>
>>> resistor to ground and a 5000 ohm resistor to the first filter
>>>
> capacitor,
>
>>> so the capacitor is isolated from the IF port by the resistors.
>>>
>>>
>>>
>> I wouldn't take too much notice of that recommendation as I have
little
>> confidence in the author's experience/knowledge.
>>
>
> Well, OK, but:
>
> Stephen Kurtz says the same thing on the third column of the third
page, a
> bit above Figure 6.
>
Off course with a capacitive IF port termination matching the RF and LO
ports becomes more critical as does the reverse isolation of the various
amplifiers driving the RF and LO ports.
It may be simpler in fact to use a level 17 mixer with high LO to RF and
LO to IF isolation with the RF port unsaturated as it relaxes the
reverse isolation specs for the isolation amplifiers.
> Nelson and Walls (paper 971), Figure 4, also shows the low pass filter
> arranged to absorb the sum signal, not allowing it to be reflected
back
> into the mixer.
>
>
>
>>>> Supposedly an SRA-1, but some caution is in order as some
>>>>
>> statements as
>>
>>>> to the effect of the input offset of an opamp based IF preamp in
the
>>>> same application note were of dubious veracity unless one
>>>>
>> were to use an
>>
>>>> inverting opamp input stage.
>>>>
>>>>
>>> This issue was mentioned in another app note, but their main issue
>>> appeared to be that the opamp bias currents could cause an offset.
>>>
>>>
>>>
>>>
>> But the circuit they suggest has no effect on bias current induced
>> offset, the same current flows into the mixer and termination
impedance
>> independent of the series resistance.
>>
>
> You're right that the proposed remedy didn't make sense. I don't know
> that this is a big problem with modern opamps, especially FET input
ones
> (if needed).
>
>
The only configuration for which it makes any sense is an inverting
input amplifier with a finite input voltage offset.
>> Yes, I should have said that when the 2 input signals are in
quadrature,
>> any capacitive crosstalk will have little effect on the phase shift.
>>
>
> Ah. Because the capacitor coupling adds a second 90 degree shift,
> bringing the total to 180 degrees.
>
> But crosstalk by ground coupling will be unaffected. As will
crosstalk by
> transformer action. Those boards are pretty crowded.
>
>
>
Yes its better to measure it rather than relying too much on conjecture.
>> The AP192 has a somewhat higher interchannel isolation than that, the
>> interchannel crosstalk spec is about -120dB.
>> With a sufficiently large number of samples the its easy to see
>> artifacts as low as -140dBFS.
>>
>
> Yep. Seems like a very good card.
>
>
>
>
>
>> It's hard to find such Firewire systems without such unnecessary
frills
>> (for this application) as high gain preamps.
>>
>
> The AP192 has high-level inputs, but I don't know if this bypasses the
> preamps, or attenuates. Given their target market, I'd bet it
bypasses.
>
>
There are no preamps other than an external differential input amplifier
that translates the 4 Vrms FS inputs at the input connector to a level
that the ADC can handle.
The ADC chip itself has no preamps built in.
There have been numerous complaint about this by some audio nuts,
however for this application not having such amplifiers is ideal.
>
>> The gain tempco and linearity of some variable gain audio preamps is
>> somewhat suspect.
>>
>
> I would think that none of these cards has a good tempco of anything,
> given the lack of necessity in their market.
>
> I would think that linearity would be quite good, given the horsepower
> competitions on linearity.
>
>
Since the 2 ADCs share the same reference their gain tracking tempco
should be quite good given that they use capacitors rather than
resistors within the ADCs.
>
>>>> Other cards using AKM 24 bit ADCs should also be suitable.
>>>>
>>>>
>>> Who is AKM?
>>>
>>>
>>>
>> Asahi Kasei EKM
>>
>> http://www.asahi-kasei.co.jp/akm/en/
>>
>> http://www.asahi-kasei.co.jp/akm/en/product/proaudio.html
>>
>>
>
> Thanks. I'll look into their data.
>
>
>
>>> 20 Log[ 2^24 ] = 144 dB, so something else will be the limit.
>>>
>>>
>>>
>>>
>> Actual ENOB ~ 19 to 20 bits.
>>
>
> Makes sense. 20 Log [ 2^19 ] = 114 dB. Still plenty good enough.
>
>
>
>>>> Ideally an external sound card with balanced XLR inputs would be
>>>>
> best.
>
> Yes.
>
>
>
>
>>>> HP produced a number of different phase comparators each with a
>>>> different type of phase detector.
>>>>
>
> OK. And the PLL folk must have a million designs.
>
>
>
>>
>> Can alleviate it to some extent by driving a pair of such phase
>> detectors so that their outputs are in quadrature.
>> One just selects the phase detector output that is in the linear
range.
>> The quadrature outputs also allow unambiguous assignment of the sign
of
>> any phase change.
>>
>
> The Symmetricom 5120A does something very clever to alleviate this
> problem. Explained in US patent 7,227,346 and "Direct-Digital
Phase-Noise
> Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger, W. Solbrig,
and
> S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics, and
> Frequency Control Joint 50th Anniversary Conference, pages 287-291.
>
> Joe
>
>
I've read the patent.
Bruce
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JM
Joseph M Gwinn
Thu, Dec 11, 2008 2:27 AM
Let's have a top-posting versus bottom-posting fight!
But we're too late, it's already been done:
http://en.wikipedia.org/wiki/Posting_style
http://allmyfaqs.net/faq.pl?Top-posting_or_bottom-posting
And so on. Many times.
Joe
"Mike Feher" mfeher@eozinc.com
Sent by: time-nuts-bounces@febo.com
12/10/2008 09:06 PM
Please respond to
Discussion of precise time and frequency measurement time-nuts@febo.com
To
"'Discussion of precise time and frequency measurement'"
time-nuts@febo.com
cc
Subject
Re: [time-nuts] Sub Pico Second Phase logger
I am now, and actually have been, at the point where I just do not read
bottom line post/replies. Bruce has a lot of good information to share,
but, now, if I click on a post, and do not immediately see a response it
is just deleted. Maybe it will be my loss, but, technology as well as
the internet is evolving, and bottom line replies totally suck. - Mike
Mike B. Feher
EOZ Inc.
89 Arnold Blvd.
Howell, NJ, 07731
732-886-5960
908-902-3831 - cell
-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com] On
Behalf Of Bruce Griffiths
Sent: Wednesday, December 10, 2008 8:38 PM
To: Discussion of precise time and frequency measurement
Subject: Re: [time-nuts] Sub Pico Second Phase logger
Joe
Joseph M Gwinn wrote:
Reflecting the sum frequency back into the mixer is actually
to reduce the noise at the IF port.
I believe that one of Agilent's simulation application notes mentions
this effect but I don't recall the actual application note number.
This will affect the mixer RF and IF port impedance so adding a
resistor may be required to improve the SWR.
How big an effect is this? Is the absolute noise decreased, or does
remain the same while the signal increase?
With the same difference frequency IF port termination impedance, noise
is actually decreased along with the mixer conversion loss.
However if the sound card input noise dominates reducing the mixer
effective output noise wont help.
If I'm understanding Walls and Stein (paper 112) correctly, the
is because with the capacitor load the beatnote waveform approaches
square, thus increasing the zero-crossing speed and therefor the phase
sensitivity. This is no doubt true, but the question was if this also
caused a small everything-dependent phase shift, something that would
have mattered in the measurement of phase noise. The object of paper
was to remedy a 10 to 20 dB error in phase noise measurements. The
critical words are in the lower left column of page 337, in the
beginning "If the mixer is terminated ...".
Saturating the RF port has a similar effect.
If one is time stamping the zero crossings an increased zero crossing
slope is an advantage.
For relative phase measurements a trapezoidal beat frequency waveform
may be less useful.
MiniCircuits AN-41-001 "FAQ about Phase Detectors" has on page 2 a
resistor to ground and a 5000 ohm resistor to the first filter
so the capacitor is isolated from the IF port by the resistors.
I wouldn't take too much notice of that recommendation as I have
confidence in the author's experience/knowledge.
Well, OK, but:
Stephen Kurtz says the same thing on the third column of the third
Off course with a capacitive IF port termination matching the RF and LO
ports becomes more critical as does the reverse isolation of the various
amplifiers driving the RF and LO ports.
It may be simpler in fact to use a level 17 mixer with high LO to RF and
LO to IF isolation with the RF port unsaturated as it relaxes the
reverse isolation specs for the isolation amplifiers.
Nelson and Walls (paper 971), Figure 4, also shows the low pass filter
arranged to absorb the sum signal, not allowing it to be reflected
Supposedly an SRA-1, but some caution is in order as some
to the effect of the input offset of an opamp based IF preamp in
same application note were of dubious veracity unless one
inverting opamp input stage.
This issue was mentioned in another app note, but their main issue
appeared to be that the opamp bias currents could cause an offset.
But the circuit they suggest has no effect on bias current induced
offset, the same current flows into the mixer and termination
independent of the series resistance.
You're right that the proposed remedy didn't make sense. I don't know
that this is a big problem with modern opamps, especially FET input
The only configuration for which it makes any sense is an inverting
input amplifier with a finite input voltage offset.
Yes, I should have said that when the 2 input signals are in
any capacitive crosstalk will have little effect on the phase shift.
Ah. Because the capacitor coupling adds a second 90 degree shift,
bringing the total to 180 degrees.
But crosstalk by ground coupling will be unaffected. As will
transformer action. Those boards are pretty crowded.
Yes its better to measure it rather than relying too much on conjecture.
The AP192 has a somewhat higher interchannel isolation than that, the
interchannel crosstalk spec is about -120dB.
With a sufficiently large number of samples the its easy to see
artifacts as low as -140dBFS.
Yep. Seems like a very good card.
It's hard to find such Firewire systems without such unnecessary
(for this application) as high gain preamps.
The AP192 has high-level inputs, but I don't know if this bypasses the
preamps, or attenuates. Given their target market, I'd bet it
There are no preamps other than an external differential input amplifier
that translates the 4 Vrms FS inputs at the input connector to a level
that the ADC can handle.
The ADC chip itself has no preamps built in.
There have been numerous complaint about this by some audio nuts,
however for this application not having such amplifiers is ideal.
The gain tempco and linearity of some variable gain audio preamps is
somewhat suspect.
I would think that none of these cards has a good tempco of anything,
given the lack of necessity in their market.
I would think that linearity would be quite good, given the horsepower
competitions on linearity.
Since the 2 ADCs share the same reference their gain tracking tempco
should be quite good given that they use capacitors rather than
resistors within the ADCs.
Other cards using AKM 24 bit ADCs should also be suitable.
Thanks. I'll look into their data.
20 Log[ 2^24 ] = 144 dB, so something else will be the limit.
Actual ENOB ~ 19 to 20 bits.
Makes sense. 20 Log [ 2^19 ] = 114 dB. Still plenty good enough.
Ideally an external sound card with balanced XLR inputs would be
HP produced a number of different phase comparators each with a
different type of phase detector.
OK. And the PLL folk must have a million designs.
Can alleviate it to some extent by driving a pair of such phase
detectors so that their outputs are in quadrature.
One just selects the phase detector output that is in the linear
The quadrature outputs also allow unambiguous assignment of the sign
The Symmetricom 5120A does something very clever to alleviate this
problem. Explained in US patent 7,227,346 and "Direct-Digital
Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger, W. Solbrig,
S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics, and
Frequency Control Joint 50th Anniversary Conference, pages 287-291.
Joe
Let's have a top-posting versus bottom-posting fight!
But we're too late, it's already been done:
<http://en.wikipedia.org/wiki/Posting_style>
<http://allmyfaqs.net/faq.pl?Top-posting_or_bottom-posting>
And so on. Many times.
Joe
"Mike Feher" <mfeher@eozinc.com>
Sent by: time-nuts-bounces@febo.com
12/10/2008 09:06 PM
Please respond to
Discussion of precise time and frequency measurement <time-nuts@febo.com>
To
"'Discussion of precise time and frequency measurement'"
<time-nuts@febo.com>
cc
Subject
Re: [time-nuts] Sub Pico Second Phase logger
I am now, and actually have been, at the point where I just do not read
bottom line post/replies. Bruce has a lot of good information to share,
but, now, if I click on a post, and do not immediately see a response it
is just deleted. Maybe it will be my loss, but, technology as well as
the internet is evolving, and bottom line replies totally suck. - Mike
Mike B. Feher
EOZ Inc.
89 Arnold Blvd.
Howell, NJ, 07731
732-886-5960
908-902-3831 - cell
-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com] On
Behalf Of Bruce Griffiths
Sent: Wednesday, December 10, 2008 8:38 PM
To: Discussion of precise time and frequency measurement
Subject: Re: [time-nuts] Sub Pico Second Phase logger
Joe
Joseph M Gwinn wrote:
> Bruce,
>
>
>> Reflecting the sum frequency back into the mixer is actually
necessary
>> to reduce the noise at the IF port.
>> I believe that one of Agilent's simulation application notes mentions
>> this effect but I don't recall the actual application note number.
>> This will affect the mixer RF and IF port impedance so adding a
series
>> resistor may be required to improve the SWR.
>>
>
> How big an effect is this? Is the absolute noise decreased, or does
it
> remain the same while the signal increase?
>
>
With the same difference frequency IF port termination impedance, noise
is actually decreased along with the mixer conversion loss.
However if the sound card input noise dominates reducing the mixer
effective output noise wont help.
> If I'm understanding Walls and Stein (paper 112) correctly, the
advantage
> is because with the capacitor load the beatnote waveform approaches
> square, thus increasing the zero-crossing speed and therefor the phase
> sensitivity. This is no doubt true, but the question was if this also
> caused a small everything-dependent phase shift, something that would
not
> have mattered in the measurement of phase noise. The object of paper
112
> was to remedy a 10 to 20 dB error in phase noise measurements. The
> critical words are in the lower left column of page 337, in the
paragraph
> beginning "If the mixer is terminated ...".
>
>
>
Saturating the RF port has a similar effect.
If one is time stamping the zero crossings an increased zero crossing
slope is an advantage.
For relative phase measurements a trapezoidal beat frequency waveform
may be less useful.
>>> MiniCircuits AN-41-001 "FAQ about Phase Detectors" has on page 2 a
500
>>>
> ohm
>
>>> resistor to ground and a 5000 ohm resistor to the first filter
>>>
> capacitor,
>
>>> so the capacitor is isolated from the IF port by the resistors.
>>>
>>>
>>>
>> I wouldn't take too much notice of that recommendation as I have
little
>> confidence in the author's experience/knowledge.
>>
>
> Well, OK, but:
>
> Stephen Kurtz says the same thing on the third column of the third
page, a
> bit above Figure 6.
>
Off course with a capacitive IF port termination matching the RF and LO
ports becomes more critical as does the reverse isolation of the various
amplifiers driving the RF and LO ports.
It may be simpler in fact to use a level 17 mixer with high LO to RF and
LO to IF isolation with the RF port unsaturated as it relaxes the
reverse isolation specs for the isolation amplifiers.
> Nelson and Walls (paper 971), Figure 4, also shows the low pass filter
> arranged to absorb the sum signal, not allowing it to be reflected
back
> into the mixer.
>
>
>
>>>> Supposedly an SRA-1, but some caution is in order as some
>>>>
>> statements as
>>
>>>> to the effect of the input offset of an opamp based IF preamp in
the
>>>> same application note were of dubious veracity unless one
>>>>
>> were to use an
>>
>>>> inverting opamp input stage.
>>>>
>>>>
>>> This issue was mentioned in another app note, but their main issue
>>> appeared to be that the opamp bias currents could cause an offset.
>>>
>>>
>>>
>>>
>> But the circuit they suggest has no effect on bias current induced
>> offset, the same current flows into the mixer and termination
impedance
>> independent of the series resistance.
>>
>
> You're right that the proposed remedy didn't make sense. I don't know
> that this is a big problem with modern opamps, especially FET input
ones
> (if needed).
>
>
The only configuration for which it makes any sense is an inverting
input amplifier with a finite input voltage offset.
>> Yes, I should have said that when the 2 input signals are in
quadrature,
>> any capacitive crosstalk will have little effect on the phase shift.
>>
>
> Ah. Because the capacitor coupling adds a second 90 degree shift,
> bringing the total to 180 degrees.
>
> But crosstalk by ground coupling will be unaffected. As will
crosstalk by
> transformer action. Those boards are pretty crowded.
>
>
>
Yes its better to measure it rather than relying too much on conjecture.
>> The AP192 has a somewhat higher interchannel isolation than that, the
>> interchannel crosstalk spec is about -120dB.
>> With a sufficiently large number of samples the its easy to see
>> artifacts as low as -140dBFS.
>>
>
> Yep. Seems like a very good card.
>
>
>
>
>
>> It's hard to find such Firewire systems without such unnecessary
frills
>> (for this application) as high gain preamps.
>>
>
> The AP192 has high-level inputs, but I don't know if this bypasses the
> preamps, or attenuates. Given their target market, I'd bet it
bypasses.
>
>
There are no preamps other than an external differential input amplifier
that translates the 4 Vrms FS inputs at the input connector to a level
that the ADC can handle.
The ADC chip itself has no preamps built in.
There have been numerous complaint about this by some audio nuts,
however for this application not having such amplifiers is ideal.
>
>> The gain tempco and linearity of some variable gain audio preamps is
>> somewhat suspect.
>>
>
> I would think that none of these cards has a good tempco of anything,
> given the lack of necessity in their market.
>
> I would think that linearity would be quite good, given the horsepower
> competitions on linearity.
>
>
Since the 2 ADCs share the same reference their gain tracking tempco
should be quite good given that they use capacitors rather than
resistors within the ADCs.
>
>>>> Other cards using AKM 24 bit ADCs should also be suitable.
>>>>
>>>>
>>> Who is AKM?
>>>
>>>
>>>
>> Asahi Kasei EKM
>>
>> http://www.asahi-kasei.co.jp/akm/en/
>>
>> http://www.asahi-kasei.co.jp/akm/en/product/proaudio.html
>>
>>
>
> Thanks. I'll look into their data.
>
>
>
>>> 20 Log[ 2^24 ] = 144 dB, so something else will be the limit.
>>>
>>>
>>>
>>>
>> Actual ENOB ~ 19 to 20 bits.
>>
>
> Makes sense. 20 Log [ 2^19 ] = 114 dB. Still plenty good enough.
>
>
>
>>>> Ideally an external sound card with balanced XLR inputs would be
>>>>
> best.
>
> Yes.
>
>
>
>
>>>> HP produced a number of different phase comparators each with a
>>>> different type of phase detector.
>>>>
>
> OK. And the PLL folk must have a million designs.
>
>
>
>>
>> Can alleviate it to some extent by driving a pair of such phase
>> detectors so that their outputs are in quadrature.
>> One just selects the phase detector output that is in the linear
range.
>> The quadrature outputs also allow unambiguous assignment of the sign
of
>> any phase change.
>>
>
> The Symmetricom 5120A does something very clever to alleviate this
> problem. Explained in US patent 7,227,346 and "Direct-Digital
Phase-Noise
> Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger, W. Solbrig,
and
> S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics, and
> Frequency Control Joint 50th Anniversary Conference, pages 287-291.
>
> Joe
>
>
I've read the patent.
Bruce
_______________________________________________
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To unsubscribe, go to
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To unsubscribe, go to
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and follow the instructions there.
MF
Mike Feher
Thu, Dec 11, 2008 2:36 AM
I do not give a shit about a fight or Wiki. I was only stating my
position. BTW, good thing you stated your post on top, otherwise I would
not have seen it :). - Mike
Mike B. Feher
EOZ Inc.
89 Arnold Blvd.
Howell, NJ, 07731
732-886-5960
908-902-3831 - cell
-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com] On
Behalf Of Joseph M Gwinn
Sent: Wednesday, December 10, 2008 9:27 PM
To: Discussion of precise time and frequency measurement
Subject: Re: [time-nuts] Sub Pico Second Phase logger (Posting Style)
Let's have a top-posting versus bottom-posting fight!
But we're too late, it's already been done:
http://en.wikipedia.org/wiki/Posting_style
http://allmyfaqs.net/faq.pl?Top-posting_or_bottom-posting
And so on. Many times.
Joe
"Mike Feher" mfeher@eozinc.com
Sent by: time-nuts-bounces@febo.com
12/10/2008 09:06 PM
Please respond to
Discussion of precise time and frequency measurement
time-nuts@febo.com
To
"'Discussion of precise time and frequency measurement'"
time-nuts@febo.com
cc
Subject
Re: [time-nuts] Sub Pico Second Phase logger
I am now, and actually have been, at the point where I just do not read
bottom line post/replies. Bruce has a lot of good information to share,
but, now, if I click on a post, and do not immediately see a response it
is just deleted. Maybe it will be my loss, but, technology as well as
the internet is evolving, and bottom line replies totally suck. - Mike
Mike B. Feher
EOZ Inc.
89 Arnold Blvd.
Howell, NJ, 07731
732-886-5960
908-902-3831 - cell
-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com] On
Behalf Of Bruce Griffiths
Sent: Wednesday, December 10, 2008 8:38 PM
To: Discussion of precise time and frequency measurement
Subject: Re: [time-nuts] Sub Pico Second Phase logger
Joe
Joseph M Gwinn wrote:
Reflecting the sum frequency back into the mixer is actually
to reduce the noise at the IF port.
I believe that one of Agilent's simulation application notes mentions
this effect but I don't recall the actual application note number.
This will affect the mixer RF and IF port impedance so adding a
resistor may be required to improve the SWR.
How big an effect is this? Is the absolute noise decreased, or does
remain the same while the signal increase?
With the same difference frequency IF port termination impedance, noise
is actually decreased along with the mixer conversion loss.
However if the sound card input noise dominates reducing the mixer
effective output noise wont help.
If I'm understanding Walls and Stein (paper 112) correctly, the
is because with the capacitor load the beatnote waveform approaches
square, thus increasing the zero-crossing speed and therefor the phase
sensitivity. This is no doubt true, but the question was if this also
caused a small everything-dependent phase shift, something that would
have mattered in the measurement of phase noise. The object of paper
was to remedy a 10 to 20 dB error in phase noise measurements. The
critical words are in the lower left column of page 337, in the
beginning "If the mixer is terminated ...".
Saturating the RF port has a similar effect.
If one is time stamping the zero crossings an increased zero crossing
slope is an advantage.
For relative phase measurements a trapezoidal beat frequency waveform
may be less useful.
MiniCircuits AN-41-001 "FAQ about Phase Detectors" has on page 2 a
resistor to ground and a 5000 ohm resistor to the first filter
so the capacitor is isolated from the IF port by the resistors.
I wouldn't take too much notice of that recommendation as I have
confidence in the author's experience/knowledge.
Well, OK, but:
Stephen Kurtz says the same thing on the third column of the third
Off course with a capacitive IF port termination matching the RF and LO
ports becomes more critical as does the reverse isolation of the various
amplifiers driving the RF and LO ports.
It may be simpler in fact to use a level 17 mixer with high LO to RF and
LO to IF isolation with the RF port unsaturated as it relaxes the
reverse isolation specs for the isolation amplifiers.
Nelson and Walls (paper 971), Figure 4, also shows the low pass filter
arranged to absorb the sum signal, not allowing it to be reflected
Supposedly an SRA-1, but some caution is in order as some
to the effect of the input offset of an opamp based IF preamp in
same application note were of dubious veracity unless one
inverting opamp input stage.
This issue was mentioned in another app note, but their main issue
appeared to be that the opamp bias currents could cause an offset.
But the circuit they suggest has no effect on bias current induced
offset, the same current flows into the mixer and termination
independent of the series resistance.
You're right that the proposed remedy didn't make sense. I don't know
that this is a big problem with modern opamps, especially FET input
The only configuration for which it makes any sense is an inverting
input amplifier with a finite input voltage offset.
Yes, I should have said that when the 2 input signals are in
any capacitive crosstalk will have little effect on the phase shift.
Ah. Because the capacitor coupling adds a second 90 degree shift,
bringing the total to 180 degrees.
But crosstalk by ground coupling will be unaffected. As will
transformer action. Those boards are pretty crowded.
Yes its better to measure it rather than relying too much on conjecture.
The AP192 has a somewhat higher interchannel isolation than that, the
interchannel crosstalk spec is about -120dB.
With a sufficiently large number of samples the its easy to see
artifacts as low as -140dBFS.
Yep. Seems like a very good card.
It's hard to find such Firewire systems without such unnecessary
(for this application) as high gain preamps.
The AP192 has high-level inputs, but I don't know if this bypasses the
preamps, or attenuates. Given their target market, I'd bet it
There are no preamps other than an external differential input amplifier
that translates the 4 Vrms FS inputs at the input connector to a level
that the ADC can handle.
The ADC chip itself has no preamps built in.
There have been numerous complaint about this by some audio nuts,
however for this application not having such amplifiers is ideal.
The gain tempco and linearity of some variable gain audio preamps is
somewhat suspect.
I would think that none of these cards has a good tempco of anything,
given the lack of necessity in their market.
I would think that linearity would be quite good, given the horsepower
competitions on linearity.
Since the 2 ADCs share the same reference their gain tracking tempco
should be quite good given that they use capacitors rather than
resistors within the ADCs.
Other cards using AKM 24 bit ADCs should also be suitable.
Thanks. I'll look into their data.
20 Log[ 2^24 ] = 144 dB, so something else will be the limit.
Actual ENOB ~ 19 to 20 bits.
Makes sense. 20 Log [ 2^19 ] = 114 dB. Still plenty good enough.
Ideally an external sound card with balanced XLR inputs would be
HP produced a number of different phase comparators each with a
different type of phase detector.
OK. And the PLL folk must have a million designs.
Can alleviate it to some extent by driving a pair of such phase
detectors so that their outputs are in quadrature.
One just selects the phase detector output that is in the linear
The quadrature outputs also allow unambiguous assignment of the sign
The Symmetricom 5120A does something very clever to alleviate this
problem. Explained in US patent 7,227,346 and "Direct-Digital
Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger, W. Solbrig,
S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics, and
Frequency Control Joint 50th Anniversary Conference, pages 287-291.
Joe
I do not give a shit about a fight or Wiki. I was only stating my
position. BTW, good thing you stated your post on top, otherwise I would
not have seen it :). - Mike
Mike B. Feher
EOZ Inc.
89 Arnold Blvd.
Howell, NJ, 07731
732-886-5960
908-902-3831 - cell
-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com] On
Behalf Of Joseph M Gwinn
Sent: Wednesday, December 10, 2008 9:27 PM
To: Discussion of precise time and frequency measurement
Subject: Re: [time-nuts] Sub Pico Second Phase logger (Posting Style)
Let's have a top-posting versus bottom-posting fight!
But we're too late, it's already been done:
<http://en.wikipedia.org/wiki/Posting_style>
<http://allmyfaqs.net/faq.pl?Top-posting_or_bottom-posting>
And so on. Many times.
Joe
"Mike Feher" <mfeher@eozinc.com>
Sent by: time-nuts-bounces@febo.com
12/10/2008 09:06 PM
Please respond to
Discussion of precise time and frequency measurement
<time-nuts@febo.com>
To
"'Discussion of precise time and frequency measurement'"
<time-nuts@febo.com>
cc
Subject
Re: [time-nuts] Sub Pico Second Phase logger
I am now, and actually have been, at the point where I just do not read
bottom line post/replies. Bruce has a lot of good information to share,
but, now, if I click on a post, and do not immediately see a response it
is just deleted. Maybe it will be my loss, but, technology as well as
the internet is evolving, and bottom line replies totally suck. - Mike
Mike B. Feher
EOZ Inc.
89 Arnold Blvd.
Howell, NJ, 07731
732-886-5960
908-902-3831 - cell
-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com] On
Behalf Of Bruce Griffiths
Sent: Wednesday, December 10, 2008 8:38 PM
To: Discussion of precise time and frequency measurement
Subject: Re: [time-nuts] Sub Pico Second Phase logger
Joe
Joseph M Gwinn wrote:
> Bruce,
>
>
>> Reflecting the sum frequency back into the mixer is actually
necessary
>> to reduce the noise at the IF port.
>> I believe that one of Agilent's simulation application notes mentions
>> this effect but I don't recall the actual application note number.
>> This will affect the mixer RF and IF port impedance so adding a
series
>> resistor may be required to improve the SWR.
>>
>
> How big an effect is this? Is the absolute noise decreased, or does
it
> remain the same while the signal increase?
>
>
With the same difference frequency IF port termination impedance, noise
is actually decreased along with the mixer conversion loss.
However if the sound card input noise dominates reducing the mixer
effective output noise wont help.
> If I'm understanding Walls and Stein (paper 112) correctly, the
advantage
> is because with the capacitor load the beatnote waveform approaches
> square, thus increasing the zero-crossing speed and therefor the phase
> sensitivity. This is no doubt true, but the question was if this also
> caused a small everything-dependent phase shift, something that would
not
> have mattered in the measurement of phase noise. The object of paper
112
> was to remedy a 10 to 20 dB error in phase noise measurements. The
> critical words are in the lower left column of page 337, in the
paragraph
> beginning "If the mixer is terminated ...".
>
>
>
Saturating the RF port has a similar effect.
If one is time stamping the zero crossings an increased zero crossing
slope is an advantage.
For relative phase measurements a trapezoidal beat frequency waveform
may be less useful.
>>> MiniCircuits AN-41-001 "FAQ about Phase Detectors" has on page 2 a
500
>>>
> ohm
>
>>> resistor to ground and a 5000 ohm resistor to the first filter
>>>
> capacitor,
>
>>> so the capacitor is isolated from the IF port by the resistors.
>>>
>>>
>>>
>> I wouldn't take too much notice of that recommendation as I have
little
>> confidence in the author's experience/knowledge.
>>
>
> Well, OK, but:
>
> Stephen Kurtz says the same thing on the third column of the third
page, a
> bit above Figure 6.
>
Off course with a capacitive IF port termination matching the RF and LO
ports becomes more critical as does the reverse isolation of the various
amplifiers driving the RF and LO ports.
It may be simpler in fact to use a level 17 mixer with high LO to RF and
LO to IF isolation with the RF port unsaturated as it relaxes the
reverse isolation specs for the isolation amplifiers.
> Nelson and Walls (paper 971), Figure 4, also shows the low pass filter
> arranged to absorb the sum signal, not allowing it to be reflected
back
> into the mixer.
>
>
>
>>>> Supposedly an SRA-1, but some caution is in order as some
>>>>
>> statements as
>>
>>>> to the effect of the input offset of an opamp based IF preamp in
the
>>>> same application note were of dubious veracity unless one
>>>>
>> were to use an
>>
>>>> inverting opamp input stage.
>>>>
>>>>
>>> This issue was mentioned in another app note, but their main issue
>>> appeared to be that the opamp bias currents could cause an offset.
>>>
>>>
>>>
>>>
>> But the circuit they suggest has no effect on bias current induced
>> offset, the same current flows into the mixer and termination
impedance
>> independent of the series resistance.
>>
>
> You're right that the proposed remedy didn't make sense. I don't know
> that this is a big problem with modern opamps, especially FET input
ones
> (if needed).
>
>
The only configuration for which it makes any sense is an inverting
input amplifier with a finite input voltage offset.
>> Yes, I should have said that when the 2 input signals are in
quadrature,
>> any capacitive crosstalk will have little effect on the phase shift.
>>
>
> Ah. Because the capacitor coupling adds a second 90 degree shift,
> bringing the total to 180 degrees.
>
> But crosstalk by ground coupling will be unaffected. As will
crosstalk by
> transformer action. Those boards are pretty crowded.
>
>
>
Yes its better to measure it rather than relying too much on conjecture.
>> The AP192 has a somewhat higher interchannel isolation than that, the
>> interchannel crosstalk spec is about -120dB.
>> With a sufficiently large number of samples the its easy to see
>> artifacts as low as -140dBFS.
>>
>
> Yep. Seems like a very good card.
>
>
>
>
>
>> It's hard to find such Firewire systems without such unnecessary
frills
>> (for this application) as high gain preamps.
>>
>
> The AP192 has high-level inputs, but I don't know if this bypasses the
> preamps, or attenuates. Given their target market, I'd bet it
bypasses.
>
>
There are no preamps other than an external differential input amplifier
that translates the 4 Vrms FS inputs at the input connector to a level
that the ADC can handle.
The ADC chip itself has no preamps built in.
There have been numerous complaint about this by some audio nuts,
however for this application not having such amplifiers is ideal.
>
>> The gain tempco and linearity of some variable gain audio preamps is
>> somewhat suspect.
>>
>
> I would think that none of these cards has a good tempco of anything,
> given the lack of necessity in their market.
>
> I would think that linearity would be quite good, given the horsepower
> competitions on linearity.
>
>
Since the 2 ADCs share the same reference their gain tracking tempco
should be quite good given that they use capacitors rather than
resistors within the ADCs.
>
>>>> Other cards using AKM 24 bit ADCs should also be suitable.
>>>>
>>>>
>>> Who is AKM?
>>>
>>>
>>>
>> Asahi Kasei EKM
>>
>> http://www.asahi-kasei.co.jp/akm/en/
>>
>> http://www.asahi-kasei.co.jp/akm/en/product/proaudio.html
>>
>>
>
> Thanks. I'll look into their data.
>
>
>
>>> 20 Log[ 2^24 ] = 144 dB, so something else will be the limit.
>>>
>>>
>>>
>>>
>> Actual ENOB ~ 19 to 20 bits.
>>
>
> Makes sense. 20 Log [ 2^19 ] = 114 dB. Still plenty good enough.
>
>
>
>>>> Ideally an external sound card with balanced XLR inputs would be
>>>>
> best.
>
> Yes.
>
>
>
>
>>>> HP produced a number of different phase comparators each with a
>>>> different type of phase detector.
>>>>
>
> OK. And the PLL folk must have a million designs.
>
>
>
>>
>> Can alleviate it to some extent by driving a pair of such phase
>> detectors so that their outputs are in quadrature.
>> One just selects the phase detector output that is in the linear
range.
>> The quadrature outputs also allow unambiguous assignment of the sign
of
>> any phase change.
>>
>
> The Symmetricom 5120A does something very clever to alleviate this
> problem. Explained in US patent 7,227,346 and "Direct-Digital
Phase-Noise
> Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger, W. Solbrig,
and
> S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics, and
> Frequency Control Joint 50th Anniversary Conference, pages 287-291.
>
> Joe
>
>
I've read the patent.
Bruce
_______________________________________________
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_______________________________________________
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To unsubscribe, go to
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_______________________________________________
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To unsubscribe, go to
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and follow the instructions there.
BG
Bruce Griffiths
Thu, Dec 11, 2008 3:32 AM
Joe
A screen shot indicating the AP192 sound card noise level with nothing
connected to the inputs is attached.
Reference level is full scale (4Vrms).
Aside from the few discrete spurs which differ between both channels the
region from 1 - 10kHz is fairly quiet.
Technical Notes:
FFT bin equivalent noise bandwidth ~ 3Hz.
96 KSPS.
Bruce
Joe
A screen shot indicating the AP192 sound card noise level with nothing
connected to the inputs is attached.
Reference level is full scale (4Vrms).
Aside from the few discrete spurs which differ between both channels the
region from 1 - 10kHz is fairly quiet.
Technical Notes:
FFT bin equivalent noise bandwidth ~ 3Hz.
96 KSPS.
Bruce
SR
Steve Rooke
Thu, Dec 11, 2008 3:52 AM
You can blow out a candle but you can't blow out a fire...
2008/12/11 Mike Feher mfeher@eozinc.com:
I do not give a shit about a fight or Wiki. I was only stating my
position. BTW, good thing you stated your post on top, otherwise I would
not have seen it :). - Mike
Mike B. Feher
EOZ Inc.
89 Arnold Blvd.
Howell, NJ, 07731
732-886-5960
908-902-3831 - cell
-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com] On
Behalf Of Joseph M Gwinn
Sent: Wednesday, December 10, 2008 9:27 PM
To: Discussion of precise time and frequency measurement
Subject: Re: [time-nuts] Sub Pico Second Phase logger (Posting Style)
Let's have a top-posting versus bottom-posting fight!
But we're too late, it's already been done:
http://en.wikipedia.org/wiki/Posting_style
http://allmyfaqs.net/faq.pl?Top-posting_or_bottom-posting
And so on. Many times.
Joe
"Mike Feher" mfeher@eozinc.com
Sent by: time-nuts-bounces@febo.com
12/10/2008 09:06 PM
Please respond to
Discussion of precise time and frequency measurement
time-nuts@febo.com
To
"'Discussion of precise time and frequency measurement'"
time-nuts@febo.com
cc
Subject
Re: [time-nuts] Sub Pico Second Phase logger
I am now, and actually have been, at the point where I just do not read
bottom line post/replies. Bruce has a lot of good information to share,
but, now, if I click on a post, and do not immediately see a response it
is just deleted. Maybe it will be my loss, but, technology as well as
the internet is evolving, and bottom line replies totally suck. - Mike
Mike B. Feher
EOZ Inc.
89 Arnold Blvd.
Howell, NJ, 07731
732-886-5960
908-902-3831 - cell
-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com] On
Behalf Of Bruce Griffiths
Sent: Wednesday, December 10, 2008 8:38 PM
To: Discussion of precise time and frequency measurement
Subject: Re: [time-nuts] Sub Pico Second Phase logger
Joe
Joseph M Gwinn wrote:
Reflecting the sum frequency back into the mixer is actually
to reduce the noise at the IF port.
I believe that one of Agilent's simulation application notes mentions
this effect but I don't recall the actual application note number.
This will affect the mixer RF and IF port impedance so adding a
resistor may be required to improve the SWR.
How big an effect is this? Is the absolute noise decreased, or does
remain the same while the signal increase?
With the same difference frequency IF port termination impedance, noise
is actually decreased along with the mixer conversion loss.
However if the sound card input noise dominates reducing the mixer
effective output noise wont help.
If I'm understanding Walls and Stein (paper 112) correctly, the
is because with the capacitor load the beatnote waveform approaches
square, thus increasing the zero-crossing speed and therefor the phase
sensitivity. This is no doubt true, but the question was if this also
caused a small everything-dependent phase shift, something that would
have mattered in the measurement of phase noise. The object of paper
was to remedy a 10 to 20 dB error in phase noise measurements. The
critical words are in the lower left column of page 337, in the
beginning "If the mixer is terminated ...".
Saturating the RF port has a similar effect.
If one is time stamping the zero crossings an increased zero crossing
slope is an advantage.
For relative phase measurements a trapezoidal beat frequency waveform
may be less useful.
MiniCircuits AN-41-001 "FAQ about Phase Detectors" has on page 2 a
resistor to ground and a 5000 ohm resistor to the first filter
so the capacitor is isolated from the IF port by the resistors.
I wouldn't take too much notice of that recommendation as I have
confidence in the author's experience/knowledge.
Well, OK, but:
Stephen Kurtz says the same thing on the third column of the third
Off course with a capacitive IF port termination matching the RF and LO
ports becomes more critical as does the reverse isolation of the various
amplifiers driving the RF and LO ports.
It may be simpler in fact to use a level 17 mixer with high LO to RF and
LO to IF isolation with the RF port unsaturated as it relaxes the
reverse isolation specs for the isolation amplifiers.
Nelson and Walls (paper 971), Figure 4, also shows the low pass filter
arranged to absorb the sum signal, not allowing it to be reflected
Supposedly an SRA-1, but some caution is in order as some
to the effect of the input offset of an opamp based IF preamp in
same application note were of dubious veracity unless one
inverting opamp input stage.
This issue was mentioned in another app note, but their main issue
appeared to be that the opamp bias currents could cause an offset.
But the circuit they suggest has no effect on bias current induced
offset, the same current flows into the mixer and termination
independent of the series resistance.
You're right that the proposed remedy didn't make sense. I don't know
that this is a big problem with modern opamps, especially FET input
The only configuration for which it makes any sense is an inverting
input amplifier with a finite input voltage offset.
Yes, I should have said that when the 2 input signals are in
any capacitive crosstalk will have little effect on the phase shift.
Ah. Because the capacitor coupling adds a second 90 degree shift,
bringing the total to 180 degrees.
But crosstalk by ground coupling will be unaffected. As will
transformer action. Those boards are pretty crowded.
Yes its better to measure it rather than relying too much on conjecture.
The AP192 has a somewhat higher interchannel isolation than that, the
interchannel crosstalk spec is about -120dB.
With a sufficiently large number of samples the its easy to see
artifacts as low as -140dBFS.
Yep. Seems like a very good card.
It's hard to find such Firewire systems without such unnecessary
(for this application) as high gain preamps.
The AP192 has high-level inputs, but I don't know if this bypasses the
preamps, or attenuates. Given their target market, I'd bet it
There are no preamps other than an external differential input amplifier
that translates the 4 Vrms FS inputs at the input connector to a level
that the ADC can handle.
The ADC chip itself has no preamps built in.
There have been numerous complaint about this by some audio nuts,
however for this application not having such amplifiers is ideal.
The gain tempco and linearity of some variable gain audio preamps is
somewhat suspect.
I would think that none of these cards has a good tempco of anything,
given the lack of necessity in their market.
I would think that linearity would be quite good, given the horsepower
competitions on linearity.
Since the 2 ADCs share the same reference their gain tracking tempco
should be quite good given that they use capacitors rather than
resistors within the ADCs.
Other cards using AKM 24 bit ADCs should also be suitable.
Thanks. I'll look into their data.
20 Log[ 2^24 ] = 144 dB, so something else will be the limit.
Actual ENOB ~ 19 to 20 bits.
Makes sense. 20 Log [ 2^19 ] = 114 dB. Still plenty good enough.
Ideally an external sound card with balanced XLR inputs would be
HP produced a number of different phase comparators each with a
different type of phase detector.
OK. And the PLL folk must have a million designs.
Can alleviate it to some extent by driving a pair of such phase
detectors so that their outputs are in quadrature.
One just selects the phase detector output that is in the linear
The quadrature outputs also allow unambiguous assignment of the sign
The Symmetricom 5120A does something very clever to alleviate this
problem. Explained in US patent 7,227,346 and "Direct-Digital
Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger, W. Solbrig,
S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics, and
Frequency Control Joint 50th Anniversary Conference, pages 287-291.
Joe
--
Steve Rooke - ZL3TUV & G8KVD
Omnium finis imminet
You can blow out a candle but you can't blow out a fire...
2008/12/11 Mike Feher <mfeher@eozinc.com>:
> I do not give a shit about a fight or Wiki. I was only stating my
> position. BTW, good thing you stated your post on top, otherwise I would
> not have seen it :). - Mike
>
>
>
> Mike B. Feher
> EOZ Inc.
> 89 Arnold Blvd.
> Howell, NJ, 07731
> 732-886-5960
> 908-902-3831 - cell
>
>
>
> -----Original Message-----
> From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com] On
> Behalf Of Joseph M Gwinn
> Sent: Wednesday, December 10, 2008 9:27 PM
> To: Discussion of precise time and frequency measurement
> Subject: Re: [time-nuts] Sub Pico Second Phase logger (Posting Style)
>
> Let's have a top-posting versus bottom-posting fight!
>
> But we're too late, it's already been done:
>
> <http://en.wikipedia.org/wiki/Posting_style>
>
> <http://allmyfaqs.net/faq.pl?Top-posting_or_bottom-posting>
>
> And so on. Many times.
>
> Joe
>
>
>
>
> "Mike Feher" <mfeher@eozinc.com>
> Sent by: time-nuts-bounces@febo.com
> 12/10/2008 09:06 PM
> Please respond to
> Discussion of precise time and frequency measurement
> <time-nuts@febo.com>
>
>
> To
> "'Discussion of precise time and frequency measurement'"
> <time-nuts@febo.com>
> cc
>
> Subject
> Re: [time-nuts] Sub Pico Second Phase logger
>
>
>
>
>
>
> I am now, and actually have been, at the point where I just do not read
> bottom line post/replies. Bruce has a lot of good information to share,
> but, now, if I click on a post, and do not immediately see a response it
> is just deleted. Maybe it will be my loss, but, technology as well as
> the internet is evolving, and bottom line replies totally suck. - Mike
>
>
> Mike B. Feher
> EOZ Inc.
> 89 Arnold Blvd.
> Howell, NJ, 07731
> 732-886-5960
> 908-902-3831 - cell
>
>
>
> -----Original Message-----
> From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com] On
> Behalf Of Bruce Griffiths
> Sent: Wednesday, December 10, 2008 8:38 PM
> To: Discussion of precise time and frequency measurement
> Subject: Re: [time-nuts] Sub Pico Second Phase logger
>
> Joe
> Joseph M Gwinn wrote:
>> Bruce,
>>
>>
>>> Reflecting the sum frequency back into the mixer is actually
> necessary
>>> to reduce the noise at the IF port.
>>> I believe that one of Agilent's simulation application notes mentions
>>> this effect but I don't recall the actual application note number.
>>> This will affect the mixer RF and IF port impedance so adding a
> series
>>> resistor may be required to improve the SWR.
>>>
>>
>> How big an effect is this? Is the absolute noise decreased, or does
> it
>> remain the same while the signal increase?
>>
>>
> With the same difference frequency IF port termination impedance, noise
> is actually decreased along with the mixer conversion loss.
> However if the sound card input noise dominates reducing the mixer
> effective output noise wont help.
>> If I'm understanding Walls and Stein (paper 112) correctly, the
> advantage
>> is because with the capacitor load the beatnote waveform approaches
>> square, thus increasing the zero-crossing speed and therefor the phase
>
>> sensitivity. This is no doubt true, but the question was if this also
>
>> caused a small everything-dependent phase shift, something that would
> not
>> have mattered in the measurement of phase noise. The object of paper
> 112
>> was to remedy a 10 to 20 dB error in phase noise measurements. The
>> critical words are in the lower left column of page 337, in the
> paragraph
>> beginning "If the mixer is terminated ...".
>>
>>
>>
> Saturating the RF port has a similar effect.
> If one is time stamping the zero crossings an increased zero crossing
> slope is an advantage.
> For relative phase measurements a trapezoidal beat frequency waveform
> may be less useful.
>>>> MiniCircuits AN-41-001 "FAQ about Phase Detectors" has on page 2 a
> 500
>>>>
>> ohm
>>
>>>> resistor to ground and a 5000 ohm resistor to the first filter
>>>>
>> capacitor,
>>
>>>> so the capacitor is isolated from the IF port by the resistors.
>>>>
>>>>
>>>>
>>> I wouldn't take too much notice of that recommendation as I have
> little
>>> confidence in the author's experience/knowledge.
>>>
>>
>> Well, OK, but:
>>
>> Stephen Kurtz says the same thing on the third column of the third
> page, a
>> bit above Figure 6.
>>
>
> Off course with a capacitive IF port termination matching the RF and LO
> ports becomes more critical as does the reverse isolation of the various
> amplifiers driving the RF and LO ports.
> It may be simpler in fact to use a level 17 mixer with high LO to RF and
> LO to IF isolation with the RF port unsaturated as it relaxes the
> reverse isolation specs for the isolation amplifiers.
>> Nelson and Walls (paper 971), Figure 4, also shows the low pass filter
>
>> arranged to absorb the sum signal, not allowing it to be reflected
> back
>> into the mixer.
>>
>>
>>
>>>>> Supposedly an SRA-1, but some caution is in order as some
>>>>>
>>> statements as
>>>
>>>>> to the effect of the input offset of an opamp based IF preamp in
> the
>>>>> same application note were of dubious veracity unless one
>>>>>
>>> were to use an
>>>
>>>>> inverting opamp input stage.
>>>>>
>>>>>
>>>> This issue was mentioned in another app note, but their main issue
>>>> appeared to be that the opamp bias currents could cause an offset.
>>>>
>>>>
>>>>
>>>>
>>> But the circuit they suggest has no effect on bias current induced
>>> offset, the same current flows into the mixer and termination
> impedance
>>> independent of the series resistance.
>>>
>>
>> You're right that the proposed remedy didn't make sense. I don't know
>
>> that this is a big problem with modern opamps, especially FET input
> ones
>> (if needed).
>>
>>
> The only configuration for which it makes any sense is an inverting
> input amplifier with a finite input voltage offset.
>>> Yes, I should have said that when the 2 input signals are in
> quadrature,
>>> any capacitive crosstalk will have little effect on the phase shift.
>>>
>>
>> Ah. Because the capacitor coupling adds a second 90 degree shift,
>> bringing the total to 180 degrees.
>>
>> But crosstalk by ground coupling will be unaffected. As will
> crosstalk by
>> transformer action. Those boards are pretty crowded.
>>
>>
>>
> Yes its better to measure it rather than relying too much on conjecture.
>>> The AP192 has a somewhat higher interchannel isolation than that, the
>>> interchannel crosstalk spec is about -120dB.
>>> With a sufficiently large number of samples the its easy to see
>>> artifacts as low as -140dBFS.
>>>
>>
>> Yep. Seems like a very good card.
>>
>>
>>
>>
>>
>>> It's hard to find such Firewire systems without such unnecessary
> frills
>>> (for this application) as high gain preamps.
>>>
>>
>> The AP192 has high-level inputs, but I don't know if this bypasses the
>
>> preamps, or attenuates. Given their target market, I'd bet it
> bypasses.
>>
>>
> There are no preamps other than an external differential input amplifier
> that translates the 4 Vrms FS inputs at the input connector to a level
> that the ADC can handle.
> The ADC chip itself has no preamps built in.
> There have been numerous complaint about this by some audio nuts,
> however for this application not having such amplifiers is ideal.
>>
>>> The gain tempco and linearity of some variable gain audio preamps is
>>> somewhat suspect.
>>>
>>
>> I would think that none of these cards has a good tempco of anything,
>> given the lack of necessity in their market.
>>
>> I would think that linearity would be quite good, given the horsepower
>
>> competitions on linearity.
>>
>>
> Since the 2 ADCs share the same reference their gain tracking tempco
> should be quite good given that they use capacitors rather than
> resistors within the ADCs.
>>
>>>>> Other cards using AKM 24 bit ADCs should also be suitable.
>>>>>
>>>>>
>>>> Who is AKM?
>>>>
>>>>
>>>>
>>> Asahi Kasei EKM
>>>
>>> http://www.asahi-kasei.co.jp/akm/en/
>>>
>>> http://www.asahi-kasei.co.jp/akm/en/product/proaudio.html
>>>
>>>
>>
>> Thanks. I'll look into their data.
>>
>>
>>
>>>> 20 Log[ 2^24 ] = 144 dB, so something else will be the limit.
>>>>
>>>>
>>>>
>>>>
>>> Actual ENOB ~ 19 to 20 bits.
>>>
>>
>> Makes sense. 20 Log [ 2^19 ] = 114 dB. Still plenty good enough.
>>
>>
>>
>>>>> Ideally an external sound card with balanced XLR inputs would be
>>>>>
>> best.
>>
>> Yes.
>>
>>
>>
>>
>>>>> HP produced a number of different phase comparators each with a
>>>>> different type of phase detector.
>>>>>
>>
>> OK. And the PLL folk must have a million designs.
>>
>>
>>
>>>
>>> Can alleviate it to some extent by driving a pair of such phase
>>> detectors so that their outputs are in quadrature.
>>> One just selects the phase detector output that is in the linear
> range.
>>> The quadrature outputs also allow unambiguous assignment of the sign
> of
>>> any phase change.
>>>
>>
>> The Symmetricom 5120A does something very clever to alleviate this
>> problem. Explained in US patent 7,227,346 and "Direct-Digital
> Phase-Noise
>> Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger, W. Solbrig,
> and
>> S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics, and
>> Frequency Control Joint 50th Anniversary Conference, pages 287-291.
>>
>> Joe
>>
>>
> I've read the patent.
>
> Bruce
>
>
> _______________________________________________
> time-nuts mailing list -- time-nuts@febo.com
> To unsubscribe, go to
> https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
> and follow the instructions there.
>
>
> _______________________________________________
> time-nuts mailing list -- time-nuts@febo.com
> To unsubscribe, go to
> https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
> and follow the instructions there.
>
>
>
>
>
> _______________________________________________
> time-nuts mailing list -- time-nuts@febo.com
> To unsubscribe, go to
> https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
> and follow the instructions there.
>
>
> _______________________________________________
> time-nuts mailing list -- time-nuts@febo.com
> To unsubscribe, go to https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
> and follow the instructions there.
>
--
Steve Rooke - ZL3TUV & G8KVD
Omnium finis imminet
W
WB6BNQ
Thu, Dec 11, 2008 5:42 AM
Mike,
The biggest problem with Bruce's postings is that he does not leave any damn
space between the OLD data and his NEW data. So you spend an unusual amount of
time mentally separating what is going on. Very frustrating !
Bill....WB6BNQ
Mike Feher wrote:
I am now, and actually have been, at the point where I just do not read
bottom line post/replies. Bruce has a lot of good information to share,
but, now, if I click on a post, and do not immediately see a response it
is just deleted. Maybe it will be my loss, but, technology as well as
the internet is evolving, and bottom line replies totally suck. - Mike
Mike B. Feher
EOZ Inc.
89 Arnold Blvd.
Howell, NJ, 07731
732-886-5960
908-902-3831 - cell
-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com] On
Behalf Of Bruce Griffiths
Sent: Wednesday, December 10, 2008 8:38 PM
To: Discussion of precise time and frequency measurement
Subject: Re: [time-nuts] Sub Pico Second Phase logger
Joe
Joseph M Gwinn wrote:
Reflecting the sum frequency back into the mixer is actually
to reduce the noise at the IF port.
I believe that one of Agilent's simulation application notes mentions
this effect but I don't recall the actual application note number.
This will affect the mixer RF and IF port impedance so adding a
resistor may be required to improve the SWR.
How big an effect is this? Is the absolute noise decreased, or does
remain the same while the signal increase?
With the same difference frequency IF port termination impedance, noise
is actually decreased along with the mixer conversion loss.
However if the sound card input noise dominates reducing the mixer
effective output noise wont help.
If I'm understanding Walls and Stein (paper 112) correctly, the
is because with the capacitor load the beatnote waveform approaches
square, thus increasing the zero-crossing speed and therefor the phase
sensitivity. This is no doubt true, but the question was if this also
caused a small everything-dependent phase shift, something that would
have mattered in the measurement of phase noise. The object of paper
was to remedy a 10 to 20 dB error in phase noise measurements. The
critical words are in the lower left column of page 337, in the
beginning "If the mixer is terminated ...".
Saturating the RF port has a similar effect.
If one is time stamping the zero crossings an increased zero crossing
slope is an advantage.
For relative phase measurements a trapezoidal beat frequency waveform
may be less useful.
MiniCircuits AN-41-001 "FAQ about Phase Detectors" has on page 2 a
resistor to ground and a 5000 ohm resistor to the first filter
so the capacitor is isolated from the IF port by the resistors.
I wouldn't take too much notice of that recommendation as I have
confidence in the author's experience/knowledge.
Well, OK, but:
Stephen Kurtz says the same thing on the third column of the third
Off course with a capacitive IF port termination matching the RF and LO
ports becomes more critical as does the reverse isolation of the various
amplifiers driving the RF and LO ports.
It may be simpler in fact to use a level 17 mixer with high LO to RF and
LO to IF isolation with the RF port unsaturated as it relaxes the
reverse isolation specs for the isolation amplifiers.
Nelson and Walls (paper 971), Figure 4, also shows the low pass filter
arranged to absorb the sum signal, not allowing it to be reflected
Supposedly an SRA-1, but some caution is in order as some
to the effect of the input offset of an opamp based IF preamp in
same application note were of dubious veracity unless one
inverting opamp input stage.
This issue was mentioned in another app note, but their main issue
appeared to be that the opamp bias currents could cause an offset.
But the circuit they suggest has no effect on bias current induced
offset, the same current flows into the mixer and termination
independent of the series resistance.
You're right that the proposed remedy didn't make sense. I don't know
that this is a big problem with modern opamps, especially FET input
The only configuration for which it makes any sense is an inverting
input amplifier with a finite input voltage offset.
Yes, I should have said that when the 2 input signals are in
any capacitive crosstalk will have little effect on the phase shift.
Ah. Because the capacitor coupling adds a second 90 degree shift,
bringing the total to 180 degrees.
But crosstalk by ground coupling will be unaffected. As will
transformer action. Those boards are pretty crowded.
Yes its better to measure it rather than relying too much on conjecture.
The AP192 has a somewhat higher interchannel isolation than that, the
interchannel crosstalk spec is about -120dB.
With a sufficiently large number of samples the its easy to see
artifacts as low as -140dBFS.
Yep. Seems like a very good card.
It's hard to find such Firewire systems without such unnecessary
(for this application) as high gain preamps.
The AP192 has high-level inputs, but I don't know if this bypasses the
preamps, or attenuates. Given their target market, I'd bet it
There are no preamps other than an external differential input amplifier
that translates the 4 Vrms FS inputs at the input connector to a level
that the ADC can handle.
The ADC chip itself has no preamps built in.
There have been numerous complaint about this by some audio nuts,
however for this application not having such amplifiers is ideal.
The gain tempco and linearity of some variable gain audio preamps is
somewhat suspect.
I would think that none of these cards has a good tempco of anything,
given the lack of necessity in their market.
I would think that linearity would be quite good, given the horsepower
competitions on linearity.
Since the 2 ADCs share the same reference their gain tracking tempco
should be quite good given that they use capacitors rather than
resistors within the ADCs.
Other cards using AKM 24 bit ADCs should also be suitable.
Thanks. I'll look into their data.
20 Log[ 2^24 ] = 144 dB, so something else will be the limit.
Actual ENOB ~ 19 to 20 bits.
Makes sense. 20 Log [ 2^19 ] = 114 dB. Still plenty good enough.
Ideally an external sound card with balanced XLR inputs would be
HP produced a number of different phase comparators each with a
different type of phase detector.
OK. And the PLL folk must have a million designs.
Can alleviate it to some extent by driving a pair of such phase
detectors so that their outputs are in quadrature.
One just selects the phase detector output that is in the linear
The quadrature outputs also allow unambiguous assignment of the sign
The Symmetricom 5120A does something very clever to alleviate this
problem. Explained in US patent 7,227,346 and "Direct-Digital
Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger, W. Solbrig,
S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics, and
Frequency Control Joint 50th Anniversary Conference, pages 287-291.
Joe
Mike,
The biggest problem with Bruce's postings is that he does not leave any damn
space between the OLD data and his NEW data. So you spend an unusual amount of
time mentally separating what is going on. Very frustrating !
Bill....WB6BNQ
Mike Feher wrote:
> I am now, and actually have been, at the point where I just do not read
> bottom line post/replies. Bruce has a lot of good information to share,
> but, now, if I click on a post, and do not immediately see a response it
> is just deleted. Maybe it will be my loss, but, technology as well as
> the internet is evolving, and bottom line replies totally suck. - Mike
>
>
> Mike B. Feher
> EOZ Inc.
> 89 Arnold Blvd.
> Howell, NJ, 07731
> 732-886-5960
> 908-902-3831 - cell
>
>
>
> -----Original Message-----
> From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com] On
> Behalf Of Bruce Griffiths
> Sent: Wednesday, December 10, 2008 8:38 PM
> To: Discussion of precise time and frequency measurement
> Subject: Re: [time-nuts] Sub Pico Second Phase logger
>
> Joe
> Joseph M Gwinn wrote:
> > Bruce,
> >
> >
> >> Reflecting the sum frequency back into the mixer is actually
> necessary
> >> to reduce the noise at the IF port.
> >> I believe that one of Agilent's simulation application notes mentions
> >> this effect but I don't recall the actual application note number.
> >> This will affect the mixer RF and IF port impedance so adding a
> series
> >> resistor may be required to improve the SWR.
> >>
> >
> > How big an effect is this? Is the absolute noise decreased, or does
> it
> > remain the same while the signal increase?
> >
> >
> With the same difference frequency IF port termination impedance, noise
> is actually decreased along with the mixer conversion loss.
> However if the sound card input noise dominates reducing the mixer
> effective output noise wont help.
> > If I'm understanding Walls and Stein (paper 112) correctly, the
> advantage
> > is because with the capacitor load the beatnote waveform approaches
> > square, thus increasing the zero-crossing speed and therefor the phase
>
> > sensitivity. This is no doubt true, but the question was if this also
>
> > caused a small everything-dependent phase shift, something that would
> not
> > have mattered in the measurement of phase noise. The object of paper
> 112
> > was to remedy a 10 to 20 dB error in phase noise measurements. The
> > critical words are in the lower left column of page 337, in the
> paragraph
> > beginning "If the mixer is terminated ...".
> >
> >
> >
> Saturating the RF port has a similar effect.
> If one is time stamping the zero crossings an increased zero crossing
> slope is an advantage.
> For relative phase measurements a trapezoidal beat frequency waveform
> may be less useful.
> >>> MiniCircuits AN-41-001 "FAQ about Phase Detectors" has on page 2 a
> 500
> >>>
> > ohm
> >
> >>> resistor to ground and a 5000 ohm resistor to the first filter
> >>>
> > capacitor,
> >
> >>> so the capacitor is isolated from the IF port by the resistors.
> >>>
> >>>
> >>>
> >> I wouldn't take too much notice of that recommendation as I have
> little
> >> confidence in the author's experience/knowledge.
> >>
> >
> > Well, OK, but:
> >
> > Stephen Kurtz says the same thing on the third column of the third
> page, a
> > bit above Figure 6.
> >
>
> Off course with a capacitive IF port termination matching the RF and LO
> ports becomes more critical as does the reverse isolation of the various
> amplifiers driving the RF and LO ports.
> It may be simpler in fact to use a level 17 mixer with high LO to RF and
> LO to IF isolation with the RF port unsaturated as it relaxes the
> reverse isolation specs for the isolation amplifiers.
> > Nelson and Walls (paper 971), Figure 4, also shows the low pass filter
>
> > arranged to absorb the sum signal, not allowing it to be reflected
> back
> > into the mixer.
> >
> >
> >
> >>>> Supposedly an SRA-1, but some caution is in order as some
> >>>>
> >> statements as
> >>
> >>>> to the effect of the input offset of an opamp based IF preamp in
> the
> >>>> same application note were of dubious veracity unless one
> >>>>
> >> were to use an
> >>
> >>>> inverting opamp input stage.
> >>>>
> >>>>
> >>> This issue was mentioned in another app note, but their main issue
> >>> appeared to be that the opamp bias currents could cause an offset.
> >>>
> >>>
> >>>
> >>>
> >> But the circuit they suggest has no effect on bias current induced
> >> offset, the same current flows into the mixer and termination
> impedance
> >> independent of the series resistance.
> >>
> >
> > You're right that the proposed remedy didn't make sense. I don't know
>
> > that this is a big problem with modern opamps, especially FET input
> ones
> > (if needed).
> >
> >
> The only configuration for which it makes any sense is an inverting
> input amplifier with a finite input voltage offset.
> >> Yes, I should have said that when the 2 input signals are in
> quadrature,
> >> any capacitive crosstalk will have little effect on the phase shift.
> >>
> >
> > Ah. Because the capacitor coupling adds a second 90 degree shift,
> > bringing the total to 180 degrees.
> >
> > But crosstalk by ground coupling will be unaffected. As will
> crosstalk by
> > transformer action. Those boards are pretty crowded.
> >
> >
> >
> Yes its better to measure it rather than relying too much on conjecture.
> >> The AP192 has a somewhat higher interchannel isolation than that, the
> >> interchannel crosstalk spec is about -120dB.
> >> With a sufficiently large number of samples the its easy to see
> >> artifacts as low as -140dBFS.
> >>
> >
> > Yep. Seems like a very good card.
> >
> >
> >
> >
> >
> >> It's hard to find such Firewire systems without such unnecessary
> frills
> >> (for this application) as high gain preamps.
> >>
> >
> > The AP192 has high-level inputs, but I don't know if this bypasses the
>
> > preamps, or attenuates. Given their target market, I'd bet it
> bypasses.
> >
> >
> There are no preamps other than an external differential input amplifier
> that translates the 4 Vrms FS inputs at the input connector to a level
> that the ADC can handle.
> The ADC chip itself has no preamps built in.
> There have been numerous complaint about this by some audio nuts,
> however for this application not having such amplifiers is ideal.
> >
> >> The gain tempco and linearity of some variable gain audio preamps is
> >> somewhat suspect.
> >>
> >
> > I would think that none of these cards has a good tempco of anything,
> > given the lack of necessity in their market.
> >
> > I would think that linearity would be quite good, given the horsepower
>
> > competitions on linearity.
> >
> >
> Since the 2 ADCs share the same reference their gain tracking tempco
> should be quite good given that they use capacitors rather than
> resistors within the ADCs.
> >
> >>>> Other cards using AKM 24 bit ADCs should also be suitable.
> >>>>
> >>>>
> >>> Who is AKM?
> >>>
> >>>
> >>>
> >> Asahi Kasei EKM
> >>
> >> http://www.asahi-kasei.co.jp/akm/en/
> >>
> >> http://www.asahi-kasei.co.jp/akm/en/product/proaudio.html
> >>
> >>
> >
> > Thanks. I'll look into their data.
> >
> >
> >
> >>> 20 Log[ 2^24 ] = 144 dB, so something else will be the limit.
> >>>
> >>>
> >>>
> >>>
> >> Actual ENOB ~ 19 to 20 bits.
> >>
> >
> > Makes sense. 20 Log [ 2^19 ] = 114 dB. Still plenty good enough.
> >
> >
> >
> >>>> Ideally an external sound card with balanced XLR inputs would be
> >>>>
> > best.
> >
> > Yes.
> >
> >
> >
> >
> >>>> HP produced a number of different phase comparators each with a
> >>>> different type of phase detector.
> >>>>
> >
> > OK. And the PLL folk must have a million designs.
> >
> >
> >
> >>
> >> Can alleviate it to some extent by driving a pair of such phase
> >> detectors so that their outputs are in quadrature.
> >> One just selects the phase detector output that is in the linear
> range.
> >> The quadrature outputs also allow unambiguous assignment of the sign
> of
> >> any phase change.
> >>
> >
> > The Symmetricom 5120A does something very clever to alleviate this
> > problem. Explained in US patent 7,227,346 and "Direct-Digital
> Phase-Noise
> > Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger, W. Solbrig,
> and
> > S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics, and
> > Frequency Control Joint 50th Anniversary Conference, pages 287-291.
> >
> > Joe
> >
> >
> I've read the patent.
>
> Bruce
>
> _______________________________________________
> time-nuts mailing list -- time-nuts@febo.com
> To unsubscribe, go to
> https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
> and follow the instructions there.
>
> _______________________________________________
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> To unsubscribe, go to https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
> and follow the instructions there.
BG
Bruce Griffiths
Thu, Dec 11, 2008 8:53 PM
Joe
I suspect that the phase detector characteristics stated in the NIST
papers only apply when the mixer RF port is saturated.
This is evident from the Kurtz application note:
http://www.wj.com/archive/documents/Tech_Notes_Archived/Mixers_phase_detectors.pdf
which indicates (Figure 14 and Figure 15 plus surrounding text) that
for a high impedance (resistive) IF port termination the phase detection
characteristic only approaches a triangular wave when the RF port is
saturated (Figure 15) whereas for an unsaturated RF port (Figure 14) the
phase detection characteristics still appears sinusoidal.
The Kurtz application note also indicates that the IF port signal
amplitude reaches a maximum when the IF port termination resistance
increases (for the particular mixer) above 400 ohms.
There are no corresponding figures for the case of an IF port terminated
in a capacitor.
It would be interesting to check this.
Saturating the RF port also degrades the isolation etc, thus another
interesting question is does capacitively terminating the IF port
degrade these parameters when the RF port is unsaturated?
This may well not be the case with 5MHz or 10MHz mixer input frequencies.
Bruce
Joe
I suspect that the phase detector characteristics stated in the NIST
papers only apply when the mixer RF port is saturated.
This is evident from the Kurtz application note:
http://www.wj.com/archive/documents/Tech_Notes_Archived/Mixers_phase_detectors.pdf
which indicates (Figure 14 and Figure 15 plus surrounding text) that
for a high impedance (resistive) IF port termination the phase detection
characteristic only approaches a triangular wave when the RF port is
saturated (Figure 15) whereas for an unsaturated RF port (Figure 14) the
phase detection characteristics still appears sinusoidal.
The Kurtz application note also indicates that the IF port signal
amplitude reaches a maximum when the IF port termination resistance
increases (for the particular mixer) above 400 ohms.
There are no corresponding figures for the case of an IF port terminated
in a capacitor.
It would be interesting to check this.
Saturating the RF port also degrades the isolation etc, thus another
interesting question is does capacitively terminating the IF port
degrade these parameters when the RF port is unsaturated?
This may well not be the case with 5MHz or 10MHz mixer input frequencies.
Bruce
BC
Brooke Clarke
Thu, Dec 11, 2008 9:09 PM
Hi Bruce:
A general comments on mixers.
Since they are very nonlinear devices the output consists of signals at:
+/-m * RF +/-n * LO.
The output will change if the termination on any of the ports at any of those
frequencies is changed. How much it changes depends on how strong that signal
is. Some mixers reflect the image frequency to improve the conversion loss of
the desired output.
For the mixers I was working with the LO needed to be strong enough to drive
the diodes into saturation and the RF needed to be small enough to not effect
the LO power.
More on that at:
http://www.prc68.com/I/Diodes.html
Have Fun,
Brooke Clarke
http://www.prc68.com
Bruce Griffiths wrote:
Joe
I suspect that the phase detector characteristics stated in the NIST
papers only apply when the mixer RF port is saturated.
This is evident from the Kurtz application note:
http://www.wj.com/archive/documents/Tech_Notes_Archived/Mixers_phase_detectors.pdf
which indicates (Figure 14 and Figure 15 plus surrounding text) that
for a high impedance (resistive) IF port termination the phase detection
characteristic only approaches a triangular wave when the RF port is
saturated (Figure 15) whereas for an unsaturated RF port (Figure 14) the
phase detection characteristics still appears sinusoidal.
The Kurtz application note also indicates that the IF port signal
amplitude reaches a maximum when the IF port termination resistance
increases (for the particular mixer) above 400 ohms.
There are no corresponding figures for the case of an IF port terminated
in a capacitor.
It would be interesting to check this.
Saturating the RF port also degrades the isolation etc, thus another
interesting question is does capacitively terminating the IF port
degrade these parameters when the RF port is unsaturated?
This may well not be the case with 5MHz or 10MHz mixer input frequencies.
Bruce
time-nuts mailing list -- time-nuts@febo.com
To unsubscribe, go to https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
and follow the instructions there.
Hi Bruce:
A general comments on mixers.
Since they are very nonlinear devices the output consists of signals at:
+/-m * RF +/-n * LO.
The output will change if the termination on any of the ports at any of those
frequencies is changed. How much it changes depends on how strong that signal
is. Some mixers reflect the image frequency to improve the conversion loss of
the desired output.
For the mixers I was working with the LO needed to be strong enough to drive
the diodes into saturation and the RF needed to be small enough to not effect
the LO power.
More on that at:
http://www.prc68.com/I/Diodes.html
Have Fun,
Brooke Clarke
http://www.prc68.com
Bruce Griffiths wrote:
> Joe
>
> I suspect that the phase detector characteristics stated in the NIST
> papers only apply when the mixer RF port is saturated.
> This is evident from the Kurtz application note:
>
> http://www.wj.com/archive/documents/Tech_Notes_Archived/Mixers_phase_detectors.pdf
>
> which indicates (Figure 14 and Figure 15 plus surrounding text) that
> for a high impedance (resistive) IF port termination the phase detection
> characteristic only approaches a triangular wave when the RF port is
> saturated (Figure 15) whereas for an unsaturated RF port (Figure 14) the
> phase detection characteristics still appears sinusoidal.
> The Kurtz application note also indicates that the IF port signal
> amplitude reaches a maximum when the IF port termination resistance
> increases (for the particular mixer) above 400 ohms.
>
> There are no corresponding figures for the case of an IF port terminated
> in a capacitor.
> It would be interesting to check this.
>
> Saturating the RF port also degrades the isolation etc, thus another
> interesting question is does capacitively terminating the IF port
> degrade these parameters when the RF port is unsaturated?
> This may well not be the case with 5MHz or 10MHz mixer input frequencies.
>
>
> Bruce
>
>
>
> _______________________________________________
> time-nuts mailing list -- time-nuts@febo.com
> To unsubscribe, go to https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
> and follow the instructions there.
>
>
BG
Bruce Griffiths
Thu, Dec 11, 2008 9:43 PM
Brooke
The NIST papers concerned were about using mixers as phase detectors:
http://tf.nist.gov/timefreq/general/pdf/112.pdf
http://tf.nist.gov/timefreq/general/pdf/971.pdf
As is all too often the case with some NIST papers the operating conditions for which the stated phase detection characteristics are true are not specified.
Its probably a case of over familiarity with the subject and forgetting
that what's obvious to the author isn't necessarily obvious to the reader.
Since the Minicircuits phase detectors (RPD, MPD series etc) are
specified for operation with a 500 ohm resistive IF termination, and
they have relatively high RF port to RF port isolation changing the RF
port termination from 50 ohms (at least with low frequency mixers using
conventional transformers) doesn't necessarily degrade the port to port
isolation significantly.
Bruce
Brooke Clarke wrote:
Hi Bruce:
A general comments on mixers.
Since they are very nonlinear devices the output consists of signals at:
+/-m * RF +/-n * LO.
The output will change if the termination on any of the ports at any of those
frequencies is changed. How much it changes depends on how strong that signal
is. Some mixers reflect the image frequency to improve the conversion loss of
the desired output.
For the mixers I was working with the LO needed to be strong enough to drive
the diodes into saturation and the RF needed to be small enough to not effect
the LO power.
More on that at:
http://www.prc68.com/I/Diodes.html
Have Fun,
Brooke Clarke
http://www.prc68.com
Bruce Griffiths wrote:
Joe
I suspect that the phase detector characteristics stated in the NIST
papers only apply when the mixer RF port is saturated.
This is evident from the Kurtz application note:
http://www.wj.com/archive/documents/Tech_Notes_Archived/Mixers_phase_detectors.pdf
which indicates (Figure 14 and Figure 15 plus surrounding text) that
for a high impedance (resistive) IF port termination the phase detection
characteristic only approaches a triangular wave when the RF port is
saturated (Figure 15) whereas for an unsaturated RF port (Figure 14) the
phase detection characteristics still appears sinusoidal.
The Kurtz application note also indicates that the IF port signal
amplitude reaches a maximum when the IF port termination resistance
increases (for the particular mixer) above 400 ohms.
There are no corresponding figures for the case of an IF port terminated
in a capacitor.
It would be interesting to check this.
Saturating the RF port also degrades the isolation etc, thus another
interesting question is does capacitively terminating the IF port
degrade these parameters when the RF port is unsaturated?
This may well not be the case with 5MHz or 10MHz mixer input frequencies.
Bruce
time-nuts mailing list -- time-nuts@febo.com
To unsubscribe, go to https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
and follow the instructions there.
Brooke
The NIST papers concerned were about using mixers as phase detectors:
http://tf.nist.gov/timefreq/general/pdf/112.pdf
http://tf.nist.gov/timefreq/general/pdf/971.pdf
As is all too often the case with some NIST papers the operating conditions for which the stated phase detection characteristics are true are not specified.
Its probably a case of over familiarity with the subject and forgetting
that what's obvious to the author isn't necessarily obvious to the reader.
Since the Minicircuits phase detectors (RPD, MPD series etc) are
specified for operation with a 500 ohm resistive IF termination, and
they have relatively high RF port to RF port isolation changing the RF
port termination from 50 ohms (at least with low frequency mixers using
conventional transformers) doesn't necessarily degrade the port to port
isolation significantly.
Bruce
Brooke Clarke wrote:
> Hi Bruce:
>
> A general comments on mixers.
>
> Since they are very nonlinear devices the output consists of signals at:
> +/-m * RF +/-n * LO.
> The output will change if the termination on any of the ports at any of those
> frequencies is changed. How much it changes depends on how strong that signal
> is. Some mixers reflect the image frequency to improve the conversion loss of
> the desired output.
>
> For the mixers I was working with the LO needed to be strong enough to drive
> the diodes into saturation and the RF needed to be small enough to not effect
> the LO power.
>
> More on that at:
> http://www.prc68.com/I/Diodes.html
>
> Have Fun,
>
> Brooke Clarke
> http://www.prc68.com
>
> Bruce Griffiths wrote:
>
>> Joe
>>
>> I suspect that the phase detector characteristics stated in the NIST
>> papers only apply when the mixer RF port is saturated.
>> This is evident from the Kurtz application note:
>>
>> http://www.wj.com/archive/documents/Tech_Notes_Archived/Mixers_phase_detectors.pdf
>>
>> which indicates (Figure 14 and Figure 15 plus surrounding text) that
>> for a high impedance (resistive) IF port termination the phase detection
>> characteristic only approaches a triangular wave when the RF port is
>> saturated (Figure 15) whereas for an unsaturated RF port (Figure 14) the
>> phase detection characteristics still appears sinusoidal.
>> The Kurtz application note also indicates that the IF port signal
>> amplitude reaches a maximum when the IF port termination resistance
>> increases (for the particular mixer) above 400 ohms.
>>
>> There are no corresponding figures for the case of an IF port terminated
>> in a capacitor.
>> It would be interesting to check this.
>>
>> Saturating the RF port also degrades the isolation etc, thus another
>> interesting question is does capacitively terminating the IF port
>> degrade these parameters when the RF port is unsaturated?
>> This may well not be the case with 5MHz or 10MHz mixer input frequencies.
>>
>>
>> Bruce
>>
>>
>>
>> _______________________________________________
>> time-nuts mailing list -- time-nuts@febo.com
>> To unsubscribe, go to https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
>> and follow the instructions there.
>>
>>
>>
>
>
> _______________________________________________
> time-nuts mailing list -- time-nuts@febo.com
> To unsubscribe, go to https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
> and follow the instructions there.
>
>
JM
Joseph M Gwinn
Thu, Dec 11, 2008 10:19 PM
Joe
Joseph M Gwinn wrote:
Reflecting the sum frequency back into the mixer is actually
to reduce the noise at the IF port.
I believe that one of Agilent's simulation application notes mentions
this effect but I don't recall the actual application note number.
This will affect the mixer RF and IF port impedance so adding a
resistor may be required to improve the SWR.
How big an effect is this? Is the absolute noise decreased, or does
remain the same while the signal increase?
With the same difference frequency IF port termination impedance, noise
is actually decreased along with the mixer conversion loss.
OK. Complicated beasts, those mixers. Do you know of a paper (or book)
on the subject?
However if the sound card input noise dominates, reducing the mixer
effective output noise won't help.
Yes. In the plots you posted in a different email, there was a big rise
below 1 KHz (scan stopped at 1 KHz, so don't know the shape). Why is
this?
If I'm understanding Walls and Stein (paper 112) correctly, the
is because with the capacitor load the beatnote waveform approaches
square, thus increasing the zero-crossing speed and therefor the phase
sensitivity. This is no doubt true, but the question was if this also
caused a small everything-dependent phase shift, something that would
have mattered in the measurement of phase noise. The object of paper
was to remedy a 10 to 20 dB error in phase noise measurements. The
critical words are in the lower left column of page 337, in the
beginning "If the mixer is terminated ...".
Saturating the RF port has a similar effect.
Yes. But there are tradeoffs pushing the other way.
If one is time stamping the zero crossings an increased zero-crossing
For relative phase measurements a trapezoidal beat frequency waveform
may be less useful.
Fitting to the approximate waveshape, sine or trapezoidal, should yield a
very robust estimate, due to the large data support, and zero-crossing
slope won't much matter. Hmm. Actually, if the slopes of the trapezoid
are too steep, we may not have all that many slope samples.
[snip]
Of course with a capacitive IF port termination, matching the RF and LO
ports becomes more critical as does the reverse isolation of the various
amplifiers driving the RF and LO ports.
It may be simpler in fact to use a level 17 mixer with high LO to RF and
LO to IF isolation with the RF port unsaturated as it relaxes the
reverse isolation specs for the isolation amplifiers.
Another tradeoff. I'll have to think about it.
I'm thinking of 6 db and 10 db attenuators on the LO and RF ports
respectively, but no isolation amplifier.
[snip]
The only configuration for which it makes any sense is an inverting
input amplifier with a finite input voltage offset.
Why would non-inverting not work? Both inputs source or sink bias
currents, and non-inverting presents a very high impedance.
It's hard to find such Firewire systems without such unnecessary
(for this application) as high gain preamps.
The AP192 has high-level inputs, but I don't know if this bypasses the
preamps, or attenuates. Given their target market, I'd bet it
There are no preamps other than an external differential input amplifier
that translates the 4 Vrms FS inputs at the input connector to a level
that the ADC can handle.
The ADC chip itself has no preamps built in.
There have been numerous complaint about this by some audio nuts,
however for this application not having such amplifiers is ideal.
The gain tempco and linearity of some variable gain audio preamps is
somewhat suspect.
I would think that none of these cards has a good tempco of anything,
given the lack of necessity in their market.
I would think that linearity would be quite good, given the horsepower
competitions on linearity.
Since the 2 ADCs share the same reference their gain tracking tempco
should be quite good given that they use capacitors rather than
resistors within the ADCs.
A happy accident, but we'll take it.
We are converging on a soundcard wishlist:
-
True balanced inputs on XLR connectors. And good ground design, so we
aren't bedeviled by ground loops.
-
24-bit ADCs, and similar DACs.
-
Very good isolation all around.
-
Digital access via firewire (or USB3 I suppose), with the soundcard in
its own box.
-
High-level input direct to the ADCs.
While use of AKM ICs may be a very good idea, it is not a requirement per
se.
[snip]
Can alleviate [oddities at end of phase range} to some extent by
a pair of such phase detectors so that their outputs are in
One just selects the phase detector output that is in the linear
The quadrature outputs also allow unambiguous assignment of the sign
The Symmetricom 5120A does something very clever to alleviate this
problem. Explained in US patent 7,227,346 and "Direct-Digital
Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger, W. Solbrig,
S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics, and
Frequency Control Joint 50th Anniversary Conference, pages 287-291.
Joe
The paper is also worthwhile, and available on the web somewhere (don't
recall where, but google found the pdf). I had to read the patent
multiple times to figure out what's going on. The correlation approach is
old as the hills, and only the digital phase detector was patentable.
Joe
Bruce,
time-nuts-bounces@febo.com wrote on 12/10/2008 08:38:13 PM:
> Joe
> Joseph M Gwinn wrote:
> > Bruce,
> >
> >
> >> Reflecting the sum frequency back into the mixer is actually
necessary
> >> to reduce the noise at the IF port.
> >> I believe that one of Agilent's simulation application notes mentions
> >> this effect but I don't recall the actual application note number.
> >> This will affect the mixer RF and IF port impedance so adding a
series
> >> resistor may be required to improve the SWR.
> >>
> >
> > How big an effect is this? Is the absolute noise decreased, or does
it
> > remain the same while the signal increase?
> >
> >
> With the same difference frequency IF port termination impedance, noise
> is actually decreased along with the mixer conversion loss.
OK. Complicated beasts, those mixers. Do you know of a paper (or book)
on the subject?
> However if the sound card input noise dominates, reducing the mixer
> effective output noise won't help.
Yes. In the plots you posted in a different email, there was a big rise
below 1 KHz (scan stopped at 1 KHz, so don't know the shape). Why is
this?
> > If I'm understanding Walls and Stein (paper 112) correctly, the
advantage
> > is because with the capacitor load the beatnote waveform approaches
> > square, thus increasing the zero-crossing speed and therefor the phase
> > sensitivity. This is no doubt true, but the question was if this also
> > caused a small everything-dependent phase shift, something that would
not
> > have mattered in the measurement of phase noise. The object of paper
112
> > was to remedy a 10 to 20 dB error in phase noise measurements. The
> > critical words are in the lower left column of page 337, in the
paragraph
> > beginning "If the mixer is terminated ...".
> >
> >
> >
> Saturating the RF port has a similar effect.
Yes. But there are tradeoffs pushing the other way.
> If one is time stamping the zero crossings an increased zero-crossing
slope is an advantage.
> For relative phase measurements a trapezoidal beat frequency waveform
may be less useful.
Fitting to the approximate waveshape, sine or trapezoidal, should yield a
very robust estimate, due to the large data support, and zero-crossing
slope won't much matter. Hmm. Actually, if the slopes of the trapezoid
are too steep, we may not have all that many slope samples.
[snip]
>
> Of course with a capacitive IF port termination, matching the RF and LO
> ports becomes more critical as does the reverse isolation of the various
> amplifiers driving the RF and LO ports.
> It may be simpler in fact to use a level 17 mixer with high LO to RF and
> LO to IF isolation with the RF port unsaturated as it relaxes the
> reverse isolation specs for the isolation amplifiers.
Another tradeoff. I'll have to think about it.
I'm thinking of 6 db and 10 db attenuators on the LO and RF ports
respectively, but no isolation amplifier.
[snip]
> >
> >
> The only configuration for which it makes any sense is an inverting
> input amplifier with a finite input voltage offset.
Why would non-inverting not work? Both inputs source or sink bias
currents, and non-inverting presents a very high impedance.
> >
> >> It's hard to find such Firewire systems without such unnecessary
frills
> >> (for this application) as high gain preamps.
> >>
> >
> > The AP192 has high-level inputs, but I don't know if this bypasses the
> > preamps, or attenuates. Given their target market, I'd bet it
bypasses.
> >
> >
> There are no preamps other than an external differential input amplifier
> that translates the 4 Vrms FS inputs at the input connector to a level
> that the ADC can handle.
> The ADC chip itself has no preamps built in.
> There have been numerous complaint about this by some audio nuts,
> however for this application not having such amplifiers is ideal.
Bingo! Good to know.
> >> The gain tempco and linearity of some variable gain audio preamps is
> >> somewhat suspect.
> >>
> >
> > I would think that none of these cards has a good tempco of anything,
> > given the lack of necessity in their market.
> >
> > I would think that linearity would be quite good, given the horsepower
> > competitions on linearity.
> >
> >
> Since the 2 ADCs share the same reference their gain tracking tempco
> should be quite good given that they use capacitors rather than
> resistors within the ADCs.
A happy accident, but we'll take it.
We are converging on a soundcard wishlist:
1. True balanced inputs on XLR connectors. And good ground design, so we
aren't bedeviled by ground loops.
2. 24-bit ADCs, and similar DACs.
3. Very good isolation all around.
4. Digital access via firewire (or USB3 I suppose), with the soundcard in
its own box.
5. High-level input direct to the ADCs.
While use of AKM ICs may be a very good idea, it is not a requirement per
se.
[snip]
> >> Can alleviate [oddities at end of phase range} to some extent by
driving
> >> a pair of such phase detectors so that their outputs are in
quadrature.
> >> One just selects the phase detector output that is in the linear
range.
> >> The quadrature outputs also allow unambiguous assignment of the sign
of
> >> any phase change.
> >>
> >
> > The Symmetricom 5120A does something very clever to alleviate this
> > problem. Explained in US patent 7,227,346 and "Direct-Digital
Phase-Noise
> > Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger, W. Solbrig,
and
> > S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics, and
> > Frequency Control Joint 50th Anniversary Conference, pages 287-291.
> >
> > Joe
> >
> >
> I've read the patent.
The paper is also worthwhile, and available on the web somewhere (don't
recall where, but google found the pdf). I had to read the patent
multiple times to figure out what's going on. The correlation approach is
old as the hills, and only the digital phase detector was patentable.
Joe
JM
John Miles
Thu, Dec 11, 2008 10:41 PM
The Symmetricom 5120A does something very clever to alleviate this
problem. Explained in US patent 7,227,346 and "Direct-Digital
Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger, W. Solbrig,
S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics, and
Frequency Control Joint 50th Anniversary Conference, pages 287-291.
Joe
The paper is also worthwhile, and available on the web somewhere (don't
recall where, but google found the pdf). I had to read the patent
multiple times to figure out what's going on. The correlation
approach is
old as the hills, and only the digital phase detector was patentable.
If you find a link to the Grove paper that's not behind an IEEE paywall,
please post it. I'd like to read that one.
-- john, KE5FX
> > > The Symmetricom 5120A does something very clever to alleviate this
> > > problem. Explained in US patent 7,227,346 and "Direct-Digital
> Phase-Noise
> > > Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger, W. Solbrig,
> and
> > > S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics, and
> > > Frequency Control Joint 50th Anniversary Conference, pages 287-291.
> > >
> > > Joe
> > >
> > >
> > I've read the patent.
>
> The paper is also worthwhile, and available on the web somewhere (don't
> recall where, but google found the pdf). I had to read the patent
> multiple times to figure out what's going on. The correlation
> approach is
> old as the hills, and only the digital phase detector was patentable.
If you find a link to the Grove paper that's not behind an IEEE paywall,
please post it. I'd like to read that one.
-- john, KE5FX
BG
Bruce Griffiths
Fri, Dec 12, 2008 12:44 AM
Joe
Isolation from mixer RF to LO port may be too low when the mixer input
frequencies are different.
Injection locking can then occur all too easily (just ask Ulrich about
this) when the mixer RF ports are driven by 2 separate OCXOs.
Detailed in line post follows:
Bruce
Joseph M Gwinn wrote:
Joe
Joseph M Gwinn wrote:
Reflecting the sum frequency back into the mixer is actually
to reduce the noise at the IF port.
I believe that one of Agilent's simulation application notes mentions
this effect but I don't recall the actual application note number.
This will affect the mixer RF and IF port impedance so adding a
resistor may be required to improve the SWR.
How big an effect is this? Is the absolute noise decreased, or does
remain the same while the signal increase?
With the same difference frequency IF port termination impedance, noise
is actually decreased along with the mixer conversion loss.
OK. Complicated beasts, those mixers. Do you know of a paper (or book)
on the subject?
Not offhand, but this crops up in lots of places usually when one least
expects it..
However if the sound card input noise dominates, reducing the mixer
effective output noise won't help.
Yes. In the plots you posted in a different email, there was a big rise
below 1 KHz (scan stopped at 1 KHz, so don't know the shape). Why is
this?
I'll expand the frequency scale and take another snapshot for the region
below 1kHz.
This rise may be due to ADC and/or input differential amplifier flicker
noise.
If I'm understanding Walls and Stein (paper 112) correctly, the
is because with the capacitor load the beatnote waveform approaches
square, thus increasing the zero-crossing speed and therefor the phase
sensitivity. This is no doubt true, but the question was if this also
caused a small everything-dependent phase shift, something that would
have mattered in the measurement of phase noise. The object of paper
was to remedy a 10 to 20 dB error in phase noise measurements. The
critical words are in the lower left column of page 337, in the
beginning "If the mixer is terminated ...".
Saturating the RF port has a similar effect.
Yes. But there are tradeoffs pushing the other way.
If one is time stamping the zero crossings an increased zero-crossing
For relative phase measurements a trapezoidal beat frequency waveform
may be less useful.
Fitting to the approximate waveshape, sine or trapezoidal, should yield a
very robust estimate, due to the large data support, and zero-crossing
slope won't much matter. Hmm. Actually, if the slopes of the trapezoid
are too steep, we may not have all that many slope samples.
If one believes the NIST papers the trapezoid zero crossing slope only
increases by a factor of 3.
If one uses a cascaded filter limiter the slope gain can be adjusted for
optimum results.
Of course with a capacitive IF port termination, matching the RF and LO
ports becomes more critical as does the reverse isolation of the various
amplifiers driving the RF and LO ports.
It may be simpler in fact to use a level 17 mixer with high LO to RF and
LO to IF isolation with the RF port unsaturated as it relaxes the
reverse isolation specs for the isolation amplifiers.
Another tradeoff. I'll have to think about it.
I'm thinking of 6 db and 10 db attenuators on the LO and RF ports
respectively, but no isolation amplifier.
You may get away with that if you use mixers with very high RF to LO
port isolation.
Minicircuits have at least 3 level 17 mixer models that typically have
80dB LO to RF isolation at 10MHz.
Using a passive splitter for the LO drives will gain at least another
30dB in isolation between the 2 RF inputs if you use an appropriate
splitter.
The only configuration for which it makes any sense is an inverting
input amplifier with a finite input voltage offset.
Why would non-inverting not work? Both inputs source or sink bias
currents, and non-inverting presents a very high impedance.
Non inverting amplifiers usually have lower noise and generally work
very well.
I was only trying to come up with a preamp circuit for which the
comments in the Minicircuits application note on the effect of amplifier
input offset voltage made any sense.
The only risk with a noninverting amplifier, is that under fault
conditions (missing supply) a very large current can flow back (with
some low noise opamps as Enrico has experienced) into the mixers and
destroy them.
For this particular application the mixer preamp gain need only be
sufficient to boost the mixer phase detector output (1V pk?, 350mV pk??
depends on mixer and its operating conditions) to the sound card input
(FSR ~ 5.6V pk for an AP192). The resultant preamp gain is relatively
low ( 5 - 15X depending on the mixer etc) and the sound card noise will
dominate (~ 100nV/rtHz midband for an AP192) thus using an ultra low
noise mixer preamp isnt necessary.
It's hard to find such Firewire systems without such unnecessary
(for this application) as high gain preamps.
The AP192 has high-level inputs, but I don't know if this bypasses the
preamps, or attenuates. Given their target market, I'd bet it
There are no preamps other than an external differential input amplifier
that translates the 4 Vrms FS inputs at the input connector to a level
that the ADC can handle.
The ADC chip itself has no preamps built in.
There have been numerous complaint about this by some audio nuts,
however for this application not having such amplifiers is ideal.
The gain tempco and linearity of some variable gain audio preamps is
somewhat suspect.
I would think that none of these cards has a good tempco of anything,
given the lack of necessity in their market.
I would think that linearity would be quite good, given the horsepower
competitions on linearity.
Since the 2 ADCs share the same reference their gain tracking tempco
should be quite good given that they use capacitors rather than
resistors within the ADCs.
A happy accident, but we'll take it.
We are converging on a soundcard wishlist:
-
True balanced inputs on XLR connectors. And good ground design, so we
aren't bedeviled by ground loops.
-
24-bit ADCs, and similar DACs.
-
Very good isolation all around.
-
Digital access via firewire (or USB3 I suppose), with the soundcard in
its own box.
-
High-level input direct to the ADCs.
While use of AKM ICs may be a very good idea, it is not a requirement per
se.
[snip]
Optical isolation of the ADC from the noisy digital interface to the PC
would also be nice.
If we design our own PCB then the AD7760 series ADCs are another
possible option.
These have a built in differential input differential output amplifier.
Can alleviate [oddities at end of phase range} to some extent by
a pair of such phase detectors so that their outputs are in
One just selects the phase detector output that is in the linear
The quadrature outputs also allow unambiguous assignment of the sign
The Symmetricom 5120A does something very clever to alleviate this
problem. Explained in US patent 7,227,346 and "Direct-Digital
Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger, W. Solbrig,
S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics, and
Frequency Control Joint 50th Anniversary Conference, pages 287-291.
Joe
The paper is also worthwhile, and available on the web somewhere (don't
recall where, but google found the pdf). I had to read the patent
multiple times to figure out what's going on. The correlation approach is
old as the hills, and only the digital phase detector was patentable.
It may be feasible to achieve the same effect by purely digital means at
least for low sample rates where FIR filters with tens of thousands of
taps are feasible.
Of course 64 bit or higher precision arithmetic is then mandatory to
avoid excessive calculation roundoff noise.
Joe
Isolation from mixer RF to LO port may be too low when the mixer input
frequencies are different.
Injection locking can then occur all too easily (just ask Ulrich about
this) when the mixer RF ports are driven by 2 separate OCXOs.
Detailed in line post follows:
Bruce
Joseph M Gwinn wrote:
> Bruce,
>
>
> time-nuts-bounces@febo.com wrote on 12/10/2008 08:38:13 PM:
>
>
>> Joe
>> Joseph M Gwinn wrote:
>>
>>> Bruce,
>>>
>>>
>>>
>>>> Reflecting the sum frequency back into the mixer is actually
>>>>
> necessary
>
>>>> to reduce the noise at the IF port.
>>>> I believe that one of Agilent's simulation application notes mentions
>>>> this effect but I don't recall the actual application note number.
>>>> This will affect the mixer RF and IF port impedance so adding a
>>>>
> series
>
>>>> resistor may be required to improve the SWR.
>>>>
>>>>
>>> How big an effect is this? Is the absolute noise decreased, or does
>>>
> it
>
>>> remain the same while the signal increase?
>>>
>>>
>>>
>> With the same difference frequency IF port termination impedance, noise
>> is actually decreased along with the mixer conversion loss.
>>
>
> OK. Complicated beasts, those mixers. Do you know of a paper (or book)
> on the subject?
>
>
Not offhand, but this crops up in lots of places usually when one least
expects it..
>> However if the sound card input noise dominates, reducing the mixer
>> effective output noise won't help.
>>
>
> Yes. In the plots you posted in a different email, there was a big rise
> below 1 KHz (scan stopped at 1 KHz, so don't know the shape). Why is
> this?
>
>
>
I'll expand the frequency scale and take another snapshot for the region
below 1kHz.
This rise may be due to ADC and/or input differential amplifier flicker
noise.
>>> If I'm understanding Walls and Stein (paper 112) correctly, the
>>>
> advantage
>
>>> is because with the capacitor load the beatnote waveform approaches
>>> square, thus increasing the zero-crossing speed and therefor the phase
>>>
>
>
>>> sensitivity. This is no doubt true, but the question was if this also
>>>
>
>
>>> caused a small everything-dependent phase shift, something that would
>>>
> not
>
>>> have mattered in the measurement of phase noise. The object of paper
>>>
> 112
>
>>> was to remedy a 10 to 20 dB error in phase noise measurements. The
>>> critical words are in the lower left column of page 337, in the
>>>
> paragraph
>
>>> beginning "If the mixer is terminated ...".
>>>
>>>
>>>
>>>
>> Saturating the RF port has a similar effect.
>>
>
> Yes. But there are tradeoffs pushing the other way.
>
>
>
>> If one is time stamping the zero crossings an increased zero-crossing
>>
> slope is an advantage.
>
>> For relative phase measurements a trapezoidal beat frequency waveform
>>
> may be less useful.
>
> Fitting to the approximate waveshape, sine or trapezoidal, should yield a
> very robust estimate, due to the large data support, and zero-crossing
> slope won't much matter. Hmm. Actually, if the slopes of the trapezoid
> are too steep, we may not have all that many slope samples.
>
>
>
If one believes the NIST papers the trapezoid zero crossing slope only
increases by a factor of 3.
If one uses a cascaded filter limiter the slope gain can be adjusted for
optimum results.
> [snip]
>
>> Of course with a capacitive IF port termination, matching the RF and LO
>> ports becomes more critical as does the reverse isolation of the various
>> amplifiers driving the RF and LO ports.
>> It may be simpler in fact to use a level 17 mixer with high LO to RF and
>> LO to IF isolation with the RF port unsaturated as it relaxes the
>> reverse isolation specs for the isolation amplifiers.
>>
>
> Another tradeoff. I'll have to think about it.
>
> I'm thinking of 6 db and 10 db attenuators on the LO and RF ports
> respectively, but no isolation amplifier.
>
>
You may get away with that if you use mixers with very high RF to LO
port isolation.
Minicircuits have at least 3 level 17 mixer models that typically have
80dB LO to RF isolation at 10MHz.
Using a passive splitter for the LO drives will gain at least another
30dB in isolation between the 2 RF inputs if you use an appropriate
splitter.
> [snip]
>
>>>
>> The only configuration for which it makes any sense is an inverting
>> input amplifier with a finite input voltage offset.
>>
>
> Why would non-inverting not work? Both inputs source or sink bias
> currents, and non-inverting presents a very high impedance.
>
>
>
>
Non inverting amplifiers usually have lower noise and generally work
very well.
I was only trying to come up with a preamp circuit for which the
comments in the Minicircuits application note on the effect of amplifier
input offset voltage made any sense.
The only risk with a noninverting amplifier, is that under fault
conditions (missing supply) a very large current can flow back (with
some low noise opamps as Enrico has experienced) into the mixers and
destroy them.
For this particular application the mixer preamp gain need only be
sufficient to boost the mixer phase detector output (1V pk?, 350mV pk??
depends on mixer and its operating conditions) to the sound card input
(FSR ~ 5.6V pk for an AP192). The resultant preamp gain is relatively
low ( 5 - 15X depending on the mixer etc) and the sound card noise will
dominate (~ 100nV/rtHz midband for an AP192) thus using an ultra low
noise mixer preamp isnt necessary.
>>>> It's hard to find such Firewire systems without such unnecessary
>>>>
> frills
>
>>>> (for this application) as high gain preamps.
>>>>
>>>>
>>> The AP192 has high-level inputs, but I don't know if this bypasses the
>>>
>
>
>>> preamps, or attenuates. Given their target market, I'd bet it
>>>
> bypasses.
>
>>>
>> There are no preamps other than an external differential input amplifier
>> that translates the 4 Vrms FS inputs at the input connector to a level
>> that the ADC can handle.
>> The ADC chip itself has no preamps built in.
>> There have been numerous complaint about this by some audio nuts,
>> however for this application not having such amplifiers is ideal.
>>
>
> Bingo! Good to know.
>
>
>
>>>> The gain tempco and linearity of some variable gain audio preamps is
>>>> somewhat suspect.
>>>>
>>>>
>>> I would think that none of these cards has a good tempco of anything,
>>> given the lack of necessity in their market.
>>>
>>> I would think that linearity would be quite good, given the horsepower
>>>
>
>
>>> competitions on linearity.
>>>
>>>
>>>
>> Since the 2 ADCs share the same reference their gain tracking tempco
>> should be quite good given that they use capacitors rather than
>> resistors within the ADCs.
>>
>
> A happy accident, but we'll take it.
>
>
> We are converging on a soundcard wishlist:
>
> 1. True balanced inputs on XLR connectors. And good ground design, so we
> aren't bedeviled by ground loops.
>
> 2. 24-bit ADCs, and similar DACs.
>
> 3. Very good isolation all around.
>
> 4. Digital access via firewire (or USB3 I suppose), with the soundcard in
> its own box.
>
> 5. High-level input direct to the ADCs.
>
>
> While use of AKM ICs may be a very good idea, it is not a requirement per
> se.
>
> [snip]
>
Optical isolation of the ADC from the noisy digital interface to the PC
would also be nice.
If we design our own PCB then the AD7760 series ADCs are another
possible option.
These have a built in differential input differential output amplifier.
>>>> Can alleviate [oddities at end of phase range} to some extent by
>>>>
> driving
>
>>>> a pair of such phase detectors so that their outputs are in
>>>>
> quadrature.
>
>>>> One just selects the phase detector output that is in the linear
>>>>
> range.
>
>>>> The quadrature outputs also allow unambiguous assignment of the sign
>>>>
> of
>
>>>> any phase change.
>>>>
>>>>
>>> The Symmetricom 5120A does something very clever to alleviate this
>>> problem. Explained in US patent 7,227,346 and "Direct-Digital
>>>
> Phase-Noise
>
>>> Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger, W. Solbrig,
>>>
> and
>
>>> S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics, and
>>> Frequency Control Joint 50th Anniversary Conference, pages 287-291.
>>>
>>> Joe
>>>
>>>
>>>
>> I've read the patent.
>>
>
> The paper is also worthwhile, and available on the web somewhere (don't
> recall where, but google found the pdf). I had to read the patent
> multiple times to figure out what's going on. The correlation approach is
> old as the hills, and only the digital phase detector was patentable.
>
>
>
It may be feasible to achieve the same effect by purely digital means at
least for low sample rates where FIR filters with tens of thousands of
taps are feasible.
Of course 64 bit or higher precision arithmetic is then mandatory to
avoid excessive calculation roundoff noise.
> Joe
>
>
BG
Bruce Griffiths
Fri, Dec 12, 2008 1:36 AM
Joe
Attached is noise spectrum (1kHz and below) of AP192 with nothing
connected to inputs.
Sampling rate 96KSPS.
Frequency bin equivalent noise bandwidth ~ 3Hz.
Noise has similar spectrum to flicker noise with a noise corner of
around 300Hz or so.
Bruce
Joe
Attached is noise spectrum (1kHz and below) of AP192 with nothing
connected to inputs.
Sampling rate 96KSPS.
Frequency bin equivalent noise bandwidth ~ 3Hz.
Noise has similar spectrum to flicker noise with a noise corner of
around 300Hz or so.
Bruce
JM
Joseph M Gwinn
Mon, Dec 15, 2008 3:25 PM
Joe
Attached is noise spectrum (1kHz and below) of AP192 with nothing
connected to inputs.
Sampling rate 96KSPS.
Frequency bin equivalent noise bandwidth ~ 3Hz.
Noise has similar spectrum to flicker noise with a noise corner of
around 300Hz or so.
The noise floor ain't so bad, -130 dB (dBm? dbV?) at maybe 10 Hz, even if
it's 20 dB worse than at 1 KHz.
Has anyone measured the Allan Deviation?
Joe
Bruce,
time-nuts-bounces@febo.com wrote on 12/11/2008 08:36:47 PM:
> Joe
>
> Attached is noise spectrum (1kHz and below) of AP192 with nothing
> connected to inputs.
> Sampling rate 96KSPS.
> Frequency bin equivalent noise bandwidth ~ 3Hz.
> Noise has similar spectrum to flicker noise with a noise corner of
> around 300Hz or so.
The noise floor ain't so bad, -130 dB (dBm? dbV?) at maybe 10 Hz, even if
it's 20 dB worse than at 1 KHz.
Has anyone measured the Allan Deviation?
Joe
JM
Joseph M Gwinn
Mon, Dec 15, 2008 4:16 PM
Joe
Isolation from mixer RF to LO port may be too low when the mixer input
frequencies are different.
Injection locking can then occur all too easily (just ask Ulrich about
this) when the mixer RF ports are driven by 2 separate OCXOs.
Detailed in line post follows:
Bruce
Joseph M Gwinn wrote:
Joe
Joseph M Gwinn wrote:
Reflecting the sum frequency back into the mixer is actually
to reduce the noise at the IF port.
I believe that one of Agilent's simulation application notes
this effect but I don't recall the actual application note number.
This will affect the mixer RF and IF port impedance so adding a
series resistor may be required to improve the SWR.
How big an effect is this? Is the absolute noise decreased, or does
it remain the same while the signal increase?
With the same difference frequency IF port termination impedance,
is actually decreased along with the mixer conversion loss.
OK. Complicated beasts, those mixers. Do you know of a paper (or
Not offhand, but this crops up in lots of places usually when one least
expects it..
I've noticed. Someone has to have poured his soul into a monograph.
However if the sound card input noise dominates, reducing the mixer
effective output noise won't help.
Yes. In the plots you posted in a different email, there was a big
below 1 KHz (scan stopped at 1 KHz, so don't know the shape). Why is
this?
I'll expand the frequency scale and take another snapshot for the region
below 1kHz.
This rise may be due to ADC and/or input differential amplifier flicker
noise.
Saw it. Thanks. Does look like flicker noise. Although it wasn't large
enough to be a real problem it seems.
If one is time stamping the zero crossings an increased zero-crossing
slope is an advantage.
For relative phase measurements a trapezoidal beat frequencywaveform
may be less useful.
Fitting to the approximate waveshape, sine or trapezoidal,
should yield a very robust estimate, due to the large data support,
slope won't much matter. Hmm. Actually, if the slopes of the
are too steep, we may not have all that many slope samples.
If one believes the NIST papers the trapezoid zero crossing slope only
increases by a factor of 3.
If one uses a cascaded filter limiter the slope gain can be adjusted for
optimum results.
The implication would be to not sharpen things up too well in before
digitizing.
Of course with a capacitive IF port termination, matching the RF and
ports becomes more critical as does the reverse isolation of the
amplifiers driving the RF and LO ports.
It may be simpler in fact to use a level 17 mixer with high LO to RF
LO to IF isolation with the RF port unsaturated as it relaxes the
reverse isolation specs for the isolation amplifiers.
Another tradeoff. I'll have to think about it.
I'm thinking of 6 db and 10 db attenuators on the LO and RF ports
respectively, but no isolation amplifier.
You may get away with that if you use mixers with very high RF to LO
port isolation.
Minicircuits have at least 3 level 17 mixer models that typically have
80dB LO to RF isolation at 10MHz.
I've used the ZRPD-1, which claims about 75 dB isolation at 10 MHz.
Using a passive splitter for the LO drives will gain at least another
30dB in isolation between the 2 RF inputs if you use an appropriate
splitter.
True.
I may have lost the thread here. If we have one oscillator driving
everything, one cannot have injection locking even if isolation isn't
perfect. What isolation does gain us is a reduction in undesired phase
shifta.
The only configuration for which it makes any sense is an inverting
input amplifier with a finite input voltage offset.
Why would non-inverting not work? Both inputs source or sink bias
currents, and non-inverting presents a very high impedance.
Non inverting amplifiers usually have lower noise and generally work
I was only trying to come up with a preamp circuit for which the
comments in the Minicircuits application note on the effect of amplifier
input offset voltage made any sense.
Ah. It may be hopeless.
My reading was that they were worried about bias currents from the amp
flowing into the mixer and causing offsets, not amplifier offset voltages
per se. The amplifier offset voltage does not cause a mixer offset, and
may be reduced by use of a chopper amp or very good balance.
The only risk with a noninverting amplifier, is that under fault
conditions (missing supply) a very large current can flow back (with
some low noise opamps as Enrico has experienced) into the mixers and
destroy them.
Yes, I recall the discussion on your website.
For this particular application the mixer preamp gain need only be
sufficient to boost the mixer phase detector output (1V pk?, 350mV pk??
depends on mixer and its operating conditions) to the sound card input
(FSR ~ 5.6V pk for an AP192). The resultant preamp gain is relatively
low ( 5 - 15X depending on the mixer etc) and the sound card noise will
dominate (~ 100nV/rtHz midband for an AP192) thus using an ultra low
noise mixer preamp isn't necessary.
We are converging on a soundcard wishlist:
- True balanced inputs on XLR connectors. And good ground design,
aren't bedeviled by ground loops.
-
24-bit ADCs, and similar DACs.
-
Very good isolation all around.
-
Digital access via firewire (or USB3 I suppose), with the
its own box.
- High-level input direct to the ADCs.
While use of AKM ICs may be a very good idea, it is not a requirement
Optical isolation of the ADC from the noisy digital interface to the PC
would also be nice.
Good point. Part of ground design I suppose. Although with noise floor
of -150 dB down there cannot be so much leakage.
If we design our own PCB then the AD7760 series ADCs are another
possible option.
These have a built in differential input differential output amplifier.
Yes. But aren't we trying to use commonly available soundcards?
Can alleviate [oddities at end of phase range} to some extent by
driving a pair of such phase detectors so that their outputs are in
One just selects the phase detector output that is in the linear
range.
The quadrature outputs also allow unambiguous assignment of the
The Symmetricom 5120A does something very clever to alleviate this
problem. Explained in US patent 7,227,346 and "Direct-Digital
Phase-Noise Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger,
and S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics,
Frequency Control Joint 50th Anniversary Conference, pages 287-291.
Joe
The paper is also worthwhile, and available on the web somewhere
recall where, but google found the pdf). I had to read the patent
multiple times to figure out what's going on. The correlation
old as the hills, and only the digital phase detector was patentable.
It may be feasible to achieve the same effect by purely digital means at
least for low sample rates where FIR filters with tens of thousands of
taps are feasible.
It is feasible, and Sam Stein is doing it. I've perhaps lost the thread
here.
Of course 64 bit or higher precision arithmetic is then mandatory to
avoid excessive calculation roundoff noise.
It may be 64 bit integer, actually.
My understanding is that the 5120A is built upon a DSP or more likely FPGA
(of unspecified make and model). The 5125A will have a top frequency of
400 MHz, so the DSP and/or FPGA better be damn fast. Little analog stuff
remains.
Joe
Bruce,
time-nuts-bounces@febo.com wrote on 12/11/2008 07:44:01 PM:
> Joe
>
> Isolation from mixer RF to LO port may be too low when the mixer input
> frequencies are different.
> Injection locking can then occur all too easily (just ask Ulrich about
> this) when the mixer RF ports are driven by 2 separate OCXOs.
>
> Detailed in line post follows:
>
> Bruce
>
> Joseph M Gwinn wrote:
> > Bruce,
> >
> >
> > time-nuts-bounces@febo.com wrote on 12/10/2008 08:38:13 PM:
> >
> >
> >> Joe
> >> Joseph M Gwinn wrote:
> >>
> >>> Bruce,
> >>>
> >>>
> >>>
> >>>> Reflecting the sum frequency back into the mixer is actually
necessary
> >>>> to reduce the noise at the IF port.
> >>>> I believe that one of Agilent's simulation application notes
mentions
> >>>> this effect but I don't recall the actual application note number.
> >>>> This will affect the mixer RF and IF port impedance so adding a
> >>>> series resistor may be required to improve the SWR.
> >>>>
> >>>>
> >>> How big an effect is this? Is the absolute noise decreased, or does
> >>> it remain the same while the signal increase?
> >>>
> >>>
> >>>
> >> With the same difference frequency IF port termination impedance,
noise
> >> is actually decreased along with the mixer conversion loss.
> >>
> >
> > OK. Complicated beasts, those mixers. Do you know of a paper (or
book)
> > on the subject?
> >
> >
>
>
> Not offhand, but this crops up in lots of places usually when one least
> expects it..
I've noticed. Someone has to have poured his soul into a monograph.
> >> However if the sound card input noise dominates, reducing the mixer
> >> effective output noise won't help.
> >>
> >
> > Yes. In the plots you posted in a different email, there was a big
rise
> > below 1 KHz (scan stopped at 1 KHz, so don't know the shape). Why is
> > this?
> >
> I'll expand the frequency scale and take another snapshot for the region
> below 1kHz.
> This rise may be due to ADC and/or input differential amplifier flicker
> noise.
Saw it. Thanks. Does look like flicker noise. Although it wasn't large
enough to be a real problem it seems.
> >
> >> If one is time stamping the zero crossings an increased zero-crossing
> >> slope is an advantage.
> >> For relative phase measurements a trapezoidal beat frequencywaveform
> >> may be less useful.
> >
> > Fitting to the approximate waveshape, sine or trapezoidal,
> > should yield a very robust estimate, due to the large data support,
and zero-crossing
> > slope won't much matter. Hmm. Actually, if the slopes of the
trapezoid
> > are too steep, we may not have all that many slope samples.
> >
> >
> >
>
> If one believes the NIST papers the trapezoid zero crossing slope only
> increases by a factor of 3.
> If one uses a cascaded filter limiter the slope gain can be adjusted for
> optimum results.
The implication would be to not sharpen things up too well in before
digitizing.
> > [snip]
> >
> >> Of course with a capacitive IF port termination, matching the RF and
LO
> >> ports becomes more critical as does the reverse isolation of the
various
> >> amplifiers driving the RF and LO ports.
> >> It may be simpler in fact to use a level 17 mixer with high LO to RF
and
> >> LO to IF isolation with the RF port unsaturated as it relaxes the
> >> reverse isolation specs for the isolation amplifiers.
> >>
> >
> > Another tradeoff. I'll have to think about it.
> >
> > I'm thinking of 6 db and 10 db attenuators on the LO and RF ports
> > respectively, but no isolation amplifier.
>
> You may get away with that if you use mixers with very high RF to LO
> port isolation.
> Minicircuits have at least 3 level 17 mixer models that typically have
> 80dB LO to RF isolation at 10MHz.
I've used the ZRPD-1, which claims about 75 dB isolation at 10 MHz.
> Using a passive splitter for the LO drives will gain at least another
> 30dB in isolation between the 2 RF inputs if you use an appropriate
> splitter.
True.
I may have lost the thread here. If we have one oscillator driving
everything, one cannot have injection locking even if isolation isn't
perfect. What isolation does gain us is a reduction in undesired phase
shifta.
> > [snip]
> >
> >>>
> >> The only configuration for which it makes any sense is an inverting
> >> input amplifier with a finite input voltage offset.
> >>
> >
> > Why would non-inverting not work? Both inputs source or sink bias
> > currents, and non-inverting presents a very high impedance.
>
> Non inverting amplifiers usually have lower noise and generally work
very well.
> I was only trying to come up with a preamp circuit for which the
> comments in the Minicircuits application note on the effect of amplifier
> input offset voltage made any sense.
Ah. It may be hopeless.
My reading was that they were worried about bias currents from the amp
flowing into the mixer and causing offsets, not amplifier offset voltages
per se. The amplifier offset voltage does not cause a mixer offset, and
may be reduced by use of a chopper amp or very good balance.
> The only risk with a noninverting amplifier, is that under fault
> conditions (missing supply) a very large current can flow back (with
> some low noise opamps as Enrico has experienced) into the mixers and
> destroy them.
Yes, I recall the discussion on your website.
> For this particular application the mixer preamp gain need only be
> sufficient to boost the mixer phase detector output (1V pk?, 350mV pk??
> depends on mixer and its operating conditions) to the sound card input
> (FSR ~ 5.6V pk for an AP192). The resultant preamp gain is relatively
> low ( 5 - 15X depending on the mixer etc) and the sound card noise will
> dominate (~ 100nV/rtHz midband for an AP192) thus using an ultra low
> noise mixer preamp isn't necessary.
Yes.
> > We are converging on a soundcard wishlist:
> >
> > 1. True balanced inputs on XLR connectors. And good ground design,
so we
> > aren't bedeviled by ground loops.
> >
> > 2. 24-bit ADCs, and similar DACs.
> >
> > 3. Very good isolation all around.
> >
> > 4. Digital access via firewire (or USB3 I suppose), with the
soundcard in
> > its own box.
> >
> > 5. High-level input direct to the ADCs.
> >
> >
> > While use of AKM ICs may be a very good idea, it is not a requirement
per
> > se.
> >
> > [snip]
> >
>
>
> Optical isolation of the ADC from the noisy digital interface to the PC
> would also be nice.
Good point. Part of ground design I suppose. Although with noise floor
of -150 dB down there cannot be so much leakage.
> If we design our own PCB then the AD7760 series ADCs are another
> possible option.
> These have a built in differential input differential output amplifier.
Yes. But aren't we trying to use commonly available soundcards?
>
> >>>> Can alleviate [oddities at end of phase range} to some extent by
> >>>> driving a pair of such phase detectors so that their outputs are in
> >>>> quadrature.
> >
> >>>> One just selects the phase detector output that is in the linear
> >>>> range.
> >
> >>>> The quadrature outputs also allow unambiguous assignment of the
sign
> >>>> of any phase change.
> >>>>
> >>>>
> >>> The Symmetricom 5120A does something very clever to alleviate this
> >>> problem. Explained in US patent 7,227,346 and "Direct-Digital
> >>> Phase-Noise Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger,
W.Solbrig,
> >>> and S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics,
and
> >>> Frequency Control Joint 50th Anniversary Conference, pages 287-291.
> >>>
> >>> Joe
> >>>
> >>>
> >>>
> >> I've read the patent.
> >>
> >
> > The paper is also worthwhile, and available on the web somewhere
(don't
> > recall where, but google found the pdf). I had to read the patent
> > multiple times to figure out what's going on. The correlation
approach is
> > old as the hills, and only the digital phase detector was patentable.
> >
>
> It may be feasible to achieve the same effect by purely digital means at
> least for low sample rates where FIR filters with tens of thousands of
> taps are feasible.
It *is* feasible, and Sam Stein is doing it. I've perhaps lost the thread
here.
> Of course 64 bit or higher precision arithmetic is then mandatory to
> avoid excessive calculation roundoff noise.
It may be 64 bit *integer*, actually.
My understanding is that the 5120A is built upon a DSP or more likely FPGA
(of unspecified make and model). The 5125A will have a top frequency of
400 MHz, so the DSP and/or FPGA better be damn fast. Little analog stuff
remains.
Joe
W
wa1zms@att.net
Mon, Dec 15, 2008 4:53 PM
Looking for comment here...
The background:
I'm working on a sub mm-wave LO chain for
a ham radio application. While chasing issues
of close-in phase (ie: within 1KHz of RF
carrier) by peeling the "layers of the onion",
I'm starting to question the performance of
the MMICs that are used as buffers and amps
following my Wenzel reference OCXOs.
Question(s):
Should any MMIC be allowed to be driven
close to compression or into compression
when striving for best close-in noise?
I know and have seen the NF of a MMIC
degrade while in compression, but my
target right now is close-in noise rather
than broadband noise.
My design, in summary, takes 5MHz up to 630GHz
via several multipliers and PLL stages.
-Brian
Looking for comment here...
The background:
I'm working on a sub mm-wave LO chain for
a ham radio application. While chasing issues
of close-in phase (ie: within 1KHz of RF
carrier) by peeling the "layers of the onion",
I'm starting to question the performance of
the MMICs that are used as buffers and amps
following my Wenzel reference OCXOs.
Question(s):
Should any MMIC be allowed to be driven
close to compression or into compression
when striving for best close-in noise?
I know and have seen the NF of a MMIC
degrade while in compression, but my
target right now is close-in noise rather
than broadband noise.
My design, in summary, takes 5MHz up to 630GHz
via several multipliers and PLL stages.
-Brian
SW
Stan W1LE
Mon, Dec 15, 2008 5:19 PM
Hello Brian,
I hope you and yours have the very best of this Holiday Season.
My considerations would be similar to spurious free dynamic range,
keeping all discrete intermodulation products very low, which
would in turn keep the intermodulation noise very low.
When you drive an amp towards and into saturation, to get harmonics
generated,
the broadband noise floor will be raised with intermodulation noise.
Most of the time this intermodulation noise is not a limiting factor,
until very high harmonic multiples are needed.
Stan, W1LE FN41sr Cape Cod
wa1zms@att.net wrote:
Looking for comment here...
The background:
I'm working on a sub mm-wave LO chain for
a ham radio application. While chasing issues
of close-in phase (ie: within 1KHz of RF
carrier) by peeling the "layers of the onion",
I'm starting to question the performance of
the MMICs that are used as buffers and amps
following my Wenzel reference OCXOs.
Question(s):
Should any MMIC be allowed to be driven
close to compression or into compression
when striving for best close-in noise?
I know and have seen the NF of a MMIC
degrade while in compression, but my
target right now is close-in noise rather
than broadband noise.
My design, in summary, takes 5MHz up to 630GHz
via several multipliers and PLL stages.
-Brian
time-nuts mailing list -- time-nuts@febo.com
To unsubscribe, go to https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
and follow the instructions there.
Hello Brian,
I hope you and yours have the very best of this Holiday Season.
My considerations would be similar to spurious free dynamic range,
keeping all discrete intermodulation products very low, which
would in turn keep the intermodulation noise very low.
When you drive an amp towards and into saturation, to get harmonics
generated,
the broadband noise floor will be raised with intermodulation noise.
Most of the time this intermodulation noise is not a limiting factor,
until very high harmonic multiples are needed.
Stan, W1LE FN41sr Cape Cod
wa1zms@att.net wrote:
> Looking for comment here...
>
> The background:
> I'm working on a sub mm-wave LO chain for
> a ham radio application. While chasing issues
> of close-in phase (ie: within 1KHz of RF
> carrier) by peeling the "layers of the onion",
> I'm starting to question the performance of
> the MMICs that are used as buffers and amps
> following my Wenzel reference OCXOs.
>
> Question(s):
> Should any MMIC be allowed to be driven
> close to compression or into compression
> when striving for best close-in noise?
>
> I know and have seen the NF of a MMIC
> degrade while in compression, but my
> target right now is close-in noise rather
> than broadband noise.
>
> My design, in summary, takes 5MHz up to 630GHz
> via several multipliers and PLL stages.
>
> -Brian
>
>
>
>
>
> _______________________________________________
> time-nuts mailing list -- time-nuts@febo.com
> To unsubscribe, go to https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
> and follow the instructions there.
>
>
JM
John Miles
Mon, Dec 15, 2008 7:14 PM
The painful part is probably the first few stages, if you are starting at 5
MHz. You probably want to do some HP 8662A-like tricks using crystal
filters to shave off the broadband noise below 1 GHz, and maybe SAW filters
above that. This will do nothing for noise within 1 kHz, though... do you
really need a clean signal that close to the carrier all the way up to 630
GHz?
The noise characteristics of the MMICs seems to depend a lot on the fab
technology. I can't seem to find my .PDF copy of it right now, but I have
one paper on microwave regenerative dividers where the authors measured the
residual PN of several contemporary parts driven to saturation. At 4.5 GHz,
the 10 dB/decade corner frequency wasn't reached until past 100 kHz for the
Stanford Microdevices SGA-4186, which didn't speak well for the PN
performance of SiGe HBT parts. They showed -143 dBc/Hz at 1 kHz for that
one.
The GaAs HBT part (Mini-Circuits ERA-5SM) they tested was among the best
(-156 dBc/Hz at 1 kHz). Second-worst was an InGaP/GaAs HBT part (Stanford
NGA-489) at about -153 dBc/Hz at 1 kHz. Still much better than the SiGe
part.
My understanding is that the newer GALI-series parts from Mini-Circuits are
InGaP HBT devices so they'd presumably perform about like the NGA-489.
You'd want to measure them to make sure, though, if your app is that
critical.
Take a look at NRAO's recent publications, especially those associated with
the ALMA array (many of which are on their site). They're doing the real
bleeding-edge work at sub-mm these days.
-- john, KE5FX
-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com]On
Behalf Of wa1zms@att.net
Sent: Monday, December 15, 2008 8:54 AM
To: time-nuts@febo.com
Subject: [time-nuts] Close-in phase noise question...
Looking for comment here...
The background:
I'm working on a sub mm-wave LO chain for
a ham radio application. While chasing issues
of close-in phase (ie: within 1KHz of RF
carrier) by peeling the "layers of the onion",
I'm starting to question the performance of
the MMICs that are used as buffers and amps
following my Wenzel reference OCXOs.
Question(s):
Should any MMIC be allowed to be driven
close to compression or into compression
when striving for best close-in noise?
I know and have seen the NF of a MMIC
degrade while in compression, but my
target right now is close-in noise rather
than broadband noise.
My design, in summary, takes 5MHz up to 630GHz
via several multipliers and PLL stages.
-Brian
The painful part is probably the first few stages, if you are starting at 5
MHz. You probably want to do some HP 8662A-like tricks using crystal
filters to shave off the broadband noise below 1 GHz, and maybe SAW filters
above that. This will do nothing for noise within 1 kHz, though... do you
really need a clean signal that close to the carrier all the way up to 630
GHz?
The noise characteristics of the MMICs seems to depend a lot on the fab
technology. I can't seem to find my .PDF copy of it right now, but I have
one paper on microwave regenerative dividers where the authors measured the
residual PN of several contemporary parts driven to saturation. At 4.5 GHz,
the 10 dB/decade corner frequency wasn't reached until past 100 kHz for the
Stanford Microdevices SGA-4186, which didn't speak well for the PN
performance of SiGe HBT parts. They showed -143 dBc/Hz at 1 kHz for that
one.
The GaAs HBT part (Mini-Circuits ERA-5SM) they tested was among the best
(-156 dBc/Hz at 1 kHz). Second-worst was an InGaP/GaAs HBT part (Stanford
NGA-489) at about -153 dBc/Hz at 1 kHz. Still much better than the SiGe
part.
My understanding is that the newer GALI-series parts from Mini-Circuits are
InGaP HBT devices so they'd presumably perform about like the NGA-489.
You'd want to measure them to make sure, though, if your app is that
critical.
Take a look at NRAO's recent publications, especially those associated with
the ALMA array (many of which are on their site). They're doing the real
bleeding-edge work at sub-mm these days.
-- john, KE5FX
> -----Original Message-----
> From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com]On
> Behalf Of wa1zms@att.net
> Sent: Monday, December 15, 2008 8:54 AM
> To: time-nuts@febo.com
> Subject: [time-nuts] Close-in phase noise question...
>
>
> Looking for comment here...
>
> The background:
> I'm working on a sub mm-wave LO chain for
> a ham radio application. While chasing issues
> of close-in phase (ie: within 1KHz of RF
> carrier) by peeling the "layers of the onion",
> I'm starting to question the performance of
> the MMICs that are used as buffers and amps
> following my Wenzel reference OCXOs.
>
> Question(s):
> Should any MMIC be allowed to be driven
> close to compression or into compression
> when striving for best close-in noise?
>
> I know and have seen the NF of a MMIC
> degrade while in compression, but my
> target right now is close-in noise rather
> than broadband noise.
>
> My design, in summary, takes 5MHz up to 630GHz
> via several multipliers and PLL stages.
>
> -Brian
>
>
JL
J. L. Trantham, M. D.
Mon, Dec 15, 2008 7:56 PM
I really enjoy reading the mail on this group, but I thought it was the
'front molecule on the cutting edge'.
Joe
-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com]On
Behalf Of John Miles
Sent: Monday, December 15, 2008 1:15 PM
To: Discussion of precise time and frequency measurement
Subject: Re: [time-nuts] Close-in phase noise question...
The painful part is probably the first few stages, if you are starting at 5
MHz. You probably want to do some HP 8662A-like tricks using crystal
filters to shave off the broadband noise below 1 GHz, and maybe SAW filters
above that. This will do nothing for noise within 1 kHz, though... do you
really need a clean signal that close to the carrier all the way up to 630
GHz?
The noise characteristics of the MMICs seems to depend a lot on the fab
technology. I can't seem to find my .PDF copy of it right now, but I have
one paper on microwave regenerative dividers where the authors measured the
residual PN of several contemporary parts driven to saturation. At 4.5 GHz,
the 10 dB/decade corner frequency wasn't reached until past 100 kHz for the
Stanford Microdevices SGA-4186, which didn't speak well for the PN
performance of SiGe HBT parts. They showed -143 dBc/Hz at 1 kHz for that
one.
The GaAs HBT part (Mini-Circuits ERA-5SM) they tested was among the best
(-156 dBc/Hz at 1 kHz). Second-worst was an InGaP/GaAs HBT part (Stanford
NGA-489) at about -153 dBc/Hz at 1 kHz. Still much better than the SiGe
part.
My understanding is that the newer GALI-series parts from Mini-Circuits are
InGaP HBT devices so they'd presumably perform about like the NGA-489.
You'd want to measure them to make sure, though, if your app is that
critical.
Take a look at NRAO's recent publications, especially those associated with
the ALMA array (many of which are on their site). They're doing the real
bleeding-edge work at sub-mm these days.
-- john, KE5FX
-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com]On
Behalf Of wa1zms@att.net
Sent: Monday, December 15, 2008 8:54 AM
To: time-nuts@febo.com
Subject: [time-nuts] Close-in phase noise question...
Looking for comment here...
The background:
I'm working on a sub mm-wave LO chain for
a ham radio application. While chasing issues
of close-in phase (ie: within 1KHz of RF
carrier) by peeling the "layers of the onion",
I'm starting to question the performance of
the MMICs that are used as buffers and amps
following my Wenzel reference OCXOs.
Question(s):
Should any MMIC be allowed to be driven
close to compression or into compression
when striving for best close-in noise?
I know and have seen the NF of a MMIC
degrade while in compression, but my
target right now is close-in noise rather
than broadband noise.
My design, in summary, takes 5MHz up to 630GHz
via several multipliers and PLL stages.
-Brian
I really enjoy reading the mail on this group, but I thought it was the
'front molecule on the cutting edge'.
Joe
-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com]On
Behalf Of John Miles
Sent: Monday, December 15, 2008 1:15 PM
To: Discussion of precise time and frequency measurement
Subject: Re: [time-nuts] Close-in phase noise question...
The painful part is probably the first few stages, if you are starting at 5
MHz. You probably want to do some HP 8662A-like tricks using crystal
filters to shave off the broadband noise below 1 GHz, and maybe SAW filters
above that. This will do nothing for noise within 1 kHz, though... do you
really need a clean signal that close to the carrier all the way up to 630
GHz?
The noise characteristics of the MMICs seems to depend a lot on the fab
technology. I can't seem to find my .PDF copy of it right now, but I have
one paper on microwave regenerative dividers where the authors measured the
residual PN of several contemporary parts driven to saturation. At 4.5 GHz,
the 10 dB/decade corner frequency wasn't reached until past 100 kHz for the
Stanford Microdevices SGA-4186, which didn't speak well for the PN
performance of SiGe HBT parts. They showed -143 dBc/Hz at 1 kHz for that
one.
The GaAs HBT part (Mini-Circuits ERA-5SM) they tested was among the best
(-156 dBc/Hz at 1 kHz). Second-worst was an InGaP/GaAs HBT part (Stanford
NGA-489) at about -153 dBc/Hz at 1 kHz. Still much better than the SiGe
part.
My understanding is that the newer GALI-series parts from Mini-Circuits are
InGaP HBT devices so they'd presumably perform about like the NGA-489.
You'd want to measure them to make sure, though, if your app is that
critical.
Take a look at NRAO's recent publications, especially those associated with
the ALMA array (many of which are on their site). They're doing the real
bleeding-edge work at sub-mm these days.
-- john, KE5FX
> -----Original Message-----
> From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com]On
> Behalf Of wa1zms@att.net
> Sent: Monday, December 15, 2008 8:54 AM
> To: time-nuts@febo.com
> Subject: [time-nuts] Close-in phase noise question...
>
>
> Looking for comment here...
>
> The background:
> I'm working on a sub mm-wave LO chain for
> a ham radio application. While chasing issues
> of close-in phase (ie: within 1KHz of RF
> carrier) by peeling the "layers of the onion",
> I'm starting to question the performance of
> the MMICs that are used as buffers and amps
> following my Wenzel reference OCXOs.
>
> Question(s):
> Should any MMIC be allowed to be driven
> close to compression or into compression
> when striving for best close-in noise?
>
> I know and have seen the NF of a MMIC
> degrade while in compression, but my
> target right now is close-in noise rather
> than broadband noise.
>
> My design, in summary, takes 5MHz up to 630GHz
> via several multipliers and PLL stages.
>
> -Brian
>
>
_______________________________________________
time-nuts mailing list -- time-nuts@febo.com
To unsubscribe, go to
https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
and follow the instructions there.
W
wa1zms@att.net
Mon, Dec 15, 2008 7:58 PM
John-
I do agree with you. The pain is in the early stages of the
LO chain. Since I'm planning to use QRSS CW (that's very
slow speed Morse Code with very narrow demod bandwidths for
non-hams on this reflector who may not be familiar with QRSS),
the noise very close to the carrier become key.
A noisy carrier can't be detected very well on a waterfall
display if it too, looks like noise.
I have a several MSA-1105 MMICs in the chain to provide
isolation and give gain while multiplying the 5MHz signal
to 10MHz and then to 20MHz. The 20MHz acts as a reference
for a 1320MHz PLL. The 1320MHz then is multiplied several
more times on it's way to a sun-harmonic mixer for 630GHz.
I am wondering if the MSA-1105s could be causing more
close-in noise than I expected. The CW note sounds a
bit "rough" by ear.
The final 600GHz mixer that I'm using comes from some of
the mm-wave boys that work with NRAO.
But my LO noise requirements are a bit different than their
needs.
-------------- Original message from "John Miles" jmiles@pop.net: --------------
The painful part is probably the first few stages, if you are starting at 5
MHz. You probably want to do some HP 8662A-like tricks using crystal
filters to shave off the broadband noise below 1 GHz, and maybe SAW filters
above that. This will do nothing for noise within 1 kHz, though... do you
really need a clean signal that close to the carrier all the way up to 630
GHz?
The noise characteristics of the MMICs seems to depend a lot on the fab
technology. I can't seem to find my .PDF copy of it right now, but I have
one paper on microwave regenerative dividers where the authors measured the
residual PN of several contemporary parts driven to saturation. At 4.5 GHz,
the 10 dB/decade corner frequency wasn't reached until past 100 kHz for the
Stanford Microdevices SGA-4186, which didn't speak well for the PN
performance of SiGe HBT parts. They showed -143 dBc/Hz at 1 kHz for that
one.
The GaAs HBT part (Mini-Circuits ERA-5SM) they tested was among the best
(-156 dBc/Hz at 1 kHz). Second-worst was an InGaP/GaAs HBT part (Stanford
NGA-489) at about -153 dBc/Hz at 1 kHz. Still much better than the SiGe
part.
My understanding is that the newer GALI-series parts from Mini-Circuits are
InGaP HBT devices so they'd presumably perform about like the NGA-489.
You'd want to measure them to make sure, though, if your app is that
critical.
Take a look at NRAO's recent publications, especially those associated with
the ALMA array (many of which are on their site). They're doing the real
bleeding-edge work at sub-mm these days.
-- john, KE5FX
-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com]On
Behalf Of wa1zms@att.net
Sent: Monday, December 15, 2008 8:54 AM
To: time-nuts@febo.com
Subject: [time-nuts] Close-in phase noise question...
Looking for comment here...
The background:
I'm working on a sub mm-wave LO chain for
a ham radio application. While chasing issues
of close-in phase (ie: within 1KHz of RF
carrier) by peeling the "layers of the onion",
I'm starting to question the performance of
the MMICs that are used as buffers and amps
following my Wenzel reference OCXOs.
Question(s):
Should any MMIC be allowed to be driven
close to compression or into compression
when striving for best close-in noise?
I know and have seen the NF of a MMIC
degrade while in compression, but my
target right now is close-in noise rather
than broadband noise.
My design, in summary, takes 5MHz up to 630GHz
via several multipliers and PLL stages.
-Brian
John-
I do agree with you. The pain is in the early stages of the
LO chain. Since I'm planning to use QRSS CW (that's very
slow speed Morse Code with very narrow demod bandwidths for
non-hams on this reflector who may not be familiar with QRSS),
the noise very close to the carrier become key.
A noisy carrier can't be detected very well on a waterfall
display if it too, looks like noise.
I have a several MSA-1105 MMICs in the chain to provide
isolation and give gain while multiplying the 5MHz signal
to 10MHz and then to 20MHz. The 20MHz acts as a reference
for a 1320MHz PLL. The 1320MHz then is multiplied several
more times on it's way to a sun-harmonic mixer for 630GHz.
I am wondering if the MSA-1105s could be causing more
close-in noise than I expected. The CW note sounds a
bit "rough" by ear.
The final 600GHz mixer that I'm using comes from some of
the mm-wave boys that work with NRAO.
But my LO noise requirements are a bit different than their
needs.
-------------- Original message from "John Miles" <jmiles@pop.net>: --------------
> The painful part is probably the first few stages, if you are starting at 5
> MHz. You probably want to do some HP 8662A-like tricks using crystal
> filters to shave off the broadband noise below 1 GHz, and maybe SAW filters
> above that. This will do nothing for noise within 1 kHz, though... do you
> really need a clean signal that close to the carrier all the way up to 630
> GHz?
>
> The noise characteristics of the MMICs seems to depend a lot on the fab
> technology. I can't seem to find my .PDF copy of it right now, but I have
> one paper on microwave regenerative dividers where the authors measured the
> residual PN of several contemporary parts driven to saturation. At 4.5 GHz,
> the 10 dB/decade corner frequency wasn't reached until past 100 kHz for the
> Stanford Microdevices SGA-4186, which didn't speak well for the PN
> performance of SiGe HBT parts. They showed -143 dBc/Hz at 1 kHz for that
> one.
>
> The GaAs HBT part (Mini-Circuits ERA-5SM) they tested was among the best
> (-156 dBc/Hz at 1 kHz). Second-worst was an InGaP/GaAs HBT part (Stanford
> NGA-489) at about -153 dBc/Hz at 1 kHz. Still much better than the SiGe
> part.
>
> My understanding is that the newer GALI-series parts from Mini-Circuits are
> InGaP HBT devices so they'd presumably perform about like the NGA-489.
> You'd want to measure them to make sure, though, if your app is that
> critical.
>
> Take a look at NRAO's recent publications, especially those associated with
> the ALMA array (many of which are on their site). They're doing the real
> bleeding-edge work at sub-mm these days.
>
> -- john, KE5FX
>
> > -----Original Message-----
> > From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com]On
> > Behalf Of wa1zms@att.net
> > Sent: Monday, December 15, 2008 8:54 AM
> > To: time-nuts@febo.com
> > Subject: [time-nuts] Close-in phase noise question...
> >
> >
> > Looking for comment here...
> >
> > The background:
> > I'm working on a sub mm-wave LO chain for
> > a ham radio application. While chasing issues
> > of close-in phase (ie: within 1KHz of RF
> > carrier) by peeling the "layers of the onion",
> > I'm starting to question the performance of
> > the MMICs that are used as buffers and amps
> > following my Wenzel reference OCXOs.
> >
> > Question(s):
> > Should any MMIC be allowed to be driven
> > close to compression or into compression
> > when striving for best close-in noise?
> >
> > I know and have seen the NF of a MMIC
> > degrade while in compression, but my
> > target right now is close-in noise rather
> > than broadband noise.
> >
> > My design, in summary, takes 5MHz up to 630GHz
> > via several multipliers and PLL stages.
> >
> > -Brian
> >
>
> >
>
>
> _______________________________________________
> time-nuts mailing list -- time-nuts@febo.com
> To unsubscribe, go to https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
> and follow the instructions there.
W
wa1zms@att.net
Mon, Dec 15, 2008 8:10 PM
I would assume that the bleeding takes place
just after the first molecule has performed it's
dissecting task.
-------------- Original message from "J. L. Trantham, M. D." jltran@worldnet.att.net: --------------
I really enjoy reading the mail on this group, but I thought it was the
'front molecule on the cutting edge'.
Joe
I would assume that the bleeding takes place
just after the first molecule has performed it's
dissecting task.
-------------- Original message from "J. L. Trantham, M. D." <jltran@worldnet.att.net>: --------------
> I really enjoy reading the mail on this group, but I thought it was the
> 'front molecule on the cutting edge'.
>
> Joe
BG
Bruce Griffiths
Mon, Dec 15, 2008 9:34 PM
Joe
Attached is noise spectrum (1kHz and below) of AP192 with nothing
connected to inputs.
Sampling rate 96KSPS.
Frequency bin equivalent noise bandwidth ~ 3Hz.
Noise has similar spectrum to flicker noise with a noise corner of
around 300Hz or so.
The noise floor ain't so bad, -130 dB (dBm? dbV?) at maybe 10 Hz, even if
it's 20 dB worse than at 1 KHz.
Has anyone measured the Allan Deviation?
Joe
Joe
Noise plot is with respect to full scale (4Vrms).
I need to build a noise source to check the absolute level.
Will use the amplified Johnson noise of a 150K resistor.
By Allan deviation do you mean calculate it from the sequential 96KSPS
ADC output samples?
I can do this, but since the dominant noise source is white the Allan
deviation will scale with the measurement bandwidth.
Bruce
Joseph M Gwinn wrote:
> Bruce,
>
>
> time-nuts-bounces@febo.com wrote on 12/11/2008 08:36:47 PM:
>
>
>> Joe
>>
>> Attached is noise spectrum (1kHz and below) of AP192 with nothing
>> connected to inputs.
>> Sampling rate 96KSPS.
>> Frequency bin equivalent noise bandwidth ~ 3Hz.
>> Noise has similar spectrum to flicker noise with a noise corner of
>> around 300Hz or so.
>>
>
> The noise floor ain't so bad, -130 dB (dBm? dbV?) at maybe 10 Hz, even if
> it's 20 dB worse than at 1 KHz.
>
> Has anyone measured the Allan Deviation?
>
> Joe
>
>
>
Joe
Noise plot is with respect to full scale (4Vrms).
I need to build a noise source to check the absolute level.
Will use the amplified Johnson noise of a 150K resistor.
By Allan deviation do you mean calculate it from the sequential 96KSPS
ADC output samples?
I can do this, but since the dominant noise source is white the Allan
deviation will scale with the measurement bandwidth.
Bruce
BG
Bruce Griffiths
Mon, Dec 15, 2008 9:56 PM
Joe
Joseph M Gwinn wrote:
Joe
Isolation from mixer RF to LO port may be too low when the mixer input
frequencies are different.
Injection locking can then occur all too easily (just ask Ulrich about
this) when the mixer RF ports are driven by 2 separate OCXOs.
Detailed in line post follows:
Bruce
Joseph M Gwinn wrote:
Joe
Joseph M Gwinn wrote:
Reflecting the sum frequency back into the mixer is actually
to reduce the noise at the IF port.
I believe that one of Agilent's simulation application notes
this effect but I don't recall the actual application note number.
This will affect the mixer RF and IF port impedance so adding a
series resistor may be required to improve the SWR.
How big an effect is this? Is the absolute noise decreased, or does
it remain the same while the signal increase?
With the same difference frequency IF port termination impedance,
is actually decreased along with the mixer conversion loss.
OK. Complicated beasts, those mixers. Do you know of a paper (or
Not offhand, but this crops up in lots of places usually when one least
expects it..
I've noticed. Someone has to have poured his soul into a monograph.
However if the sound card input noise dominates, reducing the mixer
effective output noise won't help.
Yes. In the plots you posted in a different email, there was a big
below 1 KHz (scan stopped at 1 KHz, so don't know the shape). Why is
this?
I'll expand the frequency scale and take another snapshot for the region
below 1kHz.
This rise may be due to ADC and/or input differential amplifier flicker
noise.
Saw it. Thanks. Does look like flicker noise. Although it wasn't large
enough to be a real problem it seems.
If one is time stamping the zero crossings an increased zero-crossing
slope is an advantage.
For relative phase measurements a trapezoidal beat frequencywaveform
may be less useful.
Fitting to the approximate waveshape, sine or trapezoidal,
should yield a very robust estimate, due to the large data support,
slope won't much matter. Hmm. Actually, if the slopes of the
are too steep, we may not have all that many slope samples.
If one believes the NIST papers the trapezoid zero crossing slope only
increases by a factor of 3.
If one uses a cascaded filter limiter the slope gain can be adjusted for
optimum results.
The implication would be to not sharpen things up too well in before
digitizing.
Of course with a capacitive IF port termination, matching the RF and
ports becomes more critical as does the reverse isolation of the
amplifiers driving the RF and LO ports.
It may be simpler in fact to use a level 17 mixer with high LO to RF
LO to IF isolation with the RF port unsaturated as it relaxes the
reverse isolation specs for the isolation amplifiers.
Another tradeoff. I'll have to think about it.
I'm thinking of 6 db and 10 db attenuators on the LO and RF ports
respectively, but no isolation amplifier.
You may get away with that if you use mixers with very high RF to LO
port isolation.
Minicircuits have at least 3 level 17 mixer models that typically have
80dB LO to RF isolation at 10MHz.
I've used the ZRPD-1, which claims about 75 dB isolation at 10 MHz.
Using a passive splitter for the LO drives will gain at least another
30dB in isolation between the 2 RF inputs if you use an appropriate
splitter.
True.
I may have lost the thread here. If we have one oscillator driving
everything, one cannot have injection locking even if isolation isn't
perfect. What isolation does gain us is a reduction in undesired phase
shifta.
You have 2 oscillators, the test source and the offset source, however
the >= 10Hz frequency offset between them means that the isolation
requirements are relaxed considerably.
If the offset oscillator is derived from the source then injection
locking doesnt occur.
I made the general comment to ensure that anyone following the thread,
who may be contemplating building a dual mixer setup with 2 sources very
close in frequency doesnt forget about the isolation requirements.
The only configuration for which it makes any sense is an inverting
input amplifier with a finite input voltage offset.
Why would non-inverting not work? Both inputs source or sink bias
currents, and non-inverting presents a very high impedance.
Non inverting amplifiers usually have lower noise and generally work
I was only trying to come up with a preamp circuit for which the
comments in the Minicircuits application note on the effect of amplifier
input offset voltage made any sense.
Ah. It may be hopeless.
My reading was that they were worried about bias currents from the amp
flowing into the mixer and causing offsets, not amplifier offset voltages
per se. The amplifier offset voltage does not cause a mixer offset, and
may be reduced by use of a chopper amp or very good balance.
The only risk with a noninverting amplifier, is that under fault
conditions (missing supply) a very large current can flow back (with
some low noise opamps as Enrico has experienced) into the mixers and
destroy them.
Yes, I recall the discussion on your website.
For this particular application the mixer preamp gain need only be
sufficient to boost the mixer phase detector output (1V pk?, 350mV pk??
depends on mixer and its operating conditions) to the sound card input
(FSR ~ 5.6V pk for an AP192). The resultant preamp gain is relatively
low ( 5 - 15X depending on the mixer etc) and the sound card noise will
dominate (~ 100nV/rtHz midband for an AP192) thus using an ultra low
noise mixer preamp isn't necessary.
We are converging on a soundcard wishlist:
- True balanced inputs on XLR connectors. And good ground design,
aren't bedeviled by ground loops.
-
24-bit ADCs, and similar DACs.
-
Very good isolation all around.
-
Digital access via firewire (or USB3 I suppose), with the
its own box.
- High-level input direct to the ADCs.
While use of AKM ICs may be a very good idea, it is not a requirement
Optical isolation of the ADC from the noisy digital interface to the PC
would also be nice.
Good point. Part of ground design I suppose. Although with noise floor
of -150 dB down there cannot be so much leakage.
If we design our own PCB then the AD7760 series ADCs are another
possible option.
These have a built in differential input differential output amplifier.
Yes. But aren't we trying to use commonly available soundcards?
Ideally yes, but they all seem to have built in performance limitations.
AFAIK the AP192 with its 4Vrms full scale balanced inputs with no
variable gain preamps or +48V phantom supplies seems to be one of the
best for this application.
Its major drawback is that its a PCI card located within a noisy PC.
The 4V rms input allows the mixer preamp to use devices like the THAT
1646 to drive the balance sound card inputs without degrading the noise
floor too much.
With a 1V rms full scale the noise floor degradation would be very
obvious when using a THAT1646 (equivalent devices are even noisier).
It may be better to use a mixer preamp with a transformer coupled output
stage using hybrid feedback to achieve a low frequency cutoff below 1Hz
together with low noise.
Can alleviate [oddities at end of phase range} to some extent by
driving a pair of such phase detectors so that their outputs are in
quadrature.
One just selects the phase detector output that is in the linear
range.
The quadrature outputs also allow unambiguous assignment of the
The Symmetricom 5120A does something very clever to alleviate this
problem. Explained in US patent 7,227,346 and "Direct-Digital
Phase-Noise Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger,
and S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics,
Frequency Control Joint 50th Anniversary Conference, pages 287-291.
Joe
The paper is also worthwhile, and available on the web somewhere
recall where, but google found the pdf). I had to read the patent
multiple times to figure out what's going on. The correlation
old as the hills, and only the digital phase detector was patentable.
It may be feasible to achieve the same effect by purely digital means at
least for low sample rates where FIR filters with tens of thousands of
taps are feasible.
It is feasible, and Sam Stein is doing it. I've perhaps lost the thread
here.
No, I meant replace his 90 degree hybrids with a digital equivalent.
Of course 64 bit or higher precision arithmetic is then mandatory to
avoid excessive calculation roundoff noise.
It may be 64 bit integer, actually.
My understanding is that the 5120A is built upon a DSP or more likely FPGA
(of unspecified make and model). The 5125A will have a top frequency of
400 MHz, so the DSP and/or FPGA better be damn fast. Little analog stuff
remains.
Although the 5125A appeared in the 2008 product catalog, it isn't on the
website yet.
Joe
Joseph M Gwinn wrote:
> Bruce,
>
>
> time-nuts-bounces@febo.com wrote on 12/11/2008 07:44:01 PM:
>
>
>> Joe
>>
>> Isolation from mixer RF to LO port may be too low when the mixer input
>> frequencies are different.
>> Injection locking can then occur all too easily (just ask Ulrich about
>> this) when the mixer RF ports are driven by 2 separate OCXOs.
>>
>> Detailed in line post follows:
>>
>> Bruce
>>
>> Joseph M Gwinn wrote:
>>
>>> Bruce,
>>>
>>>
>>> time-nuts-bounces@febo.com wrote on 12/10/2008 08:38:13 PM:
>>>
>>>
>>>
>>>> Joe
>>>> Joseph M Gwinn wrote:
>>>>
>>>>
>>>>> Bruce,
>>>>>
>>>>>
>>>>>
>>>>>
>>>>>> Reflecting the sum frequency back into the mixer is actually
>>>>>>
> necessary
>
>>>>>> to reduce the noise at the IF port.
>>>>>> I believe that one of Agilent's simulation application notes
>>>>>>
> mentions
>
>>>>>> this effect but I don't recall the actual application note number.
>>>>>> This will affect the mixer RF and IF port impedance so adding a
>>>>>> series resistor may be required to improve the SWR.
>>>>>>
>>>>>>
>>>>>>
>>>>> How big an effect is this? Is the absolute noise decreased, or does
>>>>>
>
>
>>>>> it remain the same while the signal increase?
>>>>>
>>>>>
>>>>>
>>>>>
>>>> With the same difference frequency IF port termination impedance,
>>>>
> noise
>
>>>> is actually decreased along with the mixer conversion loss.
>>>>
>>>>
>>> OK. Complicated beasts, those mixers. Do you know of a paper (or
>>>
> book)
>
>>> on the subject?
>>>
>>>
>>>
>> Not offhand, but this crops up in lots of places usually when one least
>> expects it..
>>
>
> I've noticed. Someone has to have poured his soul into a monograph.
>
>
>
>>>> However if the sound card input noise dominates, reducing the mixer
>>>> effective output noise won't help.
>>>>
>>>>
>>> Yes. In the plots you posted in a different email, there was a big
>>>
> rise
>
>>> below 1 KHz (scan stopped at 1 KHz, so don't know the shape). Why is
>>> this?
>>>
>>>
>> I'll expand the frequency scale and take another snapshot for the region
>> below 1kHz.
>> This rise may be due to ADC and/or input differential amplifier flicker
>> noise.
>>
>
> Saw it. Thanks. Does look like flicker noise. Although it wasn't large
> enough to be a real problem it seems.
>
>
>
>
>>>> If one is time stamping the zero crossings an increased zero-crossing
>>>>
>
>
>>>> slope is an advantage.
>>>> For relative phase measurements a trapezoidal beat frequencywaveform
>>>> may be less useful.
>>>>
>>> Fitting to the approximate waveshape, sine or trapezoidal,
>>> should yield a very robust estimate, due to the large data support,
>>>
> and zero-crossing
>
>>> slope won't much matter. Hmm. Actually, if the slopes of the
>>>
> trapezoid
>
>>> are too steep, we may not have all that many slope samples.
>>>
>>>
>>>
>>>
>> If one believes the NIST papers the trapezoid zero crossing slope only
>> increases by a factor of 3.
>> If one uses a cascaded filter limiter the slope gain can be adjusted for
>> optimum results.
>>
>
> The implication would be to not sharpen things up too well in before
> digitizing.
>
>
>
>>> [snip]
>>>
>>>
>>>> Of course with a capacitive IF port termination, matching the RF and
>>>>
> LO
>
>>>> ports becomes more critical as does the reverse isolation of the
>>>>
> various
>
>>>> amplifiers driving the RF and LO ports.
>>>> It may be simpler in fact to use a level 17 mixer with high LO to RF
>>>>
> and
>
>>>> LO to IF isolation with the RF port unsaturated as it relaxes the
>>>> reverse isolation specs for the isolation amplifiers.
>>>>
>>>>
>>> Another tradeoff. I'll have to think about it.
>>>
>>> I'm thinking of 6 db and 10 db attenuators on the LO and RF ports
>>> respectively, but no isolation amplifier.
>>>
>> You may get away with that if you use mixers with very high RF to LO
>> port isolation.
>> Minicircuits have at least 3 level 17 mixer models that typically have
>> 80dB LO to RF isolation at 10MHz.
>>
>
> I've used the ZRPD-1, which claims about 75 dB isolation at 10 MHz.
>
>
>
>> Using a passive splitter for the LO drives will gain at least another
>> 30dB in isolation between the 2 RF inputs if you use an appropriate
>> splitter.
>>
>
> True.
>
> I may have lost the thread here. If we have one oscillator driving
> everything, one cannot have injection locking even if isolation isn't
> perfect. What isolation does gain us is a reduction in undesired phase
> shifta.
>
>
>
You have 2 oscillators, the test source and the offset source, however
the >= 10Hz frequency offset between them means that the isolation
requirements are relaxed considerably.
If the offset oscillator is derived from the source then injection
locking doesnt occur.
I made the general comment to ensure that anyone following the thread,
who may be contemplating building a dual mixer setup with 2 sources very
close in frequency doesnt forget about the isolation requirements.
>
>
>>> [snip]
>>>
>>>
>>>> The only configuration for which it makes any sense is an inverting
>>>> input amplifier with a finite input voltage offset.
>>>>
>>>>
>>> Why would non-inverting not work? Both inputs source or sink bias
>>> currents, and non-inverting presents a very high impedance.
>>>
>> Non inverting amplifiers usually have lower noise and generally work
>>
> very well.
>
>> I was only trying to come up with a preamp circuit for which the
>> comments in the Minicircuits application note on the effect of amplifier
>> input offset voltage made any sense.
>>
>
> Ah. It may be hopeless.
>
> My reading was that they were worried about bias currents from the amp
> flowing into the mixer and causing offsets, not amplifier offset voltages
> per se. The amplifier offset voltage does not cause a mixer offset, and
> may be reduced by use of a chopper amp or very good balance.
>
>
>
>> The only risk with a noninverting amplifier, is that under fault
>> conditions (missing supply) a very large current can flow back (with
>> some low noise opamps as Enrico has experienced) into the mixers and
>> destroy them.
>>
>
> Yes, I recall the discussion on your website.
>
>
>
>> For this particular application the mixer preamp gain need only be
>> sufficient to boost the mixer phase detector output (1V pk?, 350mV pk??
>> depends on mixer and its operating conditions) to the sound card input
>> (FSR ~ 5.6V pk for an AP192). The resultant preamp gain is relatively
>> low ( 5 - 15X depending on the mixer etc) and the sound card noise will
>> dominate (~ 100nV/rtHz midband for an AP192) thus using an ultra low
>> noise mixer preamp isn't necessary.
>>
>
> Yes.
>
>
>
>
>>> We are converging on a soundcard wishlist:
>>>
>>> 1. True balanced inputs on XLR connectors. And good ground design,
>>>
> so we
>
>>> aren't bedeviled by ground loops.
>>>
>>> 2. 24-bit ADCs, and similar DACs.
>>>
>>> 3. Very good isolation all around.
>>>
>>> 4. Digital access via firewire (or USB3 I suppose), with the
>>>
> soundcard in
>
>>> its own box.
>>>
>>> 5. High-level input direct to the ADCs.
>>>
>>>
>>> While use of AKM ICs may be a very good idea, it is not a requirement
>>>
> per
>
>>> se.
>>>
>>> [snip]
>>>
>>>
>> Optical isolation of the ADC from the noisy digital interface to the PC
>> would also be nice.
>>
>
> Good point. Part of ground design I suppose. Although with noise floor
> of -150 dB down there cannot be so much leakage.
>
>
>
>> If we design our own PCB then the AD7760 series ADCs are another
>> possible option.
>> These have a built in differential input differential output amplifier.
>>
>
> Yes. But aren't we trying to use commonly available soundcards?
>
>
Ideally yes, but they all seem to have built in performance limitations.
AFAIK the AP192 with its 4Vrms full scale balanced inputs with no
variable gain preamps or +48V phantom supplies seems to be one of the
best for this application.
Its major drawback is that its a PCI card located within a noisy PC.
The 4V rms input allows the mixer preamp to use devices like the THAT
1646 to drive the balance sound card inputs without degrading the noise
floor too much.
With a 1V rms full scale the noise floor degradation would be very
obvious when using a THAT1646 (equivalent devices are even noisier).
It may be better to use a mixer preamp with a transformer coupled output
stage using hybrid feedback to achieve a low frequency cutoff below 1Hz
together with low noise.
>
>
>>>>>> Can alleviate [oddities at end of phase range} to some extent by
>>>>>> driving a pair of such phase detectors so that their outputs are in
>>>>>>
>
>
>>>>>> quadrature.
>>>>>>
>>>>>> One just selects the phase detector output that is in the linear
>>>>>> range.
>>>>>>
>>>>>> The quadrature outputs also allow unambiguous assignment of the
>>>>>>
> sign
>
>>>>>> of any phase change.
>>>>>>
>>>>>>
>>>>>>
>>>>> The Symmetricom 5120A does something very clever to alleviate this
>>>>> problem. Explained in US patent 7,227,346 and "Direct-Digital
>>>>> Phase-Noise Measurement"; J. Grove, J. Hein, J. Retta, P. Schweiger,
>>>>>
> W.Solbrig,
>
>>>>> and S.R. Stein; 2004 IEEE International Ultrasonics, Ferroelectrics,
>>>>>
> and
>
>>>>> Frequency Control Joint 50th Anniversary Conference, pages 287-291.
>>>>>
>>>>> Joe
>>>>>
>>>>>
>>>>>
>>>>>
>>>> I've read the patent.
>>>>
>>>>
>>> The paper is also worthwhile, and available on the web somewhere
>>>
> (don't
>
>>> recall where, but google found the pdf). I had to read the patent
>>> multiple times to figure out what's going on. The correlation
>>>
> approach is
>
>>> old as the hills, and only the digital phase detector was patentable.
>>>
>>>
>> It may be feasible to achieve the same effect by purely digital means at
>> least for low sample rates where FIR filters with tens of thousands of
>> taps are feasible.
>>
>
> It *is* feasible, and Sam Stein is doing it. I've perhaps lost the thread
> here.
>
>
>
No, I meant replace his 90 degree hybrids with a digital equivalent.
>> Of course 64 bit or higher precision arithmetic is then mandatory to
>> avoid excessive calculation roundoff noise.
>>
>
> It may be 64 bit *integer*, actually.
>
> My understanding is that the 5120A is built upon a DSP or more likely FPGA
> (of unspecified make and model). The 5125A will have a top frequency of
> 400 MHz, so the DSP and/or FPGA better be damn fast. Little analog stuff
> remains.
>
>
Although the 5125A appeared in the 2008 product catalog, it isn't on the
website yet.
> Joe
>
>
Bruce
BG
Bruce Griffiths
Mon, Dec 15, 2008 9:59 PM
Looking for comment here...
The background:
I'm working on a sub mm-wave LO chain for
a ham radio application. While chasing issues
of close-in phase (ie: within 1KHz of RF
carrier) by peeling the "layers of the onion",
I'm starting to question the performance of
the MMICs that are used as buffers and amps
following my Wenzel reference OCXOs.
Question(s):
Should any MMIC be allowed to be driven
close to compression or into compression
when striving for best close-in noise?
I know and have seen the NF of a MMIC
degrade while in compression, but my
target right now is close-in noise rather
than broadband noise.
My design, in summary, takes 5MHz up to 630GHz
via several multipliers and PLL stages.
-Brian
Brian
The increased nonlinearity when driven into compression will enhance the
conversion of low frequency noise to phase noise.
Bruce
wa1zms@att.net wrote:
> Looking for comment here...
>
> The background:
> I'm working on a sub mm-wave LO chain for
> a ham radio application. While chasing issues
> of close-in phase (ie: within 1KHz of RF
> carrier) by peeling the "layers of the onion",
> I'm starting to question the performance of
> the MMICs that are used as buffers and amps
> following my Wenzel reference OCXOs.
>
> Question(s):
> Should any MMIC be allowed to be driven
> close to compression or into compression
> when striving for best close-in noise?
>
> I know and have seen the NF of a MMIC
> degrade while in compression, but my
> target right now is close-in noise rather
> than broadband noise.
>
> My design, in summary, takes 5MHz up to 630GHz
> via several multipliers and PLL stages.
>
> -Brian
>
Brian
The increased nonlinearity when driven into compression will enhance the
conversion of low frequency noise to phase noise.
Bruce
JM
Joseph M Gwinn
Mon, Dec 15, 2008 10:12 PM
Joe
Attached is noise spectrum (1kHz and below) of AP192 with nothing
connected to inputs.
Sampling rate 96KSPS.
Frequency bin equivalent noise bandwidth ~ 3Hz.
Noise has similar spectrum to flicker noise with a noise corner of
around 300Hz or so.
The noise floor ain't so bad, -130 dB (dBm? dbV?) at maybe 10 Hz, even
it's 20 dB worse than at 1 KHz.
Has anyone measured the Allan Deviation?
Joe
Joe
Noise plot is with respect to full scale (4Vrms).
I need to build a noise source to check the absolute level.
Will use the amplified Johnson noise of a 150K resistor.
By Allan deviation do you mean calculate it from the sequential 96KSPS
ADC output samples?
Yes, although some decimation may be needed to keep compute times under
control, at least for the larger values of tau.
I can do this, but since the dominant noise source is white the Allan
deviation will scale with the measurement bandwidth.
Would modified Allan deviation be better?
I'm more interested in the general shape of the Allan curve than its
absolute value, one issue being the effect of thermal variations in your
laboratory dungeon. We had speculated as to the relative size of thermal
effects in these sound cards, and this would give us some idea.
Joe
Bruce,
time-nuts-bounces@febo.com wrote on 12/15/2008 04:34:34 PM:
> Joseph M Gwinn wrote:
> > Bruce,
> >
> >
> > time-nuts-bounces@febo.com wrote on 12/11/2008 08:36:47 PM:
> >
> >
> >> Joe
> >>
> >> Attached is noise spectrum (1kHz and below) of AP192 with nothing
> >> connected to inputs.
> >> Sampling rate 96KSPS.
> >> Frequency bin equivalent noise bandwidth ~ 3Hz.
> >> Noise has similar spectrum to flicker noise with a noise corner of
> >> around 300Hz or so.
> >>
> >
> > The noise floor ain't so bad, -130 dB (dBm? dbV?) at maybe 10 Hz, even
if
> > it's 20 dB worse than at 1 KHz.
> >
> > Has anyone measured the Allan Deviation?
> >
> > Joe
> >
> >
> >
> Joe
>
> Noise plot is with respect to full scale (4Vrms).
OK, call it dbV+12db.
> I need to build a noise source to check the absolute level.
> Will use the amplified Johnson noise of a 150K resistor.
>
> By Allan deviation do you mean calculate it from the sequential 96KSPS
> ADC output samples?
Yes, although some decimation may be needed to keep compute times under
control, at least for the larger values of tau.
> I can do this, but since the dominant noise source is white the Allan
> deviation will scale with the measurement bandwidth.
Would modified Allan deviation be better?
I'm more interested in the general shape of the Allan curve than its
absolute value, one issue being the effect of thermal variations in your
laboratory dungeon. We had speculated as to the relative size of thermal
effects in these sound cards, and this would give us some idea.
Joe
BG
Bruce Griffiths
Mon, Dec 15, 2008 10:31 PM
Joe
Attached is noise spectrum (1kHz and below) of AP192 with nothing
connected to inputs.
Sampling rate 96KSPS.
Frequency bin equivalent noise bandwidth ~ 3Hz.
Noise has similar spectrum to flicker noise with a noise corner of
around 300Hz or so.
The noise floor ain't so bad, -130 dB (dBm? dbV?) at maybe 10 Hz, even
it's 20 dB worse than at 1 KHz.
Has anyone measured the Allan Deviation?
Joe
Joe
Noise plot is with respect to full scale (4Vrms).
I need to build a noise source to check the absolute level.
Will use the amplified Johnson noise of a 150K resistor.
By Allan deviation do you mean calculate it from the sequential 96KSPS
ADC output samples?
Yes, although some decimation may be needed to keep compute times under
control, at least for the larger values of tau.
I can do this, but since the dominant noise source is white the Allan
deviation will scale with the measurement bandwidth.
Would modified Allan deviation be better?
I'm more interested in the general shape of the Allan curve than its
absolute value, one issue being the effect of thermal variations in your
laboratory dungeon. We had speculated as to the relative size of thermal
effects in these sound cards, and this would give us some idea.
Joe
Joe
Modified ADEV, ADEV etc are possible, although the maximum usable record
length probably depends more on the limits of Plotter and Windows 2K.
I'll look into doing this.
Real time filtering and decimation may be impractical, in the short term
at least, as most signal processing libraries only process 16 bit samples.
Most real time spectrum analysis programs are similarly afflicted in
that they only process 16 bit samples.
Bruce
Joseph M Gwinn wrote:
> Bruce,
>
>
> time-nuts-bounces@febo.com wrote on 12/15/2008 04:34:34 PM:
>
>
>> Joseph M Gwinn wrote:
>>
>>> Bruce,
>>>
>>>
>>> time-nuts-bounces@febo.com wrote on 12/11/2008 08:36:47 PM:
>>>
>>>
>>>
>>>> Joe
>>>>
>>>> Attached is noise spectrum (1kHz and below) of AP192 with nothing
>>>> connected to inputs.
>>>> Sampling rate 96KSPS.
>>>> Frequency bin equivalent noise bandwidth ~ 3Hz.
>>>> Noise has similar spectrum to flicker noise with a noise corner of
>>>> around 300Hz or so.
>>>>
>>>>
>>> The noise floor ain't so bad, -130 dB (dBm? dbV?) at maybe 10 Hz, even
>>>
> if
>
>>> it's 20 dB worse than at 1 KHz.
>>>
>>> Has anyone measured the Allan Deviation?
>>>
>>> Joe
>>>
>>>
>>>
>>>
>> Joe
>>
>> Noise plot is with respect to full scale (4Vrms).
>>
>
> OK, call it dbV+12db.
>
>
>
>> I need to build a noise source to check the absolute level.
>> Will use the amplified Johnson noise of a 150K resistor.
>>
>> By Allan deviation do you mean calculate it from the sequential 96KSPS
>> ADC output samples?
>>
>
> Yes, although some decimation may be needed to keep compute times under
> control, at least for the larger values of tau.
>
>
>
>> I can do this, but since the dominant noise source is white the Allan
>> deviation will scale with the measurement bandwidth.
>>
>
> Would modified Allan deviation be better?
>
> I'm more interested in the general shape of the Allan curve than its
> absolute value, one issue being the effect of thermal variations in your
> laboratory dungeon. We had speculated as to the relative size of thermal
> effects in these sound cards, and this would give us some idea.
>
>
> Joe
>
>
Joe
Modified ADEV, ADEV etc are possible, although the maximum usable record
length probably depends more on the limits of Plotter and Windows 2K.
I'll look into doing this.
Real time filtering and decimation may be impractical, in the short term
at least, as most signal processing libraries only process 16 bit samples.
Most real time spectrum analysis programs are similarly afflicted in
that they only process 16 bit samples.
Bruce
JM
Joseph M Gwinn
Mon, Dec 15, 2008 11:03 PM
Joe
Joseph M Gwinn wrote:
Using a passive splitter for the LO drives will gain at least another
30dB in isolation between the 2 RF inputs if you use an appropriate
splitter.
True.
[Joe] I may have lost the thread here. If we have one oscillator
everything, one cannot have injection locking even if isolation isn't
perfect. What isolation does gain us is a reduction in undesired
You have 2 oscillators, the test source and the offset source, however
the >= 10Hz frequency offset between them means that the isolation
requirements are relaxed considerably.
If the offset oscillator is derived from the source then injection
locking doesnt occur.
I made the general comment to ensure that anyone following the thread,
who may be contemplating building a dual mixer setup with 2 sources very
close in frequency doesnt forget about the isolation requirements.
I see. We had two intertwined threads.
The only configuration for which it makes any sense is an inverting
input amplifier with a finite input voltage offset.
Why would non-inverting not work? Both inputs source or sink bias
currents, and non-inverting presents a very high impedance.
Non inverting amplifiers usually have lower noise and generally work
very well.
I was only trying to come up with a preamp circuit for which the
comments in the Minicircuits application note on the effect of
input offset voltage made any sense.
Ah. It may be hopeless.
My reading was that they were worried about bias currents from the amp
flowing into the mixer and causing offsets, not amplifier offset
per se. The amplifier offset voltage does not cause a mixer offset,
may be reduced by use of a chopper amp or very good balance.
By the way, I've noticed that Tek TDS3012B oscilloscope inputs can cause
offsets as well, again I assume from the bias currents. The circuit has
the scope input in parallel with the Agilent 34410A 6.5-digit voltmeter.
With scope input set to DC, big effect. Set to AC, small effect. Set to
Gnd, no effect. (Input is not grounded, so voltmeter is still happy.)
Didn't try changing the input volts/cm scale. Anyway, I think that this
effect is what the mystery app note was trying to say. A bias current
from the scope would cause a voltage offset that depended on the DC
resistance through which the bias current flowed, the DC load of the mixer
in this case.
If we design our own PCB then the AD7760 series ADCs are another
possible option.
These have a built in differential input differential
Yes. But aren't we trying to use commonly available soundcards?
Ideally yes, but they all seem to have built in performance limitations.
AFAIK the AP192 with its 4Vrms full scale balanced inputs with no
variable gain preamps or +48V phantom supplies seems to be one of the
best for this application.
Its major drawback is that its a PCI card located within a noisy PC.
I think that there are many top-end firewire soundcards. Whatever the
music folk like the sound of would be a good place to start - musicians'
well-trained hearing can be quite good. At least above 20 Hz.
Actually, the people that make the AP192 do have firewire and usb
offerings:
http://www.m-audio.com/index.php?do=products.family&ID=recording
The 4V rms input allows the mixer preamp to use devices like the THAT
1646 to drive the balanced sound card inputs without degrading the noise
floor too much.
Or build an isolation amp with some gain, and kill two birds with one
stone?
With a 1V rms full scale the noise floor degradation would be very
obvious when using a THAT 1646 (equivalent devices are even noisier).
It may be better to use a mixer preamp with a transformer coupled output
stage using hybrid feedback to achieve a low frequency cutoff below 1Hz
together with low noise.
With a transformer, even if toroidal, keeping hum out may prove quite
difficult.
Can alleviate [oddities at end of phase range} to some extent by
driving a pair of such phase detectors so that their outputs are
quadrature.
One just selects the phase detector output that is in the linear
range.
The quadrature outputs also allow unambiguous assignment of the
sign of any phase change.
The Symmetricom 5120A does something very clever to alleviate this
problem. Explained in US patent 7,227,346 and "Direct-Digital
Phase-Noise Measurement"; J. Grove, J. Hein, J. Retta, P.
W.Solbrig, and S.R. Stein; 2004 IEEE International Ultrasonics,
and Frequency Control Joint 50th Anniversary Conference, pages
The paper is also worthwhile, and available on the web somewhere
(don't recall where, but google found the pdf). I had to read the
multiple times to figure out what's going on. The correlation
approach is old as the hills, and only the digital phase detector
It may be feasible to achieve the same effect by purely digital means
least for low sample rates where FIR filters with tens of thousands
It is feasible, and Sam Stein is doing it. I've perhaps lost the
No, I meant replace his 90 degree hybrids with a digital equivalent.
I believe that his 90-degree hybrids are already digital. I think that
there is a very short analog path, maybe just a buffer amplifier and a
bunch of fancy ADCs. I had to reread the patent, to read between the
lines. It is presented as if there were a number of physical components,
but one can also read it to mean that these are logical components
implemented in some kind of machine code. At 400 MHz, one would assume
that the signal processor must be a FPGA, probably one of the fastest ones
made.
One example is the DDS core. Somewhere in the patent text it in effect
says that one can also implement this by table lookup, and that if one
chooses the table length correctly, one can contain a full cycle (where
the "phase accumulator" returns to the exact same value) of full-precision
amplitude samples. This allows one to eliminate all spectral spurs in the
DDS output, which is hard to do if there is phase truncation.
My understanding is that the 5120A is built upon a DSP or more likely
(of unspecified make and model). The 5125A will have a top frequency
400 MHz, so the DSP and/or FPGA better be damn fast. Little analog
Although the 5125A appeared in the 2008 product catalog, it isn't on the
website yet.
When I last talked to Symmetricom's sales folk about the 5120A and 5125A
(about a year ago), they said that Sam was still trying to get the 5125A
to work properly. He may still be cleaning things up, as scaling up from
30 MHz to 400 MHz is quite the jump, and may have forced a redesign.
Joe
Bruce,
time-nuts-bounces@febo.com [Bruce] wrote on 12/15/2008 04:56:26 PM:
> Joe
>
> Joseph M Gwinn wrote:
> > Bruce,
> >
> >
> > time-nuts-bounces@febo.com wrote on 12/11/2008 07:44:01 PM:
> >
> >
[snip]
> >
> >
> >> Using a passive splitter for the LO drives will gain at least another
> >> 30dB in isolation between the 2 RF inputs if you use an appropriate
> >> splitter.
> >>
> >
> > True.
> >
> > [Joe] I may have lost the thread here. If we have one oscillator
driving
> > everything, one cannot have injection locking even if isolation isn't
> > perfect. What isolation does gain us is a reduction in undesired
phase
> > shifts.
> >
> >
> >
> You have 2 oscillators, the test source and the offset source, however
> the >= 10Hz frequency offset between them means that the isolation
> requirements are relaxed considerably.
> If the offset oscillator is derived from the source then injection
> locking doesnt occur.
> I made the general comment to ensure that anyone following the thread,
> who may be contemplating building a dual mixer setup with 2 sources very
> close in frequency doesnt forget about the isolation requirements.
I see. We had two intertwined threads.
> >>> [snip]
> >>>
> >>>
> >>>> The only configuration for which it makes any sense is an inverting
> >>>> input amplifier with a finite input voltage offset.
> >>>>
> >>>>
> >>> Why would non-inverting not work? Both inputs source or sink bias
> >>> currents, and non-inverting presents a very high impedance.
> >>>
> >> Non inverting amplifiers usually have lower noise and generally work
> >> very well.
> >
> >> I was only trying to come up with a preamp circuit for which the
> >> comments in the Minicircuits application note on the effect of
amplifier
> >> input offset voltage made any sense.
> >>
> >
> > Ah. It may be hopeless.
> >
> > My reading was that they were worried about bias currents from the amp
> > flowing into the mixer and causing offsets, not amplifier offset
voltages
> > per se. The amplifier offset voltage does not cause a mixer offset,
and
> > may be reduced by use of a chopper amp or very good balance.
By the way, I've noticed that Tek TDS3012B oscilloscope inputs can cause
offsets as well, again I assume from the bias currents. The circuit has
the scope input in parallel with the Agilent 34410A 6.5-digit voltmeter.
With scope input set to DC, big effect. Set to AC, small effect. Set to
Gnd, no effect. (Input is not grounded, so voltmeter is still happy.)
Didn't try changing the input volts/cm scale. Anyway, I think that this
effect is what the mystery app note was trying to say. A bias current
from the scope would cause a voltage offset that depended on the DC
resistance through which the bias current flowed, the DC load of the mixer
in this case.
> >> If we design our own PCB then the AD7760 series ADCs are another
> >> possible option.
> >> These have a built in differential input differential
outputamplifier.
> >>
> >
> > Yes. But aren't we trying to use commonly available soundcards?
> >
> >
> Ideally yes, but they all seem to have built in performance limitations.
> AFAIK the AP192 with its 4Vrms full scale balanced inputs with no
> variable gain preamps or +48V phantom supplies seems to be one of the
> best for this application.
> Its major drawback is that its a PCI card located within a noisy PC.
I think that there are many top-end firewire soundcards. Whatever the
music folk like the sound of would be a good place to start - musicians'
well-trained hearing can be quite good. At least above 20 Hz.
Actually, the people that make the AP192 do have firewire and usb
offerings:
<http://www.m-audio.com/index.php?do=products.family&ID=recording>
> The 4V rms input allows the mixer preamp to use devices like the THAT
> 1646 to drive the balanced sound card inputs without degrading the noise
> floor too much.
Or build an isolation amp with some gain, and kill two birds with one
stone?
> With a 1V rms full scale the noise floor degradation would be very
> obvious when using a THAT 1646 (equivalent devices are even noisier).
> It may be better to use a mixer preamp with a transformer coupled output
> stage using hybrid feedback to achieve a low frequency cutoff below 1Hz
> together with low noise.
With a transformer, even if toroidal, keeping hum out may prove quite
difficult.
> >>>>>> Can alleviate [oddities at end of phase range} to some extent by
> >>>>>> driving a pair of such phase detectors so that their outputs are
in
> >>>>>> quadrature.
> >>>>>>
> >>>>>> One just selects the phase detector output that is in the linear
> >>>>>> range.
> >>>>>>
> >>>>>> The quadrature outputs also allow unambiguous assignment of the
> >>>>>> sign of any phase change.
> >>>>>>
> >>>>>>
> >>>>>>
> >>>>> The Symmetricom 5120A does something very clever to alleviate this
> >>>>> problem. Explained in US patent 7,227,346 and "Direct-Digital
> >>>>> Phase-Noise Measurement"; J. Grove, J. Hein, J. Retta, P.
Schweiger,
> >>>>> W.Solbrig, and S.R. Stein; 2004 IEEE International Ultrasonics,
Ferroelectrics,
> >>>>> and Frequency Control Joint 50th Anniversary Conference, pages
287-291.
> >>>>>
> >>>>>
> >>>> I've read the patent.
> >>>>
> >>>>
> >>> The paper is also worthwhile, and available on the web somewhere
> >>> (don't recall where, but google found the pdf). I had to read the
patent
> >>> multiple times to figure out what's going on. The correlation
> >>> approach is old as the hills, and only the digital phase detector
was patentable.
> >>>
> >>>
> >> It may be feasible to achieve the same effect by purely digital means
at
> >> least for low sample rates where FIR filters with tens of thousands
of
> >> taps are feasible.
> >>
> >
> > It *is* feasible, and Sam Stein is doing it. I've perhaps lost the
thread
> > here.
> >
> >
> No, I meant replace his 90 degree hybrids with a digital equivalent.
I believe that his 90-degree hybrids are already digital. I think that
there is a very short analog path, maybe just a buffer amplifier and a
bunch of fancy ADCs. I had to reread the patent, to read between the
lines. It is presented as if there were a number of physical components,
but one can also read it to mean that these are logical components
implemented in some kind of machine code. At 400 MHz, one would assume
that the signal processor must be a FPGA, probably one of the fastest ones
made.
One example is the DDS core. Somewhere in the patent text it in effect
says that one can also implement this by table lookup, and that if one
chooses the table length correctly, one can contain a full cycle (where
the "phase accumulator" returns to the exact same value) of full-precision
amplitude samples. This allows one to eliminate all spectral spurs in the
DDS output, which is hard to do if there is phase truncation.
> >
> > My understanding is that the 5120A is built upon a DSP or more likely
FPGA
> > (of unspecified make and model). The 5125A will have a top frequency
of
> > 400 MHz, so the DSP and/or FPGA better be damn fast. Little analog
stuff
> > remains.
> >
> >
> Although the 5125A appeared in the 2008 product catalog, it isn't on the
> website yet.
When I last talked to Symmetricom's sales folk about the 5120A and 5125A
(about a year ago), they said that Sam was still trying to get the 5125A
to work properly. He may still be cleaning things up, as scaling up from
30 MHz to 400 MHz is quite the jump, and may have forced a redesign.
Joe
BG
Bruce Griffiths
Mon, Dec 15, 2008 11:42 PM
Joe
Joseph M Gwinn wrote:
Joe
Joseph M Gwinn wrote:
Using a passive splitter for the LO drives will gain at least another
30dB in isolation between the 2 RF inputs if you use an appropriate
splitter.
True.
[Joe] I may have lost the thread here. If we have one oscillator
everything, one cannot have injection locking even if isolation isn't
perfect. What isolation does gain us is a reduction in undesired
You have 2 oscillators, the test source and the offset source, however
the >= 10Hz frequency offset between them means that the isolation
requirements are relaxed considerably.
If the offset oscillator is derived from the source then injection
locking doesnt occur.
I made the general comment to ensure that anyone following the thread,
who may be contemplating building a dual mixer setup with 2 sources very
close in frequency doesnt forget about the isolation requirements.
I see. We had two intertwined threads.
The only configuration for which it makes any sense is an inverting
input amplifier with a finite input voltage offset.
Why would non-inverting not work? Both inputs source or sink bias
currents, and non-inverting presents a very high impedance.
Non inverting amplifiers usually have lower noise and generally work
very well.
I was only trying to come up with a preamp circuit for which the
comments in the Minicircuits application note on the effect of
input offset voltage made any sense.
Ah. It may be hopeless.
My reading was that they were worried about bias currents from the amp
flowing into the mixer and causing offsets, not amplifier offset
per se. The amplifier offset voltage does not cause a mixer offset,
may be reduced by use of a chopper amp or very good balance.
By the way, I've noticed that Tek TDS3012B oscilloscope inputs can cause
offsets as well, again I assume from the bias currents. The circuit has
the scope input in parallel with the Agilent 34410A 6.5-digit voltmeter.
With scope input set to DC, big effect. Set to AC, small effect. Set to
Gnd, no effect. (Input is not grounded, so voltmeter is still happy.)
Didn't try changing the input volts/cm scale. Anyway, I think that this
effect is what the mystery app note was trying to say. A bias current
from the scope would cause a voltage offset that depended on the DC
resistance through which the bias current flowed, the DC load of the mixer
in this case.
However the proposed remedy has little or no effect on the errors caused
by such bias currents (eg transistor base currents).
The series resistor could be reduced to zero without effect on the mixer
offset due to the bias current. However the preamp offset due to the
source resistance would be reduced.
If we design our own PCB then the AD7760 series ADCs are another
possible option.
These have a built in differential input differential
Yes. But aren't we trying to use commonly available soundcards?
Ideally yes, but they all seem to have built in performance limitations.
AFAIK the AP192 with its 4Vrms full scale balanced inputs with no
variable gain preamps or +48V phantom supplies seems to be one of the
best for this application.
Its major drawback is that its a PCI card located within a noisy PC.
I think that there are many top-end firewire soundcards. Whatever the
music folk like the sound of would be a good place to start - musicians'
well-trained hearing can be quite good. At least above 20 Hz.
Actually, the people that make the AP192 do have firewire and usb
offerings:
http://www.m-audio.com/index.php?do=products.family&ID=recording
I've looked at all of the M-Audio offerings.
The more expensive ones have built in preamps plus 48V phantom supplies,
which can be switched off, however the presence of the switched +48V
supply is perhaps an invitation to disaster.
I've also looked at the specs for several other high end sound cards.
Quite a few only have single ended inputs.
Maybe, I should document the various cards and highlight their
shortcomings etc for this application.
The 4V rms input allows the mixer preamp to use devices like the THAT
1646 to drive the balanced sound card inputs without degrading the noise
floor too much.
Or build an isolation amp with some gain, and kill two birds with one
stone?
A low noise isolation amplifier with a frequency response down to 1Hz or
so without using a transformer may be difficult to do.
With a 1V rms full scale the noise floor degradation would be very
obvious when using a THAT 1646 (equivalent devices are even noisier).
It may be better to use a mixer preamp with a transformer coupled output
stage using hybrid feedback to achieve a low frequency cutoff below 1Hz
together with low noise.
With a transformer, even if toroidal, keeping hum out may prove quite
difficult.
High end (eg Lundahl LL1517) line output audio transformers come with mu
metal screens and metal foil interwinding shields.
Can alleviate [oddities at end of phase range} to some extent by
driving a pair of such phase detectors so that their outputs are
quadrature.
One just selects the phase detector output that is in the linear
range.
The quadrature outputs also allow unambiguous assignment of the
sign of any phase change.
The Symmetricom 5120A does something very clever to alleviate this
problem. Explained in US patent 7,227,346 and "Direct-Digital
Phase-Noise Measurement"; J. Grove, J. Hein, J. Retta, P.
W.Solbrig, and S.R. Stein; 2004 IEEE International Ultrasonics,
and Frequency Control Joint 50th Anniversary Conference, pages
The paper is also worthwhile, and available on the web somewhere
(don't recall where, but google found the pdf). I had to read the
multiple times to figure out what's going on. The correlation
approach is old as the hills, and only the digital phase detector
It may be feasible to achieve the same effect by purely digital means
least for low sample rates where FIR filters with tens of thousands
It is feasible, and Sam Stein is doing it. I've perhaps lost the
No, I meant replace his 90 degree hybrids with a digital equivalent.
I believe that his 90-degree hybrids are already digital.
I'm not convinced of that, if only because real time 10,000+ tap FIR
filters at 30+MSPS are probably still impractical.
I think that
there is a very short analog path, maybe just a buffer amplifier and a
bunch of fancy ADCs. I had to reread the patent, to read between the
lines. It is presented as if there were a number of physical components,
but one can also read it to mean that these are logical components
implemented in some kind of machine code. At 400 MHz, one would assume
that the signal processor must be a FPGA, probably one of the fastest ones
made.
One example is the DDS core. Somewhere in the patent text it in effect
says that one can also implement this by table lookup, and that if one
chooses the table length correctly, one can contain a full cycle (where
the "phase accumulator" returns to the exact same value) of full-precision
amplitude samples. This allows one to eliminate all spectral spurs in the
DDS output, which is hard to do if there is phase truncation.
My understanding is that the 5120A is built upon a DSP or more likely
(of unspecified make and model). The 5125A will have a top frequency
400 MHz, so the DSP and/or FPGA better be damn fast. Little analog
Although the 5125A appeared in the 2008 product catalog, it isn't on the
website yet.
When I last talked to Symmetricom's sales folk about the 5120A and 5125A
(about a year ago), they said that Sam was still trying to get the 5125A
to work properly. He may still be cleaning things up, as scaling up from
30 MHz to 400 MHz is quite the jump, and may have forced a redesign.
Joe
Joe
Joseph M Gwinn wrote:
> Bruce,
>
>
> time-nuts-bounces@febo.com [Bruce] wrote on 12/15/2008 04:56:26 PM:
>
>
>> Joe
>>
>> Joseph M Gwinn wrote:
>>
>>> Bruce,
>>>
>>>
>>> time-nuts-bounces@febo.com wrote on 12/11/2008 07:44:01 PM:
>>>
>>>
>>>
> [snip]
>
>>>
>>>> Using a passive splitter for the LO drives will gain at least another
>>>> 30dB in isolation between the 2 RF inputs if you use an appropriate
>>>> splitter.
>>>>
>>>>
>>> True.
>>>
>>> [Joe] I may have lost the thread here. If we have one oscillator
>>>
> driving
>
>>> everything, one cannot have injection locking even if isolation isn't
>>> perfect. What isolation does gain us is a reduction in undesired
>>>
> phase
>
>>> shifts.
>>>
>>>
>>>
>>>
>> You have 2 oscillators, the test source and the offset source, however
>> the >= 10Hz frequency offset between them means that the isolation
>> requirements are relaxed considerably.
>> If the offset oscillator is derived from the source then injection
>> locking doesnt occur.
>> I made the general comment to ensure that anyone following the thread,
>> who may be contemplating building a dual mixer setup with 2 sources very
>> close in frequency doesnt forget about the isolation requirements.
>>
>
> I see. We had two intertwined threads.
>
>
>
>>>>> [snip]
>>>>>
>>>>>
>>>>>
>>>>>> The only configuration for which it makes any sense is an inverting
>>>>>> input amplifier with a finite input voltage offset.
>>>>>>
>>>>>>
>>>>>>
>>>>> Why would non-inverting not work? Both inputs source or sink bias
>>>>> currents, and non-inverting presents a very high impedance.
>>>>>
>>>>>
>>>> Non inverting amplifiers usually have lower noise and generally work
>>>> very well.
>>>>
>>>> I was only trying to come up with a preamp circuit for which the
>>>> comments in the Minicircuits application note on the effect of
>>>>
> amplifier
>
>>>> input offset voltage made any sense.
>>>>
>>>>
>>> Ah. It may be hopeless.
>>>
>>> My reading was that they were worried about bias currents from the amp
>>>
>
>
>>> flowing into the mixer and causing offsets, not amplifier offset
>>>
> voltages
>
>>> per se. The amplifier offset voltage does not cause a mixer offset,
>>>
> and
>
>>> may be reduced by use of a chopper amp or very good balance.
>>>
>
> By the way, I've noticed that Tek TDS3012B oscilloscope inputs can cause
> offsets as well, again I assume from the bias currents. The circuit has
> the scope input in parallel with the Agilent 34410A 6.5-digit voltmeter.
> With scope input set to DC, big effect. Set to AC, small effect. Set to
> Gnd, no effect. (Input is not grounded, so voltmeter is still happy.)
> Didn't try changing the input volts/cm scale. Anyway, I think that this
> effect is what the mystery app note was trying to say. A bias current
> from the scope would cause a voltage offset that depended on the DC
> resistance through which the bias current flowed, the DC load of the mixer
> in this case.
>
>
However the proposed remedy has little or no effect on the errors caused
by such bias currents (eg transistor base currents).
The series resistor could be reduced to zero without effect on the mixer
offset due to the bias current. However the preamp offset due to the
source resistance would be reduced.
>
>
>
>>>> If we design our own PCB then the AD7760 series ADCs are another
>>>> possible option.
>>>> These have a built in differential input differential
>>>>
> outputamplifier.
>
>>> Yes. But aren't we trying to use commonly available soundcards?
>>>
>>>
>>>
>> Ideally yes, but they all seem to have built in performance limitations.
>> AFAIK the AP192 with its 4Vrms full scale balanced inputs with no
>> variable gain preamps or +48V phantom supplies seems to be one of the
>> best for this application.
>> Its major drawback is that its a PCI card located within a noisy PC.
>>
>
> I think that there are many top-end firewire soundcards. Whatever the
> music folk like the sound of would be a good place to start - musicians'
> well-trained hearing can be quite good. At least above 20 Hz.
>
> Actually, the people that make the AP192 do have firewire and usb
> offerings:
>
> <http://www.m-audio.com/index.php?do=products.family&ID=recording>
>
>
I've looked at all of the M-Audio offerings.
The more expensive ones have built in preamps plus 48V phantom supplies,
which can be switched off, however the presence of the switched +48V
supply is perhaps an invitation to disaster.
I've also looked at the specs for several other high end sound cards.
Quite a few only have single ended inputs.
Maybe, I should document the various cards and highlight their
shortcomings etc for this application.
>> The 4V rms input allows the mixer preamp to use devices like the THAT
>> 1646 to drive the balanced sound card inputs without degrading the noise
>> floor too much.
>>
>
> Or build an isolation amp with some gain, and kill two birds with one
> stone?
>
>
>
A low noise isolation amplifier with a frequency response down to 1Hz or
so without using a transformer may be difficult to do.
>> With a 1V rms full scale the noise floor degradation would be very
>> obvious when using a THAT 1646 (equivalent devices are even noisier).
>> It may be better to use a mixer preamp with a transformer coupled output
>> stage using hybrid feedback to achieve a low frequency cutoff below 1Hz
>> together with low noise.
>>
>
> With a transformer, even if toroidal, keeping hum out may prove quite
> difficult.
>
>
High end (eg Lundahl LL1517) line output audio transformers come with mu
metal screens and metal foil interwinding shields.
>
>
>>>>>>>> Can alleviate [oddities at end of phase range} to some extent by
>>>>>>>> driving a pair of such phase detectors so that their outputs are
>>>>>>>>
> in
>
>>>>>>>> quadrature.
>>>>>>>>
>>>>>>>> One just selects the phase detector output that is in the linear
>>>>>>>> range.
>>>>>>>>
>>>>>>>> The quadrature outputs also allow unambiguous assignment of the
>>>>>>>> sign of any phase change.
>>>>>>>>
>>>>>>>>
>>>>>>>>
>>>>>>>>
>>>>>>> The Symmetricom 5120A does something very clever to alleviate this
>>>>>>>
>
>
>>>>>>> problem. Explained in US patent 7,227,346 and "Direct-Digital
>>>>>>> Phase-Noise Measurement"; J. Grove, J. Hein, J. Retta, P.
>>>>>>>
> Schweiger,
>
>>>>>>> W.Solbrig, and S.R. Stein; 2004 IEEE International Ultrasonics,
>>>>>>>
> Ferroelectrics,
>
>>>>>>> and Frequency Control Joint 50th Anniversary Conference, pages
>>>>>>>
> 287-291.
>
>>>>>>>
>>>>>> I've read the patent.
>>>>>>
>>>>>>
>>>>>>
>>>>> The paper is also worthwhile, and available on the web somewhere
>>>>> (don't recall where, but google found the pdf). I had to read the
>>>>>
> patent
>
>>>>> multiple times to figure out what's going on. The correlation
>>>>> approach is old as the hills, and only the digital phase detector
>>>>>
> was patentable.
>
>>>>>
>>>> It may be feasible to achieve the same effect by purely digital means
>>>>
> at
>
>>>> least for low sample rates where FIR filters with tens of thousands
>>>>
> of
>
>>>> taps are feasible.
>>>>
>>>>
>>> It *is* feasible, and Sam Stein is doing it. I've perhaps lost the
>>>
> thread
>
>>> here.
>>>
>>>
>>>
>> No, I meant replace his 90 degree hybrids with a digital equivalent.
>>
>
> I believe that his 90-degree hybrids are already digital.
I'm not convinced of that, if only because real time 10,000+ tap FIR
filters at 30+MSPS are probably still impractical.
> I think that
> there is a very short analog path, maybe just a buffer amplifier and a
> bunch of fancy ADCs. I had to reread the patent, to read between the
> lines. It is presented as if there were a number of physical components,
> but one can also read it to mean that these are logical components
> implemented in some kind of machine code. At 400 MHz, one would assume
> that the signal processor must be a FPGA, probably one of the fastest ones
> made.
>
> One example is the DDS core. Somewhere in the patent text it in effect
> says that one can also implement this by table lookup, and that if one
> chooses the table length correctly, one can contain a full cycle (where
> the "phase accumulator" returns to the exact same value) of full-precision
> amplitude samples. This allows one to eliminate all spectral spurs in the
> DDS output, which is hard to do if there is phase truncation.
>
>
>
>>> My understanding is that the 5120A is built upon a DSP or more likely
>>>
> FPGA
>
>>> (of unspecified make and model). The 5125A will have a top frequency
>>>
> of
>
>>> 400 MHz, so the DSP and/or FPGA better be damn fast. Little analog
>>>
> stuff
>
>>> remains.
>>>
>>>
>>>
>> Although the 5125A appeared in the 2008 product catalog, it isn't on the
>> website yet.
>>
>
> When I last talked to Symmetricom's sales folk about the 5120A and 5125A
> (about a year ago), they said that Sam was still trying to get the 5125A
> to work properly. He may still be cleaning things up, as scaling up from
> 30 MHz to 400 MHz is quite the jump, and may have forced a redesign.
>
> Joe
>
>
>
Bruce
W
wa1zms@att.net
Tue, Dec 16, 2008 2:07 AM
Bruce-
OK... So, linear operation does therefore seem to be the preferred
way to operate these MMICs rather than operation into compression.
That's what I seem to be observing if only because my final RF
frequency is so high and RX bandwidth so low.
Having said that, if my frequency synthesis scheme involves a mixer
does the same effect of low frequency noise to phase noise conversion still
take place? After all, the mixer element is typically into compression
if it's a FET based mixer. I assume a diode mixer is more immune to
similar effects?
I'm trying to grow my intuitive understanding of the subtle sources of
noise. But I don't recall Maas giving much info on this topic in
his otherwise excellent text.
As always, thanks for your sagely advice.
-Brian
-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com]On
Behalf Of Bruce Griffiths
Sent: Monday, December 15, 2008 5:00 PM
To: Discussion of precise time and frequency measurement
Subject: Re: [time-nuts] Close-in phase noise question...
wa1zms@att.net wrote:
Looking for comment here...
The background:
I'm working on a sub mm-wave LO chain for
a ham radio application. While chasing issues
of close-in phase (ie: within 1KHz of RF
carrier) by peeling the "layers of the onion",
I'm starting to question the performance of
the MMICs that are used as buffers and amps
following my Wenzel reference OCXOs.
Question(s):
Should any MMIC be allowed to be driven
close to compression or into compression
when striving for best close-in noise?
I know and have seen the NF of a MMIC
degrade while in compression, but my
target right now is close-in noise rather
than broadband noise.
My design, in summary, takes 5MHz up to 630GHz
via several multipliers and PLL stages.
-Brian
Bruce-
OK... So, linear operation does therefore seem to be the preferred
way to operate these MMICs rather than operation into compression.
That's what I seem to be observing if only because my final RF
frequency is so high and RX bandwidth so low.
Having said that, if my frequency synthesis scheme involves a mixer
does the same effect of low frequency noise to phase noise conversion still
take place? After all, the mixer element is typically into compression
if it's a FET based mixer. I assume a diode mixer is more immune to
similar effects?
I'm trying to grow my intuitive understanding of the subtle sources of
noise. But I don't recall Maas giving much info on this topic in
his otherwise excellent text.
As always, thanks for your sagely advice.
-Brian
-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com]On
Behalf Of Bruce Griffiths
Sent: Monday, December 15, 2008 5:00 PM
To: Discussion of precise time and frequency measurement
Subject: Re: [time-nuts] Close-in phase noise question...
wa1zms@att.net wrote:
> Looking for comment here...
>
> The background:
> I'm working on a sub mm-wave LO chain for
> a ham radio application. While chasing issues
> of close-in phase (ie: within 1KHz of RF
> carrier) by peeling the "layers of the onion",
> I'm starting to question the performance of
> the MMICs that are used as buffers and amps
> following my Wenzel reference OCXOs.
>
> Question(s):
> Should any MMIC be allowed to be driven
> close to compression or into compression
> when striving for best close-in noise?
>
> I know and have seen the NF of a MMIC
> degrade while in compression, but my
> target right now is close-in noise rather
> than broadband noise.
>
> My design, in summary, takes 5MHz up to 630GHz
> via several multipliers and PLL stages.
>
> -Brian
>
Brian
The increased nonlinearity when driven into compression will enhance the
conversion of low frequency noise to phase noise.
Bruce
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BG
Bruce Griffiths
Tue, Dec 16, 2008 2:18 AM
Brian
Low frequency noise modulates the switching times of the mixer
components and hence produces close in phase noise.
Diode mixers especially those using Schottky diodes have lower flicker
noise than active mixers.
Passive FET mixers are not immune to flicker noise.
The level of such flicker noise increases with the RF current level.
Bruce
wa1zms@att.net wrote:
Bruce-
OK... So, linear operation does therefore seem to be the preferred
way to operate these MMICs rather than operation into compression.
That's what I seem to be observing if only because my final RF
frequency is so high and RX bandwidth so low.
Having said that, if my frequency synthesis scheme involves a mixer
does the same effect of low frequency noise to phase noise conversion still
take place? After all, the mixer element is typically into compression
if it's a FET based mixer. I assume a diode mixer is more immune to
similar effects?
I'm trying to grow my intuitive understanding of the subtle sources of
noise. But I don't recall Maas giving much info on this topic in
his otherwise excellent text.
As always, thanks for your sagely advice.
-Brian
-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com]On
Behalf Of Bruce Griffiths
Sent: Monday, December 15, 2008 5:00 PM
To: Discussion of precise time and frequency measurement
Subject: Re: [time-nuts] Close-in phase noise question...
wa1zms@att.net wrote:
Looking for comment here...
The background:
I'm working on a sub mm-wave LO chain for
a ham radio application. While chasing issues
of close-in phase (ie: within 1KHz of RF
carrier) by peeling the "layers of the onion",
I'm starting to question the performance of
the MMICs that are used as buffers and amps
following my Wenzel reference OCXOs.
Question(s):
Should any MMIC be allowed to be driven
close to compression or into compression
when striving for best close-in noise?
I know and have seen the NF of a MMIC
degrade while in compression, but my
target right now is close-in noise rather
than broadband noise.
My design, in summary, takes 5MHz up to 630GHz
via several multipliers and PLL stages.
-Brian
Brian
The increased nonlinearity when driven into compression will enhance the
conversion of low frequency noise to phase noise.
Bruce
Brian
Low frequency noise modulates the switching times of the mixer
components and hence produces close in phase noise.
Diode mixers especially those using Schottky diodes have lower flicker
noise than active mixers.
Passive FET mixers are not immune to flicker noise.
The level of such flicker noise increases with the RF current level.
Bruce
wa1zms@att.net wrote:
> Bruce-
>
> OK... So, linear operation does therefore seem to be the preferred
> way to operate these MMICs rather than operation into compression.
> That's what I seem to be observing if only because my final RF
> frequency is so high and RX bandwidth so low.
>
> Having said that, if my frequency synthesis scheme involves a mixer
> does the same effect of low frequency noise to phase noise conversion still
> take place? After all, the mixer element is typically into compression
> if it's a FET based mixer. I assume a diode mixer is more immune to
> similar effects?
>
> I'm trying to grow my intuitive understanding of the subtle sources of
> noise. But I don't recall Maas giving much info on this topic in
> his otherwise excellent text.
>
> As always, thanks for your sagely advice.
>
> -Brian
>
>
> -----Original Message-----
> From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com]On
> Behalf Of Bruce Griffiths
> Sent: Monday, December 15, 2008 5:00 PM
> To: Discussion of precise time and frequency measurement
> Subject: Re: [time-nuts] Close-in phase noise question...
>
>
> wa1zms@att.net wrote:
>
>> Looking for comment here...
>>
>> The background:
>> I'm working on a sub mm-wave LO chain for
>> a ham radio application. While chasing issues
>> of close-in phase (ie: within 1KHz of RF
>> carrier) by peeling the "layers of the onion",
>> I'm starting to question the performance of
>> the MMICs that are used as buffers and amps
>> following my Wenzel reference OCXOs.
>>
>> Question(s):
>> Should any MMIC be allowed to be driven
>> close to compression or into compression
>> when striving for best close-in noise?
>>
>> I know and have seen the NF of a MMIC
>> degrade while in compression, but my
>> target right now is close-in noise rather
>> than broadband noise.
>>
>> My design, in summary, takes 5MHz up to 630GHz
>> via several multipliers and PLL stages.
>>
>> -Brian
>>
>>
> Brian
>
> The increased nonlinearity when driven into compression will enhance the
> conversion of low frequency noise to phase noise.
>
> Bruce
>
>
LC
Luis Cupido
Tue, Dec 16, 2008 2:44 AM
Brian,
I also think the linear operation would be better, but not
so sure if bipolar transistors wouldn't be preferred
over MMIC's for this (on bottom part of the spectrum)
(ok... is not so handy).
What I can certainly add to the discussion is that
power amplification followed by higher rate varactor
multiplication is considerably better than a multiple
lower multiplication ratios chain.
I had that experience on the 411GHz where from a 70MHz(*)
xtal osc amplified to several Watt driving varactors and
cavities. I jump with few steps to about 45GHz as opposed
to a DB6NT like LO chain which was noticeably worst in
the close in noise.
Using mixers without driving hard the LO or the RF with
same marginal level on both ports is possible but you will
be in trouble with level settings :-(
Not sure how much you would gain there... theoretically
something... but then AM to PM conversion is against you so
not sure if better or worst.
Luis Cupido
ct1dmk.
(*) x12 x9 x3 x2
w/ 9th harm corner cube harm mixer
if my memory serves me well....
wa1zms@att.net wrote:
Bruce-
OK... So, linear operation does therefore seem to be the preferred
way to operate these MMICs rather than operation into compression.
That's what I seem to be observing if only because my final RF
frequency is so high and RX bandwidth so low.
Having said that, if my frequency synthesis scheme involves a mixer
does the same effect of low frequency noise to phase noise conversion still
take place? After all, the mixer element is typically into compression
if it's a FET based mixer. I assume a diode mixer is more immune to
similar effects?
I'm trying to grow my intuitive understanding of the subtle sources of
noise. But I don't recall Maas giving much info on this topic in
his otherwise excellent text.
As always, thanks for your sagely advice.
-Brian
-----Original Message-----
From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com]On
Behalf Of Bruce Griffiths
Sent: Monday, December 15, 2008 5:00 PM
To: Discussion of precise time and frequency measurement
Subject: Re: [time-nuts] Close-in phase noise question...
wa1zms@att.net wrote:
Looking for comment here...
The background:
I'm working on a sub mm-wave LO chain for
a ham radio application. While chasing issues
of close-in phase (ie: within 1KHz of RF
carrier) by peeling the "layers of the onion",
I'm starting to question the performance of
the MMICs that are used as buffers and amps
following my Wenzel reference OCXOs.
Question(s):
Should any MMIC be allowed to be driven
close to compression or into compression
when striving for best close-in noise?
I know and have seen the NF of a MMIC
degrade while in compression, but my
target right now is close-in noise rather
than broadband noise.
My design, in summary, takes 5MHz up to 630GHz
via several multipliers and PLL stages.
-Brian
Brian,
I also think the linear operation would be better, but not
so sure if bipolar transistors wouldn't be preferred
over MMIC's for this (on bottom part of the spectrum)
(ok... is not so handy).
What I can certainly add to the discussion is that
power amplification followed by higher rate varactor
multiplication is considerably better than a multiple
lower multiplication ratios chain.
I had that experience on the 411GHz where from a 70MHz(*)
xtal osc amplified to several Watt driving varactors and
cavities. I jump with few steps to about 45GHz as opposed
to a DB6NT like LO chain which was noticeably worst in
the close in noise.
Using mixers without driving hard the LO or the RF with
same marginal level on both ports is possible but you will
be in trouble with level settings :-(
Not sure how much you would gain there... theoretically
something... but then AM to PM conversion is against you so
not sure if better or worst.
Luis Cupido
ct1dmk.
(*) x12 x9 x3 x2
w/ 9th harm corner cube harm mixer
if my memory serves me well....
wa1zms@att.net wrote:
> Bruce-
>
> OK... So, linear operation does therefore seem to be the preferred
> way to operate these MMICs rather than operation into compression.
> That's what I seem to be observing if only because my final RF
> frequency is so high and RX bandwidth so low.
>
> Having said that, if my frequency synthesis scheme involves a mixer
> does the same effect of low frequency noise to phase noise conversion still
> take place? After all, the mixer element is typically into compression
> if it's a FET based mixer. I assume a diode mixer is more immune to
> similar effects?
>
> I'm trying to grow my intuitive understanding of the subtle sources of
> noise. But I don't recall Maas giving much info on this topic in
> his otherwise excellent text.
>
> As always, thanks for your sagely advice.
>
> -Brian
>
>
> -----Original Message-----
> From: time-nuts-bounces@febo.com [mailto:time-nuts-bounces@febo.com]On
> Behalf Of Bruce Griffiths
> Sent: Monday, December 15, 2008 5:00 PM
> To: Discussion of precise time and frequency measurement
> Subject: Re: [time-nuts] Close-in phase noise question...
>
>
> wa1zms@att.net wrote:
>> Looking for comment here...
>>
>> The background:
>> I'm working on a sub mm-wave LO chain for
>> a ham radio application. While chasing issues
>> of close-in phase (ie: within 1KHz of RF
>> carrier) by peeling the "layers of the onion",
>> I'm starting to question the performance of
>> the MMICs that are used as buffers and amps
>> following my Wenzel reference OCXOs.
>>
>> Question(s):
>> Should any MMIC be allowed to be driven
>> close to compression or into compression
>> when striving for best close-in noise?
>>
>> I know and have seen the NF of a MMIC
>> degrade while in compression, but my
>> target right now is close-in noise rather
>> than broadband noise.
>>
>> My design, in summary, takes 5MHz up to 630GHz
>> via several multipliers and PLL stages.
>>
>> -Brian
>>
> Brian
>
> The increased nonlinearity when driven into compression will enhance the
> conversion of low frequency noise to phase noise.
>
> Bruce
>
> _______________________________________________
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> To unsubscribe, go to
> https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
> and follow the instructions there.
>
>
> _______________________________________________
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> and follow the instructions there.
>
JM
Joseph M Gwinn
Tue, Dec 16, 2008 11:54 PM
Joe
Joseph M Gwinn wrote:
The only configuration for which it makes any sense is an
input amplifier with a finite input voltage offset.
Why would non-inverting not work? Both inputs source or sink bias
currents, and non-inverting presents a very high impedance.
Non inverting amplifiers usually have lower noise and generally
very well.
I was only trying to come up with a preamp circuit for which the
comments in the Minicircuits application note on the effect of
amplifier input offset voltage made any sense.
Ah. It may be hopeless.
My reading was that they were worried about bias currents from the
flowing into the mixer and causing offsets, not amplifier offset
per se. The amplifier offset voltage does not cause a mixer offset,
and may be reduced by use of a chopper amp or very good balance.
By the way, I've noticed that Tek TDS3012B oscilloscope inputs can
offsets as well, again I assume from the bias currents. The circuit
the scope input in parallel with the Agilent 34410A 6.5-digit
With scope input set to DC, big effect. Set to AC, small effect. Set
Gnd, no effect. (Input is not grounded, so voltmeter is still happy.)
Didn't try changing the input volts/cm scale. Anyway, I think that
effect is what the mystery app note was trying to say. A bias current
from the scope would cause a voltage offset that depended on the DC
resistance through which the bias current flowed, the DC load of the
However the proposed remedy has little or no effect on the errors caused
by such bias currents (eg transistor base currents).
The series resistor could be reduced to zero without effect on the mixer
offset due to the bias current. However the preamp offset due to the
source resistance would be reduced.
Hmm. It may be simpler than that. With the TDS3012B and 34410A connected
in parallel across the IF output of a mixer, the bias currents from the
TDS3012B developed a voltage across the mixer load resistor, and this
voltage was sensed by the 34410A. All the phase detector had to do was
not short the bias current to ground.
If we design our own PCB then the AD7760 series ADCs are another
possible option.
These have a built in differential input differential output
Yes. But aren't we trying to use commonly available soundcards?
Ideally yes, but they all seem to have built in performance
AFAIK the AP192 with its 4Vrms full scale balanced inputs with no
variable gain preamps or +48V phantom supplies seems to be one of the
best for this application.
Its major drawback is that its a PCI card located within a noisy PC.
I think that there are many top-end firewire soundcards. Whatever the
music folk like the sound of would be a good place to start -
I've looked at all of the M-Audio offerings.
The more expensive ones have built in preamps plus 48V phantom supplies,
which can be switched off, however the presence of the switched +48V
supply is perhaps an invitation to disaster.
Given that capacitance to ground is more benefit than problem in this
application, I would be tempted to use a pair of back-to-back rectifier
diodes as a clamp to protect the mixer IF output et al. The 48 volt
phantom supply will be short-circuit protected, so current will
automatically limit.
I've also looked at the specs for several other high end sound cards.
Quite a few only have single ended inputs.
Maybe, I should document the various cards and highlight their
shortcomings etc for this application.
That would be very useful.
The 4V rms input allows the mixer preamp to use devices like the THAT
1646 to drive the balanced sound card inputs without degrading the
Or build an isolation amp with some gain, and kill two birds with one
stone?
A low noise isolation amplifier with a frequency response down to 1Hz or
so without using a transformer may be difficult to do.
People do make low noise common-base RF amplifiers, but 1 Hz would yield
some pretty large bypass capacitors, even if the flicker noise can be
controlled well enough. I would consider using ultracapacitors, which
didn't exist until very recently, and of course have very large
capacitance values.
With a 1V rms full scale the noise floor degradation would be very
obvious when using a THAT 1646 (equivalent devices are even noisier).
It may be better to use a mixer preamp with a transformer coupled
stage using hybrid feedback to achieve a low frequency cutoff below
With a transformer, even if toroidal, keeping hum out may prove quite
difficult.
High end (eg Lundahl LL1517) line output audio transformers come with mu
metal screens and metal foil interwinding shields.
They don't pass 1 Hz very well. I bet the rolloff is ~20 Hz.
Certainly one can build a VLF transformer, but it will be a project for
sure, and the transformer may be quite large.
The [5120A] paper is also worthwhile, and available on the web
(don't recall where, but google found the pdf). I had to read the
multiple times to figure out what's going on. The correlation
approach is old as the hills, and only the digital phase detector
It may be feasible to achieve the same effect by purely digital
at least for low sample rates where FIR filters with tens of
It is feasible, and Sam Stein is doing it. I've perhaps lost the
No, I meant replace his 90 degree hybrids with a digital equivalent.
I believe that his 90-degree hybrids are already digital.
I'm not convinced of that, if only because real time 10,000+ tap FIR
filters at 30+MSPS are probably still impractical.
I'm not convinced that one needs a 10,000-tap FIR to achieve this, and Sam
Stein is one smart fellow. I recall some NASA patents from twenty years
ago on how to get I+Q data from a single ADC, and while there was FIR
processing of some kind, there were only maybe 8 or 16 taps. And Tayloe
(US patent 6,230,000) gets much the same effect with one resistor, four
capacitors, an analog mux, and two differential amplifiers.
How accurately must the quadrature delay be achieved? If I recall, the
patent or paper implies that it need not be exact.
I also recall thinking that he could implement the quadrature delay
digitally. I don't recall the details, but it depended on cutting time up
into one-second batches and processing each batch independently of the
others. I suppose one can use a FFT-Multiply-IFFT to do it directly.
Joe
Bruce,
time-nuts-bounces@febo.com wrote on 12/15/2008 06:42:59 PM:
> Joe
> Joseph M Gwinn wrote:
> > Bruce,
> >
> >
> > time-nuts-bounces@febo.com [Bruce] wrote on 12/15/2008 04:56:26 PM:
> >
> >>>>> [snip]
> >>>>>
> >>>>>
> >>>>>
> >>>>>> The only configuration for which it makes any sense is an
inverting
> >>>>>> input amplifier with a finite input voltage offset.
> >>>>>>
> >>>>>>
> >>>>>>
> >>>>> Why would non-inverting not work? Both inputs source or sink bias
> >>>>> currents, and non-inverting presents a very high impedance.
> >>>>>
> >>>>>
> >>>> Non inverting amplifiers usually have lower noise and generally
work
> >>>> very well.
> >>>>
> >>>> I was only trying to come up with a preamp circuit for which the
> >>>> comments in the Minicircuits application note on the effect of
> >>>> amplifier input offset voltage made any sense.
> >>>>
> >>>>
> >>> Ah. It may be hopeless.
> >>>
> >>> My reading was that they were worried about bias currents from the
amp
> >>> flowing into the mixer and causing offsets, not amplifier offset
voltages
> >>> per se. The amplifier offset voltage does not cause a mixer offset,
> >>> and may be reduced by use of a chopper amp or very good balance.
> >>>
> >
> > By the way, I've noticed that Tek TDS3012B oscilloscope inputs can
cause
> > offsets as well, again I assume from the bias currents. The circuit
has
> > the scope input in parallel with the Agilent 34410A 6.5-digit
voltmeter.
> > With scope input set to DC, big effect. Set to AC, small effect. Set
to
> > Gnd, no effect. (Input is not grounded, so voltmeter is still happy.)
> > Didn't try changing the input volts/cm scale. Anyway, I think that
this
> > effect is what the mystery app note was trying to say. A bias current
> > from the scope would cause a voltage offset that depended on the DC
> > resistance through which the bias current flowed, the DC load of the
mixer
> > in this case.
> >
> >
> However the proposed remedy has little or no effect on the errors caused
> by such bias currents (eg transistor base currents).
> The series resistor could be reduced to zero without effect on the mixer
> offset due to the bias current. However the preamp offset due to the
> source resistance would be reduced.
Hmm. It may be simpler than that. With the TDS3012B and 34410A connected
in parallel across the IF output of a mixer, the bias currents from the
TDS3012B developed a voltage across the mixer load resistor, and this
voltage was sensed by the 34410A. All the phase detector had to do was
not short the bias current to ground.
> >>>> If we design our own PCB then the AD7760 series ADCs are another
> >>>> possible option.
> >>>> These have a built in differential input differential output
amplifier.
> >
> >>> Yes. But aren't we trying to use commonly available soundcards?
> >>>
> >>>
> >>>
> >> Ideally yes, but they all seem to have built in performance
limitations.
> >> AFAIK the AP192 with its 4Vrms full scale balanced inputs with no
> >> variable gain preamps or +48V phantom supplies seems to be one of the
> >> best for this application.
> >> Its major drawback is that its a PCI card located within a noisy PC.
> >>
> >
> > I think that there are many top-end firewire soundcards. Whatever the
> > music folk like the sound of would be a good place to start -
musicians'
> > well-trained hearing can be quite good. At least above 20 Hz.
> >
> > Actually, the people that make the AP192 do have firewire and usb
> > offerings:
> >
> > <http://www.m-audio.com/index.php?do=products.family&ID=recording>
> >
> >
>
> I've looked at all of the M-Audio offerings.
> The more expensive ones have built in preamps plus 48V phantom supplies,
> which can be switched off, however the presence of the switched +48V
> supply is perhaps an invitation to disaster.
Given that capacitance to ground is more benefit than problem in this
application, I would be tempted to use a pair of back-to-back rectifier
diodes as a clamp to protect the mixer IF output et al. The 48 volt
phantom supply will be short-circuit protected, so current will
automatically limit.
> I've also looked at the specs for several other high end sound cards.
> Quite a few only have single ended inputs.
> Maybe, I should document the various cards and highlight their
> shortcomings etc for this application.
That would be very useful.
> >> The 4V rms input allows the mixer preamp to use devices like the THAT
> >> 1646 to drive the balanced sound card inputs without degrading the
noise
> >> floor too much.
> >>
> >
> > Or build an isolation amp with some gain, and kill two birds with one
> > stone?
> >
> >
> >
> A low noise isolation amplifier with a frequency response down to 1Hz or
> so without using a transformer may be difficult to do.
People do make low noise common-base RF amplifiers, but 1 Hz would yield
some pretty large bypass capacitors, even if the flicker noise can be
controlled well enough. I would consider using ultracapacitors, which
didn't exist until very recently, and of course have very large
capacitance values.
> >> With a 1V rms full scale the noise floor degradation would be very
> >> obvious when using a THAT 1646 (equivalent devices are even noisier).
> >> It may be better to use a mixer preamp with a transformer coupled
output
> >> stage using hybrid feedback to achieve a low frequency cutoff below
1Hz
> >> together with low noise.
> >>
> >
> > With a transformer, even if toroidal, keeping hum out may prove quite
> > difficult.
> >
> >
> High end (eg Lundahl LL1517) line output audio transformers come with mu
> metal screens and metal foil interwinding shields.
They don't pass 1 Hz very well. I bet the rolloff is ~20 Hz.
Certainly one can build a VLF transformer, but it will be a project for
sure, and the transformer may be quite large.
> >>>>>>
> >>>>> The [5120A] paper is also worthwhile, and available on the web
somewhere
> >>>>> (don't recall where, but google found the pdf). I had to read the
patent
> >>>>> multiple times to figure out what's going on. The correlation
> >>>>> approach is old as the hills, and only the digital phase detector
was patentable.
> >>>>>
> >>>> It may be feasible to achieve the same effect by purely digital
means
> >>>> at least for low sample rates where FIR filters with tens of
thousands
> >>>> of taps are feasible.
> >>>>
> >>>>
> >>> It *is* feasible, and Sam Stein is doing it. I've perhaps lost the
thread
> >>> here.
> >>>
> >>>
> >>>
> >> No, I meant replace his 90 degree hybrids with a digital equivalent.
> >>
> >
> > I believe that his 90-degree hybrids are already digital.
> I'm not convinced of that, if only because real time 10,000+ tap FIR
> filters at 30+MSPS are probably still impractical.
I'm not convinced that one needs a 10,000-tap FIR to achieve this, and Sam
Stein is one smart fellow. I recall some NASA patents from twenty years
ago on how to get I+Q data from a single ADC, and while there was FIR
processing of some kind, there were only maybe 8 or 16 taps. And Tayloe
(US patent 6,230,000) gets much the same effect with one resistor, four
capacitors, an analog mux, and two differential amplifiers.
How accurately must the quadrature delay be achieved? If I recall, the
patent or paper implies that it need not be exact.
I also recall thinking that he could implement the quadrature delay
digitally. I don't recall the details, but it depended on cutting time up
into one-second batches and processing each batch independently of the
others. I suppose one can use a FFT-Multiply-IFFT to do it directly.
Joe
JM
Joseph M Gwinn
Wed, Dec 17, 2008 12:07 AM
I need to build a noise source to check the absolute level.
Will use the amplified Johnson noise of a 150K resistor.
By Allan deviation do you mean calculate it from the sequential
Yes, although some decimation may be needed to keep compute times
control, at least for the larger values of tau.
I can do this, but since the dominant noise source is white the Allan
deviation will scale with the measurement bandwidth.
Would modified Allan deviation be better?
I'm more interested in the general shape of the Allan curve than its
absolute value, one issue being the effect of thermal variations in
laboratory dungeon. We had speculated as to the relative size of
effects in these sound cards, and this would give us some idea.
Joe
Joe
Modified ADEV, ADEV etc are possible, although the maximum usable record
length probably depends more on the limits of Plotter and Windows 2K.
I'll look into doing this.
Real time filtering and decimation may be impractical, in the short term
at least, as most signal processing libraries only process 16 bit
Most real time spectrum analysis programs are similarly afflicted in
that they only process 16 bit samples.
I don't see why we would need realtime filtering. Data reduction can be
offline, so we ought to be able to use 32-bit or 64-bit arithmetic.
Given that we will inspect Allan Deviation data in a log-log plot, one can
save much processing time by spacing the tau values to be computed
uniformly in log tau. I've played with this in Mathematica, and it does
work and yields a large speedup factor. It should also help with Plotter
and Win2K limits. One trick is to ensure that one computes each tau value
at most once. This check is needed because with close spacing, the round
function will yield the same tau values multiple times for small values of
tau.
Joe
Bruce,
time-nuts-bounces@febo.com wrote on 12/15/2008 05:31:27 PM:
> Joseph M Gwinn wrote:
> > Bruce,
> >
> >
> > time-nuts-bounces@febo.com wrote on 12/15/2008 04:34:34 PM:
> >
[snip]
> >
> >> I need to build a noise source to check the absolute level.
> >> Will use the amplified Johnson noise of a 150K resistor.
> >>
> >> By Allan deviation do you mean calculate it from the sequential
96KSPS
> >> ADC output samples?
> >>
> >
> > Yes, although some decimation may be needed to keep compute times
under
> > control, at least for the larger values of tau.
> >
> >
> >
> >> I can do this, but since the dominant noise source is white the Allan
> >> deviation will scale with the measurement bandwidth.
> >>
> >
> > Would modified Allan deviation be better?
> >
> > I'm more interested in the general shape of the Allan curve than its
> > absolute value, one issue being the effect of thermal variations in
your
> > laboratory dungeon. We had speculated as to the relative size of
thermal
> > effects in these sound cards, and this would give us some idea.
> >
> >
> > Joe
> >
> >
>
> Joe
>
> Modified ADEV, ADEV etc are possible, although the maximum usable record
> length probably depends more on the limits of Plotter and Windows 2K.
>
>
> I'll look into doing this.
> Real time filtering and decimation may be impractical, in the short term
> at least, as most signal processing libraries only process 16 bit
samples.
> Most real time spectrum analysis programs are similarly afflicted in
> that they only process 16 bit samples.
I don't see why we would need realtime filtering. Data reduction can be
offline, so we ought to be able to use 32-bit or 64-bit arithmetic.
Given that we will inspect Allan Deviation data in a log-log plot, one can
save much processing time by spacing the tau values to be computed
uniformly in log tau. I've played with this in Mathematica, and it does
work and yields a large speedup factor. It should also help with Plotter
and Win2K limits. One trick is to ensure that one computes each tau value
at most once. This check is needed because with close spacing, the round
function will yield the same tau values multiple times for small values of
tau.
Joe
BG
Bruce Griffiths
Wed, Dec 17, 2008 1:43 AM
Joe
Joseph M Gwinn wrote:
Joe
Joseph M Gwinn wrote:
The only configuration for which it makes any sense is an
input amplifier with a finite input voltage offset.
Why would non-inverting not work? Both inputs source or sink bias
currents, and non-inverting presents a very high impedance.
Non inverting amplifiers usually have lower noise and generally
very well.
I was only trying to come up with a preamp circuit for which the
comments in the Minicircuits application note on the effect of
amplifier input offset voltage made any sense.
Ah. It may be hopeless.
My reading was that they were worried about bias currents from the
flowing into the mixer and causing offsets, not amplifier offset
per se. The amplifier offset voltage does not cause a mixer offset,
and may be reduced by use of a chopper amp or very good balance.
By the way, I've noticed that Tek TDS3012B oscilloscope inputs can
offsets as well, again I assume from the bias currents. The circuit
the scope input in parallel with the Agilent 34410A 6.5-digit
With scope input set to DC, big effect. Set to AC, small effect. Set
Gnd, no effect. (Input is not grounded, so voltmeter is still happy.)
Didn't try changing the input volts/cm scale. Anyway, I think that
effect is what the mystery app note was trying to say. A bias current
from the scope would cause a voltage offset that depended on the DC
resistance through which the bias current flowed, the DC load of the
However the proposed remedy has little or no effect on the errors caused
by such bias currents (eg transistor base currents).
The series resistor could be reduced to zero without effect on the mixer
offset due to the bias current. However the preamp offset due to the
source resistance would be reduced.
Hmm. It may be simpler than that. With the TDS3012B and 34410A connected
in parallel across the IF output of a mixer, the bias currents from the
TDS3012B developed a voltage across the mixer load resistor, and this
voltage was sensed by the 34410A. All the phase detector had to do was
not short the bias current to ground.
If we design our own PCB then the AD7760 series ADCs are another
possible option.
These have a built in differential input differential output
Yes. But aren't we trying to use commonly available soundcards?
Ideally yes, but they all seem to have built in performance
AFAIK the AP192 with its 4Vrms full scale balanced inputs with no
variable gain preamps or +48V phantom supplies seems to be one of the
best for this application.
Its major drawback is that its a PCI card located within a noisy PC.
I think that there are many top-end firewire soundcards. Whatever the
music folk like the sound of would be a good place to start -
I've looked at all of the M-Audio offerings.
The more expensive ones have built in preamps plus 48V phantom supplies,
which can be switched off, however the presence of the switched +48V
supply is perhaps an invitation to disaster.
Given that capacitance to ground is more benefit than problem in this
application, I would be tempted to use a pair of back-to-back rectifier
diodes as a clamp to protect the mixer IF output et al. The 48 volt
phantom supply will be short-circuit protected, so current will
automatically limit.
A pair of coupling capacitors at the preamp output combined with clamp
diodes to the amplifier power supply rails would work well even if the
+48V cant be switched off.
The +48V appears between the balanced pair conductors and ground.
Unfortunately the power available from the phantom supply may not be
sufficient to power the mixer preamp.
I've also looked at the specs for several other high end sound cards.
Quite a few only have single ended inputs.
Maybe, I should document the various cards and highlight their
shortcomings etc for this application.
That would be very useful.
I'll start on this shortly.
The 4V rms input allows the mixer preamp to use devices like the THAT
1646 to drive the balanced sound card inputs without degrading the
Or build an isolation amp with some gain, and kill two birds with one
stone?
A low noise isolation amplifier with a frequency response down to 1Hz or
so without using a transformer may be difficult to do.
People do make low noise common-base RF amplifiers, but 1 Hz would yield
some pretty large bypass capacitors, even if the flicker noise can be
controlled well enough. I would consider using ultracapacitors, which
didn't exist until very recently, and of course have very large
capacitance values.
A CB stage probably isnt optimum for the mixer preamp so that lower
value caps can be used provided that they effectively short the
amplifier input resistor Johnson noise at the frequencies of interest.
With a 1V rms full scale the noise floor degradation would be very
obvious when using a THAT 1646 (equivalent devices are even noisier).
It may be better to use a mixer preamp with a transformer coupled
stage using hybrid feedback to achieve a low frequency cutoff below
With a transformer, even if toroidal, keeping hum out may prove quite
difficult.
High end (eg Lundahl LL1517) line output audio transformers come with mu
metal screens and metal foil interwinding shields.
They don't pass 1 Hz very well. I bet the rolloff is ~20 Hz.
When driven conventionally the transformer cutoff is around 20Hz,
however if one uses the appropriate driver with a controlled negative
output R to cancel the transformer primary internal winding resistance,
the low frequency response can be extended significantly. This also
reduces the low frequency distortion.
However individual adjustment of driver to suit transformer is required
and tracking the winding resistance over temperature may be an issue.
Certainly one can build a VLF transformer, but it will be a project for
sure, and the transformer may be quite large.
The transformers only weigh about 65g.
It may be simpler just to select a mixer for which the IF ground can be
isolated from the RF and IF grounds.
However a preamp with a transformer output may be useful if one uses a
mixer where all the grounds are connected together by the package.
The [5120A] paper is also worthwhile, and available on the web
(don't recall where, but google found the pdf). I had to read the
multiple times to figure out what's going on. The correlation
approach is old as the hills, and only the digital phase detector
It may be feasible to achieve the same effect by purely digital
at least for low sample rates where FIR filters with tens of
It is feasible, and Sam Stein is doing it. I've perhaps lost the
No, I meant replace his 90 degree hybrids with a digital equivalent.
I believe that his 90-degree hybrids are already digital.
I'm not convinced of that, if only because real time 10,000+ tap FIR
filters at 30+MSPS are probably still impractical.
I'm not convinced that one needs a 10,000-tap FIR to achieve this, and Sam
Stein is one smart fellow. I recall some NASA patents from twenty years
ago on how to get I+Q data from a single ADC, and while there was FIR
processing of some kind, there were only maybe 8 or 16 taps. And Tayloe
(US patent 6,230,000) gets much the same effect with one resistor, four
capacitors, an analog mux, and two differential amplifiers.
I have read similar papers from that era on radar signal processing.
They either used a Hilbert transform or a pair of digital filters whose
outputs were in phase quadrature.
The quadrature accuracy for a given bandwidth depends on the the number
of taps.
The beat frequencies (in a dual mixer system) won't match exactly and
some correction for the resultant phase shift errors will need to be made.
This may be less of a problem when the 2 beat frequency signals are
identical in frequency and just differ in phase.
How accurately must the quadrature delay be achieved? If I recall, the
patent or paper implies that it need not be exact.
I also recall thinking that he could implement the quadrature delay
digitally. I don't recall the details, but it depended on cutting time up
into one-second batches and processing each batch independently of the
others. I suppose one can use a FFT-Multiply-IFFT to do it directly.
Joe
Joe
Joseph M Gwinn wrote:
> Bruce,
>
>
> time-nuts-bounces@febo.com wrote on 12/15/2008 06:42:59 PM:
>
>
>> Joe
>> Joseph M Gwinn wrote:
>>
>>> Bruce,
>>>
>>>
>>> time-nuts-bounces@febo.com [Bruce] wrote on 12/15/2008 04:56:26 PM:
>>>
>>>
>
>
>>>>>>> [snip]
>>>>>>>
>>>>>>>
>>>>>>>
>>>>>>>
>>>>>>>> The only configuration for which it makes any sense is an
>>>>>>>>
> inverting
>
>>>>>>>> input amplifier with a finite input voltage offset.
>>>>>>>>
>>>>>>>>
>>>>>>>>
>>>>>>>>
>>>>>>> Why would non-inverting not work? Both inputs source or sink bias
>>>>>>>
>
>
>>>>>>> currents, and non-inverting presents a very high impedance.
>>>>>>>
>>>>>>>
>>>>>>>
>>>>>> Non inverting amplifiers usually have lower noise and generally
>>>>>>
> work
>
>>>>>> very well.
>>>>>>
>>>>>> I was only trying to come up with a preamp circuit for which the
>>>>>> comments in the Minicircuits application note on the effect of
>>>>>> amplifier input offset voltage made any sense.
>>>>>>
>>>>>>
>>>>>>
>>>>> Ah. It may be hopeless.
>>>>>
>>>>> My reading was that they were worried about bias currents from the
>>>>>
> amp
>
>>>>> flowing into the mixer and causing offsets, not amplifier offset
>>>>>
> voltages
>
>>>>> per se. The amplifier offset voltage does not cause a mixer offset,
>>>>>
>
>
>>>>> and may be reduced by use of a chopper amp or very good balance.
>>>>>
>>>>>
>>> By the way, I've noticed that Tek TDS3012B oscilloscope inputs can
>>>
> cause
>
>>> offsets as well, again I assume from the bias currents. The circuit
>>>
> has
>
>>> the scope input in parallel with the Agilent 34410A 6.5-digit
>>>
> voltmeter.
>
>>> With scope input set to DC, big effect. Set to AC, small effect. Set
>>>
> to
>
>>> Gnd, no effect. (Input is not grounded, so voltmeter is still happy.)
>>>
>
>
>>> Didn't try changing the input volts/cm scale. Anyway, I think that
>>>
> this
>
>>> effect is what the mystery app note was trying to say. A bias current
>>>
>
>
>>> from the scope would cause a voltage offset that depended on the DC
>>> resistance through which the bias current flowed, the DC load of the
>>>
> mixer
>
>>> in this case.
>>>
>>>
>>>
>> However the proposed remedy has little or no effect on the errors caused
>> by such bias currents (eg transistor base currents).
>> The series resistor could be reduced to zero without effect on the mixer
>> offset due to the bias current. However the preamp offset due to the
>> source resistance would be reduced.
>>
>
> Hmm. It may be simpler than that. With the TDS3012B and 34410A connected
> in parallel across the IF output of a mixer, the bias currents from the
> TDS3012B developed a voltage across the mixer load resistor, and this
> voltage was sensed by the 34410A. All the phase detector had to do was
> not short the bias current to ground.
>
>
>
AC coupling?
>
>>>>>> If we design our own PCB then the AD7760 series ADCs are another
>>>>>> possible option.
>>>>>> These have a built in differential input differential output
>>>>>>
> amplifier.
>
>>>>> Yes. But aren't we trying to use commonly available soundcards?
>>>>>
>>>>>
>>>>>
>>>>>
>>>> Ideally yes, but they all seem to have built in performance
>>>>
> limitations.
>
>>>> AFAIK the AP192 with its 4Vrms full scale balanced inputs with no
>>>> variable gain preamps or +48V phantom supplies seems to be one of the
>>>> best for this application.
>>>> Its major drawback is that its a PCI card located within a noisy PC.
>>>>
>>>>
>>> I think that there are many top-end firewire soundcards. Whatever the
>>>
>
>
>>> music folk like the sound of would be a good place to start -
>>>
> musicians'
>
>>> well-trained hearing can be quite good. At least above 20 Hz.
>>>
>>> Actually, the people that make the AP192 do have firewire and usb
>>> offerings:
>>>
>>> <http://www.m-audio.com/index.php?do=products.family&ID=recording>
>>>
>>>
>>>
>> I've looked at all of the M-Audio offerings.
>> The more expensive ones have built in preamps plus 48V phantom supplies,
>> which can be switched off, however the presence of the switched +48V
>> supply is perhaps an invitation to disaster.
>>
>
> Given that capacitance to ground is more benefit than problem in this
> application, I would be tempted to use a pair of back-to-back rectifier
> diodes as a clamp to protect the mixer IF output et al. The 48 volt
> phantom supply will be short-circuit protected, so current will
> automatically limit.
>
>
A pair of coupling capacitors at the preamp output combined with clamp
diodes to the amplifier power supply rails would work well even if the
+48V cant be switched off.
The +48V appears between the balanced pair conductors and ground.
Unfortunately the power available from the phantom supply may not be
sufficient to power the mixer preamp.
>
>> I've also looked at the specs for several other high end sound cards.
>> Quite a few only have single ended inputs.
>> Maybe, I should document the various cards and highlight their
>> shortcomings etc for this application.
>>
>
> That would be very useful.
>
I'll start on this shortly.
>
>
>>>> The 4V rms input allows the mixer preamp to use devices like the THAT
>>>> 1646 to drive the balanced sound card inputs without degrading the
>>>>
> noise
>
>>>> floor too much.
>>>>
>>>>
>>> Or build an isolation amp with some gain, and kill two birds with one
>>> stone?
>>>
>>>
>>>
>>>
>> A low noise isolation amplifier with a frequency response down to 1Hz or
>> so without using a transformer may be difficult to do.
>>
>
> People do make low noise common-base RF amplifiers, but 1 Hz would yield
> some pretty large bypass capacitors, even if the flicker noise can be
> controlled well enough. I would consider using ultracapacitors, which
> didn't exist until very recently, and of course have very large
> capacitance values.
>
>
A CB stage probably isnt optimum for the mixer preamp so that lower
value caps can be used provided that they effectively short the
amplifier input resistor Johnson noise at the frequencies of interest.
>
>>>> With a 1V rms full scale the noise floor degradation would be very
>>>> obvious when using a THAT 1646 (equivalent devices are even noisier).
>>>> It may be better to use a mixer preamp with a transformer coupled
>>>>
> output
>
>>>> stage using hybrid feedback to achieve a low frequency cutoff below
>>>>
> 1Hz
>
>>>> together with low noise.
>>>>
>>>>
>>> With a transformer, even if toroidal, keeping hum out may prove quite
>>> difficult.
>>>
>>>
>>>
>> High end (eg Lundahl LL1517) line output audio transformers come with mu
>> metal screens and metal foil interwinding shields.
>>
>
> They don't pass 1 Hz very well. I bet the rolloff is ~20 Hz.
>
>
When driven conventionally the transformer cutoff is around 20Hz,
however if one uses the appropriate driver with a controlled negative
output R to cancel the transformer primary internal winding resistance,
the low frequency response can be extended significantly. This also
reduces the low frequency distortion.
However individual adjustment of driver to suit transformer is required
and tracking the winding resistance over temperature may be an issue.
> Certainly one can build a VLF transformer, but it will be a project for
> sure, and the transformer may be quite large.
>
>
The transformers only weigh about 65g.
It may be simpler just to select a mixer for which the IF ground can be
isolated from the RF and IF grounds.
However a preamp with a transformer output may be useful if one uses a
mixer where all the grounds are connected together by the package.
>
>
>>>>>>> The [5120A] paper is also worthwhile, and available on the web
>>>>>>>
> somewhere
>
>>>>>>> (don't recall where, but google found the pdf). I had to read the
>>>>>>>
> patent
>
>>>>>>> multiple times to figure out what's going on. The correlation
>>>>>>> approach is old as the hills, and only the digital phase detector
>>>>>>>
> was patentable.
>
>>>>>> It may be feasible to achieve the same effect by purely digital
>>>>>>
> means
>
>>>>>> at least for low sample rates where FIR filters with tens of
>>>>>>
> thousands
>
>>>>>> of taps are feasible.
>>>>>>
>>>>>>
>>>>>>
>>>>> It *is* feasible, and Sam Stein is doing it. I've perhaps lost the
>>>>>
> thread
>
>>>>> here.
>>>>>
>>>>>
>>>>>
>>>>>
>>>> No, I meant replace his 90 degree hybrids with a digital equivalent.
>>>>
>>>>
>>> I believe that his 90-degree hybrids are already digital.
>>>
>> I'm not convinced of that, if only because real time 10,000+ tap FIR
>> filters at 30+MSPS are probably still impractical.
>>
>
> I'm not convinced that one needs a 10,000-tap FIR to achieve this, and Sam
> Stein is one smart fellow. I recall some NASA patents from twenty years
> ago on how to get I+Q data from a single ADC, and while there was FIR
> processing of some kind, there were only maybe 8 or 16 taps. And Tayloe
> (US patent 6,230,000) gets much the same effect with one resistor, four
> capacitors, an analog mux, and two differential amplifiers.
>
>
I have read similar papers from that era on radar signal processing.
They either used a Hilbert transform or a pair of digital filters whose
outputs were in phase quadrature.
The quadrature accuracy for a given bandwidth depends on the the number
of taps.
The beat frequencies (in a dual mixer system) won't match exactly and
some correction for the resultant phase shift errors will need to be made.
This may be less of a problem when the 2 beat frequency signals are
identical in frequency and just differ in phase.
> How accurately must the quadrature delay be achieved? If I recall, the
> patent or paper implies that it need not be exact.
>
> I also recall thinking that he could implement the quadrature delay
> digitally. I don't recall the details, but it depended on cutting time up
> into one-second batches and processing each batch independently of the
> others. I suppose one can use a FFT-Multiply-IFFT to do it directly.
>
>
> Joe
>
>
>
Bruce
BG
Bruce Griffiths
Wed, Dec 17, 2008 2:05 AM
I need to build a noise source to check the absolute level.
Will use the amplified Johnson noise of a 150K resistor.
By Allan deviation do you mean calculate it from the sequential
Yes, although some decimation may be needed to keep compute times
control, at least for the larger values of tau.
I can do this, but since the dominant noise source is white the Allan
deviation will scale with the measurement bandwidth.
Would modified Allan deviation be better?
I'm more interested in the general shape of the Allan curve than its
absolute value, one issue being the effect of thermal variations in
laboratory dungeon. We had speculated as to the relative size of
effects in these sound cards, and this would give us some idea.
Joe
Joe
Modified ADEV, ADEV etc are possible, although the maximum usable record
length probably depends more on the limits of Plotter and Windows 2K.
I'll look into doing this.
Real time filtering and decimation may be impractical, in the short term
at least, as most signal processing libraries only process 16 bit
Most real time spectrum analysis programs are similarly afflicted in
that they only process 16 bit samples.
I don't see why we would need realtime filtering. Data reduction can be
offline, so we ought to be able to use 32-bit or 64-bit arithmetic.
Given that we will inspect Allan Deviation data in a log-log plot, one can
save much processing time by spacing the tau values to be computed
uniformly in log tau. I've played with this in Mathematica, and it does
work and yields a large speedup factor. It should also help with Plotter
and Win2K limits. One trick is to ensure that one computes each tau value
at most once. This check is needed because with close spacing, the round
function will yield the same tau values multiple times for small values of
tau.
Joe
Joe
Real time processing certainly isn't required to characterise the
performance.
However some may be tempted to do this, its probably possible with a
sufficently fast machine.
I was just highlighting a problem with some available signal processing
libraries which may have been developed before sound cards with
resolutions of more than 16 bits became available.
Some (perhaps most) real time spectrum display software also has this
problem (eg baudline, Virtins etc).
It isnt necessary to use a pair of mixers and an offset source to
characterise the sound card, driving both sound card inputs from the
same audio source should suffice.
The audio source need not have low ultra low distortion (the IF output
signals in a dual mixer system wont have ultra low distortion) or very
high frequency stability ( the IF output signals in a dual mixer system
wont necessarily have particularly high frequency stability).
A standard RC audio oscillator with distortion lower than 1% or so
should suffice.
At least the resultant frequency fluctuations should thoroughly exercise
the phase extraction algorithms.
Another option would be to low pass filter the output of a divider.
Using a sound card to generate the test signal is also possible but it
can potentially introduce extraneous noise and other artifacts such as
phase truncation spurs.
Bruce
Joseph M Gwinn wrote:
> Bruce,
>
>
> time-nuts-bounces@febo.com wrote on 12/15/2008 05:31:27 PM:
>
>
>> Joseph M Gwinn wrote:
>>
>>> Bruce,
>>>
>>>
>>> time-nuts-bounces@febo.com wrote on 12/15/2008 04:34:34 PM:
>>>
>>>
> [snip]
>
>>>> I need to build a noise source to check the absolute level.
>>>> Will use the amplified Johnson noise of a 150K resistor.
>>>>
>>>> By Allan deviation do you mean calculate it from the sequential
>>>>
> 96KSPS
>
>>>> ADC output samples?
>>>>
>>>>
>>> Yes, although some decimation may be needed to keep compute times
>>>
> under
>
>>> control, at least for the larger values of tau.
>>>
>>>
>>>
>>>
>>>> I can do this, but since the dominant noise source is white the Allan
>>>> deviation will scale with the measurement bandwidth.
>>>>
>>>>
>>> Would modified Allan deviation be better?
>>>
>>> I'm more interested in the general shape of the Allan curve than its
>>> absolute value, one issue being the effect of thermal variations in
>>>
> your
>
>>> laboratory dungeon. We had speculated as to the relative size of
>>>
> thermal
>
>>> effects in these sound cards, and this would give us some idea.
>>>
>>>
>>> Joe
>>>
>>>
>>>
>> Joe
>>
>> Modified ADEV, ADEV etc are possible, although the maximum usable record
>> length probably depends more on the limits of Plotter and Windows 2K.
>>
>>
>> I'll look into doing this.
>> Real time filtering and decimation may be impractical, in the short term
>> at least, as most signal processing libraries only process 16 bit
>>
> samples.
>
>> Most real time spectrum analysis programs are similarly afflicted in
>> that they only process 16 bit samples.
>>
>
> I don't see why we would need realtime filtering. Data reduction can be
> offline, so we ought to be able to use 32-bit or 64-bit arithmetic.
>
> Given that we will inspect Allan Deviation data in a log-log plot, one can
> save much processing time by spacing the tau values to be computed
> uniformly in log tau. I've played with this in Mathematica, and it does
> work and yields a large speedup factor. It should also help with Plotter
> and Win2K limits. One trick is to ensure that one computes each tau value
> at most once. This check is needed because with close spacing, the round
> function will yield the same tau values multiple times for small values of
> tau.
>
> Joe
>
>
Joe
Real time processing certainly isn't required to characterise the
performance.
However some may be tempted to do this, its probably possible with a
sufficently fast machine.
I was just highlighting a problem with some available signal processing
libraries which may have been developed before sound cards with
resolutions of more than 16 bits became available.
Some (perhaps most) real time spectrum display software also has this
problem (eg baudline, Virtins etc).
It isnt necessary to use a pair of mixers and an offset source to
characterise the sound card, driving both sound card inputs from the
same audio source should suffice.
The audio source need not have low ultra low distortion (the IF output
signals in a dual mixer system wont have ultra low distortion) or very
high frequency stability ( the IF output signals in a dual mixer system
wont necessarily have particularly high frequency stability).
A standard RC audio oscillator with distortion lower than 1% or so
should suffice.
At least the resultant frequency fluctuations should thoroughly exercise
the phase extraction algorithms.
Another option would be to low pass filter the output of a divider.
Using a sound card to generate the test signal is also possible but it
can potentially introduce extraneous noise and other artifacts such as
phase truncation spurs.
Bruce
JM
Joseph M Gwinn
Wed, Dec 17, 2008 2:19 AM
Joe
Joseph M Gwinn wrote:
However the proposed remedy has little or no effect on the errors
by such bias currents (eg transistor base currents).
The series resistor could be reduced to zero without effect on the
offset due to the bias current. However the preamp offset due to the
source resistance would be reduced.
Hmm. It may be simpler than that. With the TDS3012B and 34410A
in parallel across the IF output of a mixer, the bias currents from
TDS3012B developed a voltage across the mixer load resistor, and this
voltage was sensed by the 34410A. All the phase detector hadto do was
not short the bias current to ground.
At the expense of phase shifts and temperature sensitivity, but yes.
And it makes it hard to sense a DC signal, if that's the intent.
I think that there are many top-end firewire soundcards. Whatever
music folk like the sound of would be a good place to start -
I've looked at all of the M-Audio offerings.
The more expensive ones have built in preamps plus 48V phantom
which can be switched off, however the presence of the switched +48V
supply is perhaps an invitation to disaster.
Given that capacitance to ground is more benefit than problem in this
application, I would be tempted to use a pair of back-to-back
diodes as a clamp to protect the mixer IF output et al. The 48 volt
phantom supply will be short-circuit protected, so current will
automatically limit.
A pair of coupling capacitors at the preamp output combined with clamp
diodes to the amplifier power supply rails would work well even if the
+48V can't be switched off.
The +48V appears between the balanced pair conductors and ground.
Unfortunately the power available from the phantom supply may not be
sufficient to power the mixer preamp.
OK. The power limit does make protection easy. I gather that the limit
is a few milliamps, so even a 1N4148 would work.
I've also looked at the specs for several other high end sound cards.
Quite a few only have single ended inputs.
Maybe, I should document the various cards and highlight their
shortcomings etc for this application.
That would be very useful.
I'll start on this shortly.
The 4V rms input allows the mixer preamp to use devices like the
1646 to drive the balanced sound card inputs without degrading the
noise floor too much.
Or build an isolation amp with some gain, and kill two birds with
A low noise isolation amplifier with a frequency response down to 1Hz
so without using a transformer may be difficult to do.
People do make low noise common-base RF amplifiers, but 1 Hz would
some pretty large bypass capacitors, even if the flicker noise can be
controlled well enough. I would consider using ultracapacitors, which
didn't exist until very recently, and of course have very large
capacitance values.
A CB stage probably isn't optimum for the mixer preamp so that lower
value caps can be used provided that they effectively short the
amplifier input resistor Johnson noise at the frequencies of interest.
But is a CB stage adequate? Elimination of hum pickup is worth a lot.
With a 1V rms full scale the noise floor degradation would be very
obvious when using a THAT 1646 (equivalent devices are even
It may be better to use a mixer preamp with a transformer coupled
stage using hybrid feedback to achieve a low frequency cutoff below
1Hz together with low noise.
With a transformer, even if toroidal, keeping hum out may prove
High end (eg Lundahl LL1517) line output audio transformers come with
metal screens and metal foil interwinding shields.
They don't pass 1 Hz very well. I bet the rolloff is ~20 Hz.
When driven conventionally the transformer cutoff is around 20Hz,
however if one uses the appropriate driver with a controlled negative
output R to cancel the transformer primary internal winding resistance,
the low frequency response can be extended significantly. This also
reduces the low frequency distortion.
However individual adjustment of driver to suit transformer is required
and tracking the winding resistance over temperature may be an issue.
That would certainly work. See next.
Certainly one can build a VLF transformer, but it will be a project
sure, and the transformer may be quite large.
The transformers only weigh about 65g.
This weight estimate assumes the negative resistance circuit I assume, the
intent being to allow use of the Lundahl LL1517 transformer.
You might be interested in the following article: "A Broadband Active
Antenna for ELF Magnetic Fields" by John F. Sutton and G. Craig Spaniol"
in Physics Essays March 1993, Vol 6, #1, 1993. The negative-impedance
trick is expanded upon. Sutton also has some US patents on this. US
patent 5,296,866 covers the antenna, but is hard to understand without the
article.
It may be simpler just to select a mixer for which the IF ground can be
isolated from the RF and IF grounds.
However a preamp with a transformer output may be useful if one uses a
mixer where all the grounds are connected together by the package.
It has to be far easier to select the right mixer than to deal with a 1 Hz
transformer. And cheaper.
No, I meant replace his 90 degree hybrids with a digital
I believe that his 90-degree hybrids are already digital.
I'm not convinced of that, if only because real time 10,000+ tap FIR
filters at 30+MSPS are probably still impractical.
I'm not convinced that one needs a 10,000-tap FIR to achieve this, and
Stein is one smart fellow. I recall some NASA patents from twenty
ago on how to get I+Q data from a single ADC, and while there was FIR
processing of some kind, there were only maybe 8 or 16 taps. And
(US patent 6,230,000) gets much the same effect with one resistor,
capacitors, an analog mux, and two differential amplifiers.
I have read similar papers from that era on radar signal processing.
They either used a Hilbert transform or a pair of digital filters whose
outputs were in phase quadrature.
The quadrature accuracy for a given bandwidth depends on the the number
of taps.
The beat frequencies (in a dual mixer system) won't match exactly and
some correction for the resultant phase shift errors will need
to be made.
This may be less of a problem when the 2 beat frequency signals are
identical in frequency and just differ in phase.
So long as we know the exact frequency, even if it isn't the exact desired
frequency, all may be well.
Joe
Bruce,
time-nuts-bounces@febo.com wrote on 12/16/2008 08:43:29 PM:
> Joe
>
> Joseph M Gwinn wrote:
> > Bruce,
> >
> >
> > time-nuts-bounces@febo.com wrote on 12/15/2008 06:42:59 PM:
> >
[snip]
> >>>
> >>>
> >> However the proposed remedy has little or no effect on the errors
caused
> >> by such bias currents (eg transistor base currents).
> >> The series resistor could be reduced to zero without effect on the
mixer
> >> offset due to the bias current. However the preamp offset due to the
> >> source resistance would be reduced.
> >>
> >
> > Hmm. It may be simpler than that. With the TDS3012B and 34410A
connected
> > in parallel across the IF output of a mixer, the bias currents from
the
> > TDS3012B developed a voltage across the mixer load resistor, and this
> > voltage was sensed by the 34410A. All the phase detector hadto do was
> > not short the bias current to ground.
> >
> >
> >
> AC coupling?
At the expense of phase shifts and temperature sensitivity, but yes.
And it makes it hard to sense a DC signal, if that's the intent.
> >>>>
> >>> I think that there are many top-end firewire soundcards. Whatever
the
> >>> music folk like the sound of would be a good place to start -
musicians'
> >>> well-trained hearing can be quite good. At least above 20 Hz.
> >>>
> >>> Actually, the people that make the AP192 do have firewire and usb
> >>> offerings:
> >>>
> >>> <http://www.m-audio.com/index.php?do=products.family&ID=recording>
> >>>
> >>>
> >>>
> >> I've looked at all of the M-Audio offerings.
> >> The more expensive ones have built in preamps plus 48V phantom
supplies,
> >> which can be switched off, however the presence of the switched +48V
> >> supply is perhaps an invitation to disaster.
> >>
> >
> > Given that capacitance to ground is more benefit than problem in this
> > application, I would be tempted to use a pair of back-to-back
rectifier
> > diodes as a clamp to protect the mixer IF output et al. The 48 volt
> > phantom supply will be short-circuit protected, so current will
> > automatically limit.
> >
> >
> A pair of coupling capacitors at the preamp output combined with clamp
> diodes to the amplifier power supply rails would work well even if the
> +48V can't be switched off.
> The +48V appears between the balanced pair conductors and ground.
> Unfortunately the power available from the phantom supply may not be
> sufficient to power the mixer preamp.
OK. The power limit does make protection easy. I gather that the limit
is a few milliamps, so even a 1N4148 would work.
> >> I've also looked at the specs for several other high end sound cards.
> >> Quite a few only have single ended inputs.
> >> Maybe, I should document the various cards and highlight their
> >> shortcomings etc for this application.
> >>
> >
> > That would be very useful.
> >
> I'll start on this shortly.
Thanks.
> >>>> The 4V rms input allows the mixer preamp to use devices like the
THAT
> >>>> 1646 to drive the balanced sound card inputs without degrading the
> >>>> noise floor too much.
> >>>>
> >>>>
> >>> Or build an isolation amp with some gain, and kill two birds with
one
> >>> stone?
> >>>
> >>>
> >>>
> >>>
> >> A low noise isolation amplifier with a frequency response down to 1Hz
or
> >> so without using a transformer may be difficult to do.
> >>
> >
> > People do make low noise common-base RF amplifiers, but 1 Hz would
yield
> > some pretty large bypass capacitors, even if the flicker noise can be
> > controlled well enough. I would consider using ultracapacitors, which
> > didn't exist until very recently, and of course have very large
> > capacitance values.
> >
> >
> A CB stage probably isn't optimum for the mixer preamp so that lower
> value caps can be used provided that they effectively short the
> amplifier input resistor Johnson noise at the frequencies of interest.
But is a CB stage adequate? Elimination of hum pickup is worth a lot.
> >>>> With a 1V rms full scale the noise floor degradation would be very
> >>>> obvious when using a THAT 1646 (equivalent devices are even
noisier).
> >>>> It may be better to use a mixer preamp with a transformer coupled
output
> >>>> stage using hybrid feedback to achieve a low frequency cutoff below
> >>>> 1Hz together with low noise.
> >>>>
> >>>>
> >>> With a transformer, even if toroidal, keeping hum out may prove
quite
> >>> difficult.
> >>>
> >>>
> >>>
> >> High end (eg Lundahl LL1517) line output audio transformers come with
mu
> >> metal screens and metal foil interwinding shields.
> >>
> >
> > They don't pass 1 Hz very well. I bet the rolloff is ~20 Hz.
> >
> >
>
> When driven conventionally the transformer cutoff is around 20Hz,
> however if one uses the appropriate driver with a controlled negative
> output R to cancel the transformer primary internal winding resistance,
> the low frequency response can be extended significantly. This also
> reduces the low frequency distortion.
> However individual adjustment of driver to suit transformer is required
> and tracking the winding resistance over temperature may be an issue.
That would certainly work. See next.
> > Certainly one can build a VLF transformer, but it will be a project
for
> > sure, and the transformer may be quite large.
> >
> >
> The transformers only weigh about 65g.
This weight estimate assumes the negative resistance circuit I assume, the
intent being to allow use of the Lundahl LL1517 transformer.
You might be interested in the following article: "A Broadband Active
Antenna for ELF Magnetic Fields" by John F. Sutton and G. Craig Spaniol"
in Physics Essays March 1993, Vol 6, #1, 1993. The negative-impedance
trick is expanded upon. Sutton also has some US patents on this. US
patent 5,296,866 covers the antenna, but is hard to understand without the
article.
> It may be simpler just to select a mixer for which the IF ground can be
> isolated from the RF and IF grounds.
> However a preamp with a transformer output may be useful if one uses a
> mixer where all the grounds are connected together by the package.
It has to be far easier to select the right mixer than to deal with a 1 Hz
transformer. And cheaper.
> >>>>>
> >>>>>
> >>>> No, I meant replace his 90 degree hybrids with a digital
equivalent.
> >>>>
> >>>>
> >>> I believe that his 90-degree hybrids are already digital.
> >>>
> >> I'm not convinced of that, if only because real time 10,000+ tap FIR
> >> filters at 30+MSPS are probably still impractical.
> >>
> >
> > I'm not convinced that one needs a 10,000-tap FIR to achieve this, and
Sam
> > Stein is one smart fellow. I recall some NASA patents from twenty
years
> > ago on how to get I+Q data from a single ADC, and while there was FIR
> > processing of some kind, there were only maybe 8 or 16 taps. And
Tayloe
> > (US patent 6,230,000) gets much the same effect with one resistor,
four
> > capacitors, an analog mux, and two differential amplifiers.
> >
> >
> I have read similar papers from that era on radar signal processing.
> They either used a Hilbert transform or a pair of digital filters whose
> outputs were in phase quadrature.
> The quadrature accuracy for a given bandwidth depends on the the number
> of taps.
> The beat frequencies (in a dual mixer system) won't match exactly and
> some correction for the resultant phase shift errors will need
> to be made.
> This may be less of a problem when the 2 beat frequency signals are
> identical in frequency and just differ in phase.
So long as we know the exact frequency, even if it isn't the exact desired
frequency, all may be well.
Joe
BG
Bruce Griffiths
Wed, Dec 17, 2008 3:21 AM
Joe
Joseph M Gwinn wrote:
Joe
Joseph M Gwinn wrote:
However the proposed remedy has little or no effect on the errors
by such bias currents (eg transistor base currents).
The series resistor could be reduced to zero without effect on the
offset due to the bias current. However the preamp offset due to the
source resistance would be reduced.
Hmm. It may be simpler than that. With the TDS3012B and 34410A
in parallel across the IF output of a mixer, the bias currents from
TDS3012B developed a voltage across the mixer load resistor, and this
voltage was sensed by the 34410A. All the phase detector hadto do was
not short the bias current to ground.
At the expense of phase shifts and temperature sensitivity, but yes.
And it makes it hard to sense a DC signal, if that's the intent.
I think that there are many top-end firewire soundcards. Whatever
music folk like the sound of would be a good place to start -
I've looked at all of the M-Audio offerings.
The more expensive ones have built in preamps plus 48V phantom
which can be switched off, however the presence of the switched +48V
supply is perhaps an invitation to disaster.
Given that capacitance to ground is more benefit than problem in this
application, I would be tempted to use a pair of back-to-back
diodes as a clamp to protect the mixer IF output et al. The 48 volt
phantom supply will be short-circuit protected, so current will
automatically limit.
A pair of coupling capacitors at the preamp output combined with clamp
diodes to the amplifier power supply rails would work well even if the
+48V can't be switched off.
The +48V appears between the balanced pair conductors and ground.
Unfortunately the power available from the phantom supply may not be
sufficient to power the mixer preamp.
OK. The power limit does make protection easy. I gather that the limit
is a few milliamps, so even a 1N4148 would work.
I've also looked at the specs for several other high end sound cards.
Quite a few only have single ended inputs.
Maybe, I should document the various cards and highlight their
shortcomings etc for this application.
That would be very useful.
I'll start on this shortly.
The 4V rms input allows the mixer preamp to use devices like the
1646 to drive the balanced sound card inputs without degrading the
noise floor too much.
Or build an isolation amp with some gain, and kill two birds with
A low noise isolation amplifier with a frequency response down to 1Hz
so without using a transformer may be difficult to do.
People do make low noise common-base RF amplifiers, but 1 Hz would
some pretty large bypass capacitors, even if the flicker noise can be
controlled well enough. I would consider using ultracapacitors, which
didn't exist until very recently, and of course have very large
capacitance values.
A CB stage probably isn't optimum for the mixer preamp so that lower
value caps can be used provided that they effectively short the
amplifier input resistor Johnson noise at the frequencies of interest.
But is a CB stage adequate? Elimination of hum pickup is worth a lot.
Text should have been:
A CB stage probably isn't optimum for the mixer preamp so that a low noise preamp with a higher input impedance can be employed allowing lower
value coupling caps to be used provided that they effectively short the amplifier input resistor Johnson noise at the frequencies of interest.
With a 1V rms full scale the noise floor degradation would be very
obvious when using a THAT 1646 (equivalent devices are even
It may be better to use a mixer preamp with a transformer coupled
stage using hybrid feedback to achieve a low frequency cutoff below
1Hz together with low noise.
With a transformer, even if toroidal, keeping hum out may prove
High end (eg Lundahl LL1517) line output audio transformers come with
metal screens and metal foil interwinding shields.
They don't pass 1 Hz very well. I bet the rolloff is ~20 Hz.
When driven conventionally the transformer cutoff is around 20Hz,
however if one uses the appropriate driver with a controlled negative
output R to cancel the transformer primary internal winding resistance,
the low frequency response can be extended significantly. This also
reduces the low frequency distortion.
However individual adjustment of driver to suit transformer is required
and tracking the winding resistance over temperature may be an issue.
That would certainly work. See next.
Certainly one can build a VLF transformer, but it will be a project
sure, and the transformer may be quite large.
The transformers only weigh about 65g.
This weight estimate assumes the negative resistance circuit I assume, the
intent being to allow use of the Lundahl LL1517 transformer.
You might be interested in the following article: "A Broadband Active
Antenna for ELF Magnetic Fields" by John F. Sutton and G. Craig Spaniol"
in Physics Essays March 1993, Vol 6, #1, 1993. The negative-impedance
trick is expanded upon. Sutton also has some US patents on this. US
patent 5,296,866 covers the antenna, but is hard to understand without the
article.
It may be simpler just to select a mixer for which the IF ground can be
isolated from the RF and IF grounds.
However a preamp with a transformer output may be useful if one uses a
mixer where all the grounds are connected together by the package.
It has to be far easier to select the right mixer than to deal with a 1 Hz
transformer. And cheaper.
I've been advocating this for some time, however one can then no longer
just buy an off the shelf mixer complete with SMA connectors, one has to
design and assemble a suitable PCB.
Obtaining suitable mixers for 5MHz and 10MHz input frequencies or even
100MHz is easy.
However for the higher microwave frequencies most mixers come complete
with connectors attached and share a common ground.
The noise problem with audio balanced drive chips can easily be overcome
with a discrete implementation.
That is discrete resistors and IC opamps.
No, I meant replace his 90 degree hybrids with a digital
I believe that his 90-degree hybrids are already digital.
I'm not convinced of that, if only because real time 10,000+ tap FIR
filters at 30+MSPS are probably still impractical.
I'm not convinced that one needs a 10,000-tap FIR to achieve this, and
Stein is one smart fellow. I recall some NASA patents from twenty
ago on how to get I+Q data from a single ADC, and while there was FIR
processing of some kind, there were only maybe 8 or 16 taps. And
(US patent 6,230,000) gets much the same effect with one resistor,
capacitors, an analog mux, and two differential amplifiers.
I have read similar papers from that era on radar signal processing.
They either used a Hilbert transform or a pair of digital filters whose
outputs were in phase quadrature.
The quadrature accuracy for a given bandwidth depends on the the number
of taps.
The beat frequencies (in a dual mixer system) won't match exactly and
some correction for the resultant phase shift errors will need
to be made.
This may be less of a problem when the 2 beat frequency signals are
identical in frequency and just differ in phase.
So long as we know the exact frequency, even if it isn't the exact desired
frequency, all may be well.
Joe
I'm reminded of some phase recovery algorithms used in phase shift
interferometry that largely negate the effect of small fixed phase errors.
If we can devise a suitable test setup then one could just log the
samples to a file for whatever sound card one has and make the data
available to others for analysis.
This allows a wide variety of sound cards to be evaluated without one
person having to test them all.
Bruce
Joe
Joseph M Gwinn wrote:
> Bruce,
>
>
> time-nuts-bounces@febo.com wrote on 12/16/2008 08:43:29 PM:
>
>
>> Joe
>>
>> Joseph M Gwinn wrote:
>>
>>> Bruce,
>>>
>>>
>>> time-nuts-bounces@febo.com wrote on 12/15/2008 06:42:59 PM:
>>>
>>>
> [snip]
>
>>>>>
>>>> However the proposed remedy has little or no effect on the errors
>>>>
> caused
>
>>>> by such bias currents (eg transistor base currents).
>>>> The series resistor could be reduced to zero without effect on the
>>>>
> mixer
>
>>>> offset due to the bias current. However the preamp offset due to the
>>>> source resistance would be reduced.
>>>>
>>>>
>>> Hmm. It may be simpler than that. With the TDS3012B and 34410A
>>>
> connected
>
>>> in parallel across the IF output of a mixer, the bias currents from
>>>
> the
>
>>> TDS3012B developed a voltage across the mixer load resistor, and this
>>> voltage was sensed by the 34410A. All the phase detector hadto do was
>>>
>
>
>>> not short the bias current to ground.
>>>
>>>
>>>
>>>
>> AC coupling?
>>
>
> At the expense of phase shifts and temperature sensitivity, but yes.
>
> And it makes it hard to sense a DC signal, if that's the intent.
>
>
>
>
>
>>>>> I think that there are many top-end firewire soundcards. Whatever
>>>>>
> the
>
>>>>> music folk like the sound of would be a good place to start -
>>>>>
> musicians'
>
>>>>> well-trained hearing can be quite good. At least above 20 Hz.
>>>>>
>>>>> Actually, the people that make the AP192 do have firewire and usb
>>>>> offerings:
>>>>>
>>>>> <http://www.m-audio.com/index.php?do=products.family&ID=recording>
>>>>>
>>>>>
>>>>>
>>>>>
>>>> I've looked at all of the M-Audio offerings.
>>>> The more expensive ones have built in preamps plus 48V phantom
>>>>
> supplies,
>
>>>> which can be switched off, however the presence of the switched +48V
>>>> supply is perhaps an invitation to disaster.
>>>>
>>>>
>>> Given that capacitance to ground is more benefit than problem in this
>>> application, I would be tempted to use a pair of back-to-back
>>>
> rectifier
>
>>> diodes as a clamp to protect the mixer IF output et al. The 48 volt
>>> phantom supply will be short-circuit protected, so current will
>>> automatically limit.
>>>
>>>
>>>
>> A pair of coupling capacitors at the preamp output combined with clamp
>> diodes to the amplifier power supply rails would work well even if the
>> +48V can't be switched off.
>> The +48V appears between the balanced pair conductors and ground.
>> Unfortunately the power available from the phantom supply may not be
>> sufficient to power the mixer preamp.
>>
>
> OK. The power limit does make protection easy. I gather that the limit
> is a few milliamps, so even a 1N4148 would work.
>
>
>
>>>> I've also looked at the specs for several other high end sound cards.
>>>> Quite a few only have single ended inputs.
>>>> Maybe, I should document the various cards and highlight their
>>>> shortcomings etc for this application.
>>>>
>>>>
>>> That would be very useful.
>>>
>>>
>> I'll start on this shortly.
>>
>
> Thanks.
>
>
>
>>>>>> The 4V rms input allows the mixer preamp to use devices like the
>>>>>>
> THAT
>
>>>>>> 1646 to drive the balanced sound card inputs without degrading the
>>>>>> noise floor too much.
>>>>>>
>>>>>>
>>>>>>
>>>>> Or build an isolation amp with some gain, and kill two birds with
>>>>>
> one
>
>>>>> stone?
>>>>>
>>>>>
>>>>>
>>>>>
>>>>>
>>>> A low noise isolation amplifier with a frequency response down to 1Hz
>>>>
> or
>
>>>> so without using a transformer may be difficult to do.
>>>>
>>>>
>>> People do make low noise common-base RF amplifiers, but 1 Hz would
>>>
> yield
>
>>> some pretty large bypass capacitors, even if the flicker noise can be
>>> controlled well enough. I would consider using ultracapacitors, which
>>>
>
>
>>> didn't exist until very recently, and of course have very large
>>> capacitance values.
>>>
>>>
>>>
>> A CB stage probably isn't optimum for the mixer preamp so that lower
>> value caps can be used provided that they effectively short the
>> amplifier input resistor Johnson noise at the frequencies of interest.
>>
>
> But is a CB stage adequate? Elimination of hum pickup is worth a lot.
>
>
>
Text should have been:
A CB stage probably isn't optimum for the mixer preamp so that a low noise preamp with a higher input impedance can be employed allowing lower
value coupling caps to be used provided that they effectively short the amplifier input resistor Johnson noise at the frequencies of interest.
>>>>>> With a 1V rms full scale the noise floor degradation would be very
>>>>>> obvious when using a THAT 1646 (equivalent devices are even
>>>>>>
> noisier).
>
>>>>>> It may be better to use a mixer preamp with a transformer coupled
>>>>>>
> output
>
>>>>>> stage using hybrid feedback to achieve a low frequency cutoff below
>>>>>>
>
>
>>>>>> 1Hz together with low noise.
>>>>>>
>>>>>>
>>>>>>
>>>>> With a transformer, even if toroidal, keeping hum out may prove
>>>>>
> quite
>
>>>>> difficult.
>>>>>
>>>>>
>>>>>
>>>>>
>>>> High end (eg Lundahl LL1517) line output audio transformers come with
>>>>
> mu
>
>>>> metal screens and metal foil interwinding shields.
>>>>
>>>>
>>> They don't pass 1 Hz very well. I bet the rolloff is ~20 Hz.
>>>
>>>
>>>
>> When driven conventionally the transformer cutoff is around 20Hz,
>> however if one uses the appropriate driver with a controlled negative
>> output R to cancel the transformer primary internal winding resistance,
>> the low frequency response can be extended significantly. This also
>> reduces the low frequency distortion.
>> However individual adjustment of driver to suit transformer is required
>> and tracking the winding resistance over temperature may be an issue.
>>
>
> That would certainly work. See next.
>
>
>
>>> Certainly one can build a VLF transformer, but it will be a project
>>>
> for
>
>>> sure, and the transformer may be quite large.
>>>
>>>
>>>
>> The transformers only weigh about 65g.
>>
>
> This weight estimate assumes the negative resistance circuit I assume, the
> intent being to allow use of the Lundahl LL1517 transformer.
>
>
Yes.
> You might be interested in the following article: "A Broadband Active
> Antenna for ELF Magnetic Fields" by John F. Sutton and G. Craig Spaniol"
> in Physics Essays March 1993, Vol 6, #1, 1993. The negative-impedance
> trick is expanded upon. Sutton also has some US patents on this. US
> patent 5,296,866 covers the antenna, but is hard to understand without the
> article.
>
>
>
>> It may be simpler just to select a mixer for which the IF ground can be
>> isolated from the RF and IF grounds.
>> However a preamp with a transformer output may be useful if one uses a
>> mixer where all the grounds are connected together by the package.
>>
>
> It has to be far easier to select the right mixer than to deal with a 1 Hz
> transformer. And cheaper.
>
>
I've been advocating this for some time, however one can then no longer
just buy an off the shelf mixer complete with SMA connectors, one has to
design and assemble a suitable PCB.
Obtaining suitable mixers for 5MHz and 10MHz input frequencies or even
100MHz is easy.
However for the higher microwave frequencies most mixers come complete
with connectors attached and share a common ground.
The noise problem with audio balanced drive chips can easily be overcome
with a discrete implementation.
That is discrete resistors and IC opamps.
>
>
>
>>>>>>>
>>>>>> No, I meant replace his 90 degree hybrids with a digital
>>>>>>
> equivalent.
>
>>>>>>
>>>>> I believe that his 90-degree hybrids are already digital.
>>>>>
>>>>>
>>>> I'm not convinced of that, if only because real time 10,000+ tap FIR
>>>> filters at 30+MSPS are probably still impractical.
>>>>
>>>>
>>> I'm not convinced that one needs a 10,000-tap FIR to achieve this, and
>>>
> Sam
>
>>> Stein is one smart fellow. I recall some NASA patents from twenty
>>>
> years
>
>>> ago on how to get I+Q data from a single ADC, and while there was FIR
>>> processing of some kind, there were only maybe 8 or 16 taps. And
>>>
> Tayloe
>
>>> (US patent 6,230,000) gets much the same effect with one resistor,
>>>
> four
>
>>> capacitors, an analog mux, and two differential amplifiers.
>>>
>>>
>>>
>> I have read similar papers from that era on radar signal processing.
>> They either used a Hilbert transform or a pair of digital filters whose
>> outputs were in phase quadrature.
>> The quadrature accuracy for a given bandwidth depends on the the number
>> of taps.
>> The beat frequencies (in a dual mixer system) won't match exactly and
>> some correction for the resultant phase shift errors will need
>> to be made.
>> This may be less of a problem when the 2 beat frequency signals are
>> identical in frequency and just differ in phase.
>>
>
> So long as we know the exact frequency, even if it isn't the exact desired
> frequency, all may be well.
>
> Joe
>
>
>
I'm reminded of some phase recovery algorithms used in phase shift
interferometry that largely negate the effect of small fixed phase errors.
If we can devise a suitable test setup then one could just log the
samples to a file for whatever sound card one has and make the data
available to others for analysis.
This allows a wide variety of sound cards to be evaluated without one
person having to test them all.
Bruce
JM
Joseph M Gwinn
Wed, Dec 17, 2008 7:21 PM
Joe
Joseph M Gwinn wrote:
A CB stage probably isn't optimum for the mixer preamp so that lower
value caps can be used provided that they effectively short the
amplifier input resistor Johnson noise at the frequencies ofinterest.
But is a CB stage adequate? Elimination of hum pickup is worth a lot.
[Bruce] Text should have been:
A CB stage probably isn't optimum for the mixer preamp so that
a low noise preamp with a higher input impedance can be
employed allowing lower
value coupling caps to be used provided that they effectively
short the amplifier input resistor Johnson noise at the
frequencies of interest.
OK. The form of input amplifier is one of the tradeoffs one must make.
It may be simpler just to select a mixer for which the IF ground can
isolated from the RF and LO grounds.
However a preamp with a transformer output may be useful if one uses
mixer where all the grounds are connected together by the package.
It has to be far easier to select the right mixer than to deal with a
transformer. And cheaper.
I've been advocating this for some time, however one can then no longer
just buy an off the shelf mixer complete with SMA connectors, one has to
design and assemble a suitable PCB.
Actually, for quantity one, I don't bother with PCBs. I use 1/16 inch
thick glass-epoxy Vectorboard with a 0.1" hole pitch, and thread bare wire
through the holes. Given that these are effectively prototype boards, the
ease of wholesale change is very useful. For transmission lines, I would
just run miniature coax or homebrew twisted pair from point to point on
the board, although it has not been neecessary yet. If microphonics is an
issue, the board can be conformal coated to glue the wires in place.
The traditional coat, wax, allows easy alteration and repair. In the
1960s, a friend was building things using RTL (Resistor Transistor Logic)
ICs and solderable magnet wire between the leads. What a rats' nest that
was. The problem was how to package this so it wouldn't fail when
bumped. The solution was to build the circuit on a sheet of vectorboard
in a 12" by 12" rectangular baking pan, and then fill the pan with hot
wax. Whenever something had to be changed, melt the wax, pour it off,
make the change, pour the wax back into the pan, allow to cool.
Modern ICs connected with point-to-point wires and potted in wax - it's an
odd mix of technology ages. Like implementing stone-age tools with modern
ceramics.
If I were building receivers, I suppose I would be forced to use
surface-mount comoponents and PCBs.
Obtaining suitable mixers for 5MHz and 10MHz input frequencies or even
However for the higher microwave frequencies most mixers come complete
with connectors attached and share a common ground.
True. However, I don't think we will be going from 1 GHz to 1 Hz in a
single step, and the last mixer can have separate grounds.
The noise problem with audio balanced drive chips can easily be overcome
with a discrete implementation.
That is discrete resistors and IC opamps.
Yes, even on vectorboard. I do it all the time.
Eventually, the supply of through-hole components will dry up, but it
hasn't happened yet, and some components from the 1970s are still
available. Even if the original manufacturer is long dead.
I have read similar papers from that era on radar signal processing.
They either used a Hilbert transform or a pair of digital filters
outputs were in phase quadrature.
The quadrature accuracy for a given bandwidth depends on the number
of taps.
The beat frequencies (in a dual mixer system) won't match exactly and
some correction for the resultant phase shift errors will need
to be made.
This may be less of a problem when the 2 beat frequency signals are
identical in frequency and just differ in phase.
So long as we know the exact frequency, even if it isn't the exact
frequency, all may be well.
Joe
I'm reminded of some phase recovery algorithms used in phase shift
interferometry that largely negate the effect of small fixed
phase errors.
Yes. A detailed math analysis of the test setup will be needed.
If we can devise a suitable test setup then one could just log the
samples to a file for whatever sound card one has and make the data
available to others for analysis.
This allows a wide variety of sound cards to be evaluated without one
person having to test them all.
And evaluation of the same test data by multiple people using different
tools also allows us to distinguish test artifacts from processing
artifacts.
Joe
Bruce,
time-nuts-bounces@febo.com wrote on 12/16/2008 10:21:55 PM:
> Joe
>
> Joseph M Gwinn wrote:
> > Bruce,
> >
[snip]
> >
Bruce wrote:
> >>
> >> A CB stage probably isn't optimum for the mixer preamp so that lower
> >> value caps can be used provided that they effectively short the
> >> amplifier input resistor Johnson noise at the frequencies ofinterest.
> >>
> >
> > But is a CB stage adequate? Elimination of hum pickup is worth a lot.
> >
> >
> >
> [Bruce] Text should have been:
>
> A CB stage probably isn't optimum for the mixer preamp so that
> a low noise preamp with a higher input impedance can be
> employed allowing lower
> value coupling caps to be used provided that they effectively
> short the amplifier input resistor Johnson noise at the
> frequencies of interest.
OK. The form of input amplifier is one of the tradeoffs one must make.
> >> It may be simpler just to select a mixer for which the IF ground can
be
> >> isolated from the RF and LO grounds.
> >> However a preamp with a transformer output may be useful if one uses
a
> >> mixer where all the grounds are connected together by the package.
> >>
> >
> > It has to be far easier to select the right mixer than to deal with a
1 Hz
> > transformer. And cheaper.
> >
> >
>
> I've been advocating this for some time, however one can then no longer
> just buy an off the shelf mixer complete with SMA connectors, one has to
> design and assemble a suitable PCB.
Actually, for quantity one, I don't bother with PCBs. I use 1/16 inch
thick glass-epoxy Vectorboard with a 0.1" hole pitch, and thread bare wire
through the holes. Given that these are effectively prototype boards, the
ease of wholesale change is very useful. For transmission lines, I would
just run miniature coax or homebrew twisted pair from point to point on
the board, although it has not been neecessary yet. If microphonics is an
issue, the board can be conformal coated to glue the wires in place.
The traditional coat, wax, allows easy alteration and repair. In the
1960s, a friend was building things using RTL (Resistor Transistor Logic)
ICs and solderable magnet wire between the leads. What a rats' nest that
was. The problem was how to package this so it wouldn't fail when
bumped. The solution was to build the circuit on a sheet of vectorboard
in a 12" by 12" rectangular baking pan, and then fill the pan with hot
wax. Whenever something had to be changed, melt the wax, pour it off,
make the change, pour the wax back into the pan, allow to cool.
Modern ICs connected with point-to-point wires and potted in wax - it's an
odd mix of technology ages. Like implementing stone-age tools with modern
ceramics.
If I were building receivers, I suppose I would be forced to use
surface-mount comoponents and PCBs.
> Obtaining suitable mixers for 5MHz and 10MHz input frequencies or even
100MHz is easy.
> However for the higher microwave frequencies most mixers come complete
> with connectors attached and share a common ground.
True. However, I don't think we will be going from 1 GHz to 1 Hz in a
single step, and the last mixer can have separate grounds.
> The noise problem with audio balanced drive chips can easily be overcome
> with a discrete implementation.
> That is discrete resistors and IC opamps.
Yes, even on vectorboard. I do it all the time.
Eventually, the supply of through-hole components will dry up, but it
hasn't happened yet, and some components from the 1970s are still
available. Even if the original manufacturer is long dead.
> >
> >> I have read similar papers from that era on radar signal processing.
> >> They either used a Hilbert transform or a pair of digital filters
whose
> >> outputs were in phase quadrature.
> >> The quadrature accuracy for a given bandwidth depends on the number
> >> of taps.
> >> The beat frequencies (in a dual mixer system) won't match exactly and
> >> some correction for the resultant phase shift errors will need
> >> to be made.
> >> This may be less of a problem when the 2 beat frequency signals are
> >> identical in frequency and just differ in phase.
> >>
> >
> > So long as we know the exact frequency, even if it isn't the exact
desired
> > frequency, all may be well.
> >
> > Joe
> >
> >
> >
> I'm reminded of some phase recovery algorithms used in phase shift
> interferometry that largely negate the effect of small fixed
> phase errors.
Yes. A detailed math analysis of the test setup will be needed.
> If we can devise a suitable test setup then one could just log the
> samples to a file for whatever sound card one has and make the data
> available to others for analysis.
Yes.
> This allows a wide variety of sound cards to be evaluated without one
> person having to test them all.
And evaluation of the same test data by multiple people using different
tools also allows us to distinguish test artifacts from processing
artifacts.
Joe
BG
Bruce Griffiths
Wed, Dec 17, 2008 8:43 PM
Joe
Joseph M Gwinn wrote:
Joe
Joseph M Gwinn wrote:
A CB stage probably isn't optimum for the mixer preamp so that lower
value caps can be used provided that they effectively short the
amplifier input resistor Johnson noise at the frequencies ofinterest.
But is a CB stage adequate? Elimination of hum pickup is worth a lot.
[Bruce] Text should have been:
A CB stage probably isn't optimum for the mixer preamp so that
a low noise preamp with a higher input impedance can be
employed allowing lower
value coupling caps to be used provided that they effectively
short the amplifier input resistor Johnson noise at the
frequencies of interest.
OK. The form of input amplifier is one of the tradeoffs one must make.
It may be simpler just to select a mixer for which the IF ground can
isolated from the RF and LO grounds.
However a preamp with a transformer output may be useful if one uses
mixer where all the grounds are connected together by the package.
It has to be far easier to select the right mixer than to deal with a
transformer. And cheaper.
I've been advocating this for some time, however one can then no longer
just buy an off the shelf mixer complete with SMA connectors, one has to
design and assemble a suitable PCB.
Actually, for quantity one, I don't bother with PCBs. I use 1/16 inch
thick glass-epoxy Vectorboard with a 0.1" hole pitch, and thread bare wire
through the holes. Given that these are effectively prototype boards, the
ease of wholesale change is very useful. For transmission lines, I would
just run miniature coax or homebrew twisted pair from point to point on
the board, although it has not been neecessary yet. If microphonics is an
issue, the board can be conformal coated to glue the wires in place.
The traditional coat, wax, allows easy alteration and repair. In the
1960s, a friend was building things using RTL (Resistor Transistor Logic)
ICs and solderable magnet wire between the leads. What a rats' nest that
was. The problem was how to package this so it wouldn't fail when
bumped. The solution was to build the circuit on a sheet of vectorboard
in a 12" by 12" rectangular baking pan, and then fill the pan with hot
wax. Whenever something had to be changed, melt the wax, pour it off,
make the change, pour the wax back into the pan, allow to cool.
Modern ICs connected with point-to-point wires and potted in wax - it's an
odd mix of technology ages. Like implementing stone-age tools with modern
ceramics.
If I were building receivers, I suppose I would be forced to use
surface-mount comoponents and PCBs.
Obtaining suitable mixers for 5MHz and 10MHz input frequencies or even
However for the higher microwave frequencies most mixers come complete
with connectors attached and share a common ground.
True. However, I don't think we will be going from 1 GHz to 1 Hz in a
single step, and the last mixer can have separate grounds.
An upper limit of at least 100MHz should be feasible for the final mixer.
A dual conversion scheme will be essential if one uses a triple balanced
or similar first mixer that has an IF response that doesn't extend down
to the low frequencies that a sound card can use.
The noise problem with audio balanced drive chips can easily be overcome
with a discrete implementation.
That is discrete resistors and IC opamps.
Yes, even on vectorboard. I do it all the time.
Eventually, the supply of through-hole components will dry up, but it
hasn't happened yet, and some components from the 1970s are still
available. Even if the original manufacturer is long dead.
I have read similar papers from that era on radar signal processing.
They either used a Hilbert transform or a pair of digital filters
outputs were in phase quadrature.
The quadrature accuracy for a given bandwidth depends on the number
of taps.
The beat frequencies (in a dual mixer system) won't match exactly and
some correction for the resultant phase shift errors will need
to be made.
This may be less of a problem when the 2 beat frequency signals are
identical in frequency and just differ in phase.
So long as we know the exact frequency, even if it isn't the exact
frequency, all may be well.
Joe
I'm reminded of some phase recovery algorithms used in phase shift
interferometry that largely negate the effect of small fixed
phase errors.
Yes. A detailed math analysis of the test setup will be needed.
If we can devise a suitable test setup then one could just log the
samples to a file for whatever sound card one has and make the data
available to others for analysis.
This allows a wide variety of sound cards to be evaluated without one
person having to test them all.
And evaluation of the same test data by multiple people using different
tools also allows us to distinguish test artifacts from processing
artifacts.
Joe
Proposed test setup:
(preliminary to be refined)
Drive 2 sound card inputs in parallel with the same source.
Source amplitude:
Max sound card input -3dB
Sources:
-
Wien bridge or equivalent (eg state variable oscillator with soft
clamping) relatively low distortion oscillator.
-
Buffered low pass filtered output of binary divider driven by a
crystal oscillator
Test frequencies:
100Hz
1kHz
Sound card sample rate:
~24KSPS
Test duration:
1000 sec
File format:
Wave file??
Resolution 24 bits for 24 bit sound cards, 16 bits for 16bit and lower
resolution sound cards, etc.
Some refinement of sample rates and test duration is required to keep
the data file sizes manageable.
With a 24 bit sound card sampling at 96KSPS or 192KSPS for 1000sec can
produce file sizes of 1GB or more.
Some preprocessing (low pass filter and decimation) may also be required.
Bruce
Joe
Joseph M Gwinn wrote:
> Bruce,
>
>
> time-nuts-bounces@febo.com wrote on 12/16/2008 10:21:55 PM:
>
>
>> Joe
>>
>> Joseph M Gwinn wrote:
>>
>>> Bruce,
>>>
>>>
> [snip]
>
> Bruce wrote:
>
>>>> A CB stage probably isn't optimum for the mixer preamp so that lower
>>>> value caps can be used provided that they effectively short the
>>>> amplifier input resistor Johnson noise at the frequencies ofinterest.
>>>>
>>>>
>>> But is a CB stage adequate? Elimination of hum pickup is worth a lot.
>>>
>>>
>>>
>>>
>> [Bruce] Text should have been:
>>
>> A CB stage probably isn't optimum for the mixer preamp so that
>> a low noise preamp with a higher input impedance can be
>> employed allowing lower
>> value coupling caps to be used provided that they effectively
>> short the amplifier input resistor Johnson noise at the
>> frequencies of interest.
>>
>
> OK. The form of input amplifier is one of the tradeoffs one must make.
>
>
>
>
>>>> It may be simpler just to select a mixer for which the IF ground can
>>>>
> be
>
>>>> isolated from the RF and LO grounds.
>>>> However a preamp with a transformer output may be useful if one uses
>>>>
> a
>
>>>> mixer where all the grounds are connected together by the package.
>>>>
>>>>
>>> It has to be far easier to select the right mixer than to deal with a
>>>
> 1 Hz
>
>>> transformer. And cheaper.
>>>
>>>
>>>
>> I've been advocating this for some time, however one can then no longer
>> just buy an off the shelf mixer complete with SMA connectors, one has to
>> design and assemble a suitable PCB.
>>
>
> Actually, for quantity one, I don't bother with PCBs. I use 1/16 inch
> thick glass-epoxy Vectorboard with a 0.1" hole pitch, and thread bare wire
> through the holes. Given that these are effectively prototype boards, the
> ease of wholesale change is very useful. For transmission lines, I would
> just run miniature coax or homebrew twisted pair from point to point on
> the board, although it has not been neecessary yet. If microphonics is an
> issue, the board can be conformal coated to glue the wires in place.
>
> The traditional coat, wax, allows easy alteration and repair. In the
> 1960s, a friend was building things using RTL (Resistor Transistor Logic)
> ICs and solderable magnet wire between the leads. What a rats' nest that
> was. The problem was how to package this so it wouldn't fail when
> bumped. The solution was to build the circuit on a sheet of vectorboard
> in a 12" by 12" rectangular baking pan, and then fill the pan with hot
> wax. Whenever something had to be changed, melt the wax, pour it off,
> make the change, pour the wax back into the pan, allow to cool.
>
> Modern ICs connected with point-to-point wires and potted in wax - it's an
> odd mix of technology ages. Like implementing stone-age tools with modern
> ceramics.
>
> If I were building receivers, I suppose I would be forced to use
> surface-mount comoponents and PCBs.
>
>
>
>> Obtaining suitable mixers for 5MHz and 10MHz input frequencies or even
>>
> 100MHz is easy.
>
>> However for the higher microwave frequencies most mixers come complete
>> with connectors attached and share a common ground.
>>
>
> True. However, I don't think we will be going from 1 GHz to 1 Hz in a
> single step, and the last mixer can have separate grounds.
>
>
>
An upper limit of at least 100MHz should be feasible for the final mixer.
A dual conversion scheme will be essential if one uses a triple balanced
or similar first mixer that has an IF response that doesn't extend down
to the low frequencies that a sound card can use.
>> The noise problem with audio balanced drive chips can easily be overcome
>> with a discrete implementation.
>> That is discrete resistors and IC opamps.
>>
>
> Yes, even on vectorboard. I do it all the time.
>
> Eventually, the supply of through-hole components will dry up, but it
> hasn't happened yet, and some components from the 1970s are still
> available. Even if the original manufacturer is long dead.
>
>
>
>>>> I have read similar papers from that era on radar signal processing.
>>>> They either used a Hilbert transform or a pair of digital filters
>>>>
> whose
>
>>>> outputs were in phase quadrature.
>>>> The quadrature accuracy for a given bandwidth depends on the number
>>>> of taps.
>>>> The beat frequencies (in a dual mixer system) won't match exactly and
>>>> some correction for the resultant phase shift errors will need
>>>> to be made.
>>>> This may be less of a problem when the 2 beat frequency signals are
>>>> identical in frequency and just differ in phase.
>>>>
>>>>
>>> So long as we know the exact frequency, even if it isn't the exact
>>>
> desired
>
>>> frequency, all may be well.
>>>
>>> Joe
>>>
>>>
>>>
>>>
>> I'm reminded of some phase recovery algorithms used in phase shift
>> interferometry that largely negate the effect of small fixed
>> phase errors.
>>
>
> Yes. A detailed math analysis of the test setup will be needed.
>
>
>
>> If we can devise a suitable test setup then one could just log the
>> samples to a file for whatever sound card one has and make the data
>> available to others for analysis.
>>
>
> Yes.
>
>
>
>> This allows a wide variety of sound cards to be evaluated without one
>> person having to test them all.
>>
>
> And evaluation of the same test data by multiple people using different
> tools also allows us to distinguish test artifacts from processing
> artifacts.
>
>
> Joe
>
Proposed test setup:
(preliminary to be refined)
Drive 2 sound card inputs in parallel with the same source.
Source amplitude:
Max sound card input -3dB
Sources:
1) Wien bridge or equivalent (eg state variable oscillator with soft
clamping) relatively low distortion oscillator.
2) Buffered low pass filtered output of binary divider driven by a
crystal oscillator
Test frequencies:
100Hz
1kHz
Sound card sample rate:
~24KSPS
Test duration:
1000 sec
File format:
Wave file??
Resolution 24 bits for 24 bit sound cards, 16 bits for 16bit and lower
resolution sound cards, etc.
Some refinement of sample rates and test duration is required to keep
the data file sizes manageable.
With a 24 bit sound card sampling at 96KSPS or 192KSPS for 1000sec can
produce file sizes of 1GB or more.
Some preprocessing (low pass filter and decimation) may also be required.
Bruce
JM
Joseph M Gwinn
Wed, Dec 17, 2008 10:36 PM
I'll look into doing this [MDEV and ADEV].
Real time filtering and decimation may be impractical, in the short
at least, as most signal processing libraries only process 16 bit
samples.
Most real time spectrum analysis programs are similarly afflicted in
that they only process 16 bit samples.
I don't see why we would need realtime filtering. Data reduction can
offline, so we ought to be able to use 32-bit or 64-bit arithmetic.
Given that we will inspect Allan Deviation data in a log-log plot, one
save much processing time by spacing the tau values to be computed
uniformly in log tau. I've played with this in Mathematica, and it
work and yields a large speedup factor. It should also help with
and Win2K limits. One trick is to ensure that one computes each tau
at most once. This check is needed because with close spacing, the
function will yield the same tau values multiple times for small
Joe
Real time processing certainly isn't required to characterise the
However some may be tempted to do this, it's probably possible with a
sufficently fast machine.
If we are looking for thermal effects, with a characteristic timescale of
tens of minutes to hours, the concept of realtime can be very generous.
I was just highlighting a problem with some available signal processing
libraries which may have been developed before sound cards with
resolutions of more than 16 bits became available.
Some (perhaps most) real time spectrum display software also has this
problem (eg baudline, Virtins etc).
I would assume that there are newer libraries now, and libraries available
as source code can be updated and recompiled.
20 Log[ 2^16 ]= 96 dB. This isn't awful, and we will get the entire
16-bit range if the ADC is 24 bits (with ENOB of 19-20 bits) and we scale
and round the samples properly.
As I think about it, the 16-bit limit must be for embedded signal
processing code, and math libraries intended for use on ordinary computers
will be at least 32 bit or 64-bit float, so it should not be difficult to
come by the necessary code.
It isn't necessary to use a pair of mixers and an offset source to
characterise the sound card, driving both sound card inputs from the
same audio source should suffice.
Yes. One input at a time, with the other input shorted, so we can also
see the crosstalk.
The audio source need not have low ultra low distortion (the IF output
signals in a dual mixer system won't have ultra low distortion) or very
high frequency stability (the IF output signals in a dual mixer system
won't necessarily have particularly high frequency stability).
But ... but ... but ... I thought Time Nuts used only atomic frequency
refs, and crystals only if oven stabilized.
A standard RC audio oscillator with distortion lower than 1% or so
should suffice.
At least the resultant frequency fluctuations should thoroughly exercise
the phase extraction algorithms.
Another option would be to low pass filter the output of a divider.
Using a sound card to generate the test signal is also possible but it
can potentially introduce extraneous noise and other artifacts such as
phase truncation spurs.
If one chooses the test frequencies correctly, one can eliminate the
spurs. The trick is to choose frequencies that lead to DDS tuning words
that have zeroes in the accumulator bits that are truncated (that is, do
not make it into the sin/cos lookup table).
Step one of planning an experiment is to decide on the objectives. The
large scale objective is to determine which sound cards are suitable for a
number of time-related tasks, so we should enumerate and describe these
tasks.
Task 1. The immediate task is to receive and digitize the sinewave output
from a mixer, the sinewave being the offset frequency coming out of a DMTD
apparatus. Offset frequencies will range from 10 Hz to 1 KHz, will be
known with great precision from the design of the apparatus, and need not
be measured. This sinewave is high amplitude (at least one volt rms,
matched to the needs of the soundcard) and very high SNR. This will be
done in two channels in parallel. The signals are at the same frequency
but differ in phase. The intent is to extract the phases of these two
sinewaves, the difference in phase being the ultimate output. The phase
of a signal will be extracted by least-squares fitting of a sine function
to the measured data.
And so on. We need to list the tasks, and to use this task list to inform
the experiment design.
Bruce,
time-nuts-bounces@febo.com wrote on 12/16/2008 09:05:55 PM:
> Joseph M Gwinn wrote:
> > Bruce,
> >
> >
> > time-nuts-bounces@febo.com wrote on 12/15/2008 05:31:27 PM:
> >
> >
> >> Joseph M Gwinn wrote:
> >>
> >>> Bruce,
> >>>
> >>>
> >>> time-nuts-bounces@febo.com wrote on 12/15/2008 04:34:34 PM:
> >>>
> >>>
> > [snip]
> >>
> >> I'll look into doing this [MDEV and ADEV].
> >> Real time filtering and decimation may be impractical, in the short
term
> >> at least, as most signal processing libraries only process 16 bit
> >> samples.
> >
> >> Most real time spectrum analysis programs are similarly afflicted in
> >> that they only process 16 bit samples.
> >>
> >
> > I don't see why we would need realtime filtering. Data reduction can
be
> > offline, so we ought to be able to use 32-bit or 64-bit arithmetic.
> >
> > Given that we will inspect Allan Deviation data in a log-log plot, one
can
> > save much processing time by spacing the tau values to be computed
> > uniformly in log tau. I've played with this in Mathematica, and it
does
> > work and yields a large speedup factor. It should also help with
Plotter
> > and Win2K limits. One trick is to ensure that one computes each tau
value
> > at most once. This check is needed because with close spacing, the
round
> > function will yield the same tau values multiple times for small
values of
> > tau.
> >
> > Joe
> >
> >
> Joe
>
> Real time processing certainly isn't required to characterise the
performance.
> However some may be tempted to do this, it's probably possible with a
sufficently fast machine.
If we are looking for thermal effects, with a characteristic timescale of
tens of minutes to hours, the concept of realtime can be very generous.
> I was just highlighting a problem with some available signal processing
> libraries which may have been developed before sound cards with
> resolutions of more than 16 bits became available.
> Some (perhaps most) real time spectrum display software also has this
> problem (eg baudline, Virtins etc).
I would assume that there are newer libraries now, and libraries available
as source code can be updated and recompiled.
20 Log[ 2^16 ]= 96 dB. This isn't awful, and we will get the entire
16-bit range if the ADC is 24 bits (with ENOB of 19-20 bits) and we scale
and round the samples properly.
As I think about it, the 16-bit limit must be for embedded signal
processing code, and math libraries intended for use on ordinary computers
will be at least 32 bit or 64-bit float, so it should not be difficult to
come by the necessary code.
> It isn't necessary to use a pair of mixers and an offset source to
> characterise the sound card, driving both sound card inputs from the
> same audio source should suffice.
Yes. One input at a time, with the other input shorted, so we can also
see the crosstalk.
> The audio source need not have low ultra low distortion (the IF output
> signals in a dual mixer system won't have ultra low distortion) or very
> high frequency stability (the IF output signals in a dual mixer system
> won't necessarily have particularly high frequency stability).
But ... but ... but ... I thought Time Nuts used only atomic frequency
refs, and crystals only if oven stabilized.
> A standard RC audio oscillator with distortion lower than 1% or so
> should suffice.
> At least the resultant frequency fluctuations should thoroughly exercise
> the phase extraction algorithms.
>
> Another option would be to low pass filter the output of a divider.
> Using a sound card to generate the test signal is also possible but it
> can potentially introduce extraneous noise and other artifacts such as
> phase truncation spurs.
If one chooses the test frequencies correctly, one can eliminate the
spurs. The trick is to choose frequencies that lead to DDS tuning words
that have zeroes in the accumulator bits that are truncated (that is, do
not make it into the sin/cos lookup table).
Step one of planning an experiment is to decide on the objectives. The
large scale objective is to determine which sound cards are suitable for a
number of time-related tasks, so we should enumerate and describe these
tasks.
Task 1. The immediate task is to receive and digitize the sinewave output
from a mixer, the sinewave being the offset frequency coming out of a DMTD
apparatus. Offset frequencies will range from 10 Hz to 1 KHz, will be
known with great precision from the design of the apparatus, and need not
be measured. This sinewave is high amplitude (at least one volt rms,
matched to the needs of the soundcard) and very high SNR. This will be
done in two channels in parallel. The signals are at the same frequency
but differ in phase. The intent is to extract the phases of these two
sinewaves, the difference in phase being the ultimate output. The phase
of a signal will be extracted by least-squares fitting of a sine function
to the measured data.
And so on. We need to list the tasks, and to use this task list to inform
the experiment design.
BG
Bruce Griffiths
Wed, Dec 17, 2008 11:26 PM
I'll look into doing this [MDEV and ADEV].
Real time filtering and decimation may be impractical, in the short
at least, as most signal processing libraries only process 16 bit
samples.
Most real time spectrum analysis programs are similarly afflicted in
that they only process 16 bit samples.
I don't see why we would need realtime filtering. Data reduction can
offline, so we ought to be able to use 32-bit or 64-bit arithmetic.
Given that we will inspect Allan Deviation data in a log-log plot, one
save much processing time by spacing the tau values to be computed
uniformly in log tau. I've played with this in Mathematica, and it
work and yields a large speedup factor. It should also help with
and Win2K limits. One trick is to ensure that one computes each tau
at most once. This check is needed because with close spacing, the
function will yield the same tau values multiple times for small
Joe
Real time processing certainly isn't required to characterise the
However some may be tempted to do this, it's probably possible with a
sufficently fast machine.
If we are looking for thermal effects, with a characteristic timescale of
tens of minutes to hours, the concept of realtime can be very generous.
I was just highlighting a problem with some available signal processing
libraries which may have been developed before sound cards with
resolutions of more than 16 bits became available.
Some (perhaps most) real time spectrum display software also has this
problem (eg baudline, Virtins etc).
I would assume that there are newer libraries now, and libraries available
as source code can be updated and recompiled.
20 Log[ 2^16 ]= 96 dB. This isn't awful, and we will get the entire
16-bit range if the ADC is 24 bits (with ENOB of 19-20 bits) and we scale
and round the samples properly.
As I think about it, the 16-bit limit must be for embedded signal
processing code, and math libraries intended for use on ordinary computers
will be at least 32 bit or 64-bit float, so it should not be difficult to
come by the necessary code.
It isn't necessary to use a pair of mixers and an offset source to
characterise the sound card, driving both sound card inputs from the
same audio source should suffice.
Yes. One input at a time, with the other input shorted, so we can also
see the crosstalk.
The audio source need not have low ultra low distortion (the IF output
signals in a dual mixer system won't have ultra low distortion) or very
high frequency stability (the IF output signals in a dual mixer system
won't necessarily have particularly high frequency stability).
But ... but ... but ... I thought Time Nuts used only atomic frequency
refs, and crystals only if oven stabilized.
If one mixes down a 10MHz source to 100Hz the fractional frequency
instability (of the beat frequency) is magnified by a factor of 1E5 over
that of the 10MHz source.
This assumes that the offset source has significantly lower instability
than the source under test.
In the special case when the offset source and the test source are phase
locked the offset frequency will have much greater stability.
A standard RC audio oscillator with distortion lower than 1% or so
should suffice.
At least the resultant frequency fluctuations should thoroughly exercise
the phase extraction algorithms.
Another option would be to low pass filter the output of a divider.
Using a sound card to generate the test signal is also possible but it
can potentially introduce extraneous noise and other artifacts such as
phase truncation spurs.
If one chooses the test frequencies correctly, one can eliminate the
spurs. The trick is to choose frequencies that lead to DDS tuning words
that have zeroes in the accumulator bits that are truncated (that is, do
not make it into the sin/cos lookup table).
This just adds another layer of complexity for little immediate gain.
Making the algorithms robust against small drifts in beat frequency is
more useful in the general case (when 2 different test sources are being
compared) than just assuming that the the beat frequency is very stable
and fixed.
Step one of planning an experiment is to decide on the objectives. The
large scale objective is to determine which sound cards are suitable for a
number of time-related tasks, so we should enumerate and describe these
tasks.
Task 1. The immediate task is to receive and digitize the sinewave output
from a mixer, the sinewave being the offset frequency coming out of a DMTD
apparatus. Offset frequencies will range from 10 Hz to 1 KHz, will be
known with great precision from the design of the apparatus, and need not
be measured. This sinewave is high amplitude (at least one volt rms,
matched to the needs of the soundcard) and very high SNR. This will be
done in two channels in parallel. The signals are at the same frequency
but differ in phase. The intent is to extract the phases of these two
sinewaves, the difference in phase being the ultimate output. The phase
of a signal will be extracted by least-squares fitting of a sine function
to the measured data.
And so on. We need to list the tasks, and to use this task list to inform
the experiment design.
The immediate task is actually to evaluate sound cards for their
suitability, preferably without the added cost and complexity of a DDS
LO and mixer.
Once this evaluation is done, using a mixer and a DDS based LO to
generate a beat frequency is the next step.
Eliminating the mixer and DDS allows a greater number of participants at
this stage than would otherwise be the case.
10Hz resolution whilst avoiding phase truncation spurs is impractical
with a DDS chip by itself.
Depending on the DDS and its clock frequency, the frequency spacing of
phase truncation spur free outputs may be as large as several kHz.
A few divide and mix stages will be required to achieve a spur free
resolution of 10Hz.
A DDS chip with higher resolution phase outputs after truncation such as
the AD99XX series are better in this respect than the earlier AD98XX series.
To broaden participation we need to broaden the scope of the project to
include dual mixer system with statistically independent test sources as
well as the more specialised case where the 2 input frequencies differ
only in phase.
-
Evaluate sound cards for suitablility.
Initially use simple less stable sources and follow up with more stable
test sources for the more promising cards.
Need to measure crosstalk, temporal instability of interchannel phase
shift, system noise etc.
-
Develop robust algorithms for phase extraction.
Use the data produced by the less stable sources and that produced by
the more stable sources
-
Repeat testing using a dual mixer system complete with offset LO.
Test frequencies identical to evaluate system noise floor.
-
Repeat testing using a dual mixer system complete with offset LO.
Test frequencies differ to help the effect of residual crosstalk and
other artifacts.
-
Split the project into 2 branches:
A) where mixer inputs differ only by a phase shift to be measured.
Useful for measuring effect on ADEV of various components and their
phase shift tempcos etc.
B) Where the mixer input test sources are statistically independent.
Useful for measuring pairwise source ADEV etc.
Bruce
Joseph M Gwinn wrote:
> Bruce,
>
>
> time-nuts-bounces@febo.com wrote on 12/16/2008 09:05:55 PM:
>
>
>> Joseph M Gwinn wrote:
>>
>>> Bruce,
>>>
>>>
>>> time-nuts-bounces@febo.com wrote on 12/15/2008 05:31:27 PM:
>>>
>>>
>>>
>>>> Joseph M Gwinn wrote:
>>>>
>>>>
>>>>> Bruce,
>>>>>
>>>>>
>>>>> time-nuts-bounces@febo.com wrote on 12/15/2008 04:34:34 PM:
>>>>>
>>>>>
>>>>>
>>> [snip]
>>>
>>>> I'll look into doing this [MDEV and ADEV].
>>>> Real time filtering and decimation may be impractical, in the short
>>>>
> term
>
>>>> at least, as most signal processing libraries only process 16 bit
>>>> samples.
>>>>
>>>> Most real time spectrum analysis programs are similarly afflicted in
>>>> that they only process 16 bit samples.
>>>>
>>>>
>>> I don't see why we would need realtime filtering. Data reduction can
>>>
> be
>
>>> offline, so we ought to be able to use 32-bit or 64-bit arithmetic.
>>>
>>> Given that we will inspect Allan Deviation data in a log-log plot, one
>>>
> can
>
>>> save much processing time by spacing the tau values to be computed
>>> uniformly in log tau. I've played with this in Mathematica, and it
>>>
> does
>
>>> work and yields a large speedup factor. It should also help with
>>>
> Plotter
>
>>> and Win2K limits. One trick is to ensure that one computes each tau
>>>
> value
>
>>> at most once. This check is needed because with close spacing, the
>>>
> round
>
>>> function will yield the same tau values multiple times for small
>>>
> values of
>
>>> tau.
>>>
>>> Joe
>>>
>>>
>>>
>> Joe
>>
>> Real time processing certainly isn't required to characterise the
>>
> performance.
>
>> However some may be tempted to do this, it's probably possible with a
>>
> sufficently fast machine.
>
> If we are looking for thermal effects, with a characteristic timescale of
> tens of minutes to hours, the concept of realtime can be very generous.
>
>
>
>> I was just highlighting a problem with some available signal processing
>> libraries which may have been developed before sound cards with
>> resolutions of more than 16 bits became available.
>> Some (perhaps most) real time spectrum display software also has this
>> problem (eg baudline, Virtins etc).
>>
>
> I would assume that there are newer libraries now, and libraries available
> as source code can be updated and recompiled.
>
> 20 Log[ 2^16 ]= 96 dB. This isn't awful, and we will get the entire
> 16-bit range if the ADC is 24 bits (with ENOB of 19-20 bits) and we scale
> and round the samples properly.
>
> As I think about it, the 16-bit limit must be for embedded signal
> processing code, and math libraries intended for use on ordinary computers
> will be at least 32 bit or 64-bit float, so it should not be difficult to
> come by the necessary code.
>
>
>
>> It isn't necessary to use a pair of mixers and an offset source to
>> characterise the sound card, driving both sound card inputs from the
>> same audio source should suffice.
>>
>
> Yes. One input at a time, with the other input shorted, so we can also
> see the crosstalk.
>
>
>
>> The audio source need not have low ultra low distortion (the IF output
>> signals in a dual mixer system won't have ultra low distortion) or very
>> high frequency stability (the IF output signals in a dual mixer system
>> won't necessarily have particularly high frequency stability).
>>
>
> But ... but ... but ... I thought Time Nuts used only atomic frequency
> refs, and crystals only if oven stabilized.
>
>
If one mixes down a 10MHz source to 100Hz the fractional frequency
instability (of the beat frequency) is magnified by a factor of 1E5 over
that of the 10MHz source.
This assumes that the offset source has significantly lower instability
than the source under test.
In the special case when the offset source and the test source are phase
locked the offset frequency will have much greater stability.
>
>
>> A standard RC audio oscillator with distortion lower than 1% or so
>> should suffice.
>> At least the resultant frequency fluctuations should thoroughly exercise
>> the phase extraction algorithms.
>>
>> Another option would be to low pass filter the output of a divider.
>> Using a sound card to generate the test signal is also possible but it
>> can potentially introduce extraneous noise and other artifacts such as
>> phase truncation spurs.
>>
>
> If one chooses the test frequencies correctly, one can eliminate the
> spurs. The trick is to choose frequencies that lead to DDS tuning words
> that have zeroes in the accumulator bits that are truncated (that is, do
> not make it into the sin/cos lookup table).
>
>
>
This just adds another layer of complexity for little immediate gain.
Making the algorithms robust against small drifts in beat frequency is
more useful in the general case (when 2 different test sources are being
compared) than just assuming that the the beat frequency is very stable
and fixed.
> Step one of planning an experiment is to decide on the objectives. The
> large scale objective is to determine which sound cards are suitable for a
> number of time-related tasks, so we should enumerate and describe these
> tasks.
>
> Task 1. The immediate task is to receive and digitize the sinewave output
> from a mixer, the sinewave being the offset frequency coming out of a DMTD
> apparatus. Offset frequencies will range from 10 Hz to 1 KHz, will be
> known with great precision from the design of the apparatus, and need not
> be measured. This sinewave is high amplitude (at least one volt rms,
> matched to the needs of the soundcard) and very high SNR. This will be
> done in two channels in parallel. The signals are at the same frequency
> but differ in phase. The intent is to extract the phases of these two
> sinewaves, the difference in phase being the ultimate output. The phase
> of a signal will be extracted by least-squares fitting of a sine function
> to the measured data.
>
> And so on. We need to list the tasks, and to use this task list to inform
> the experiment design.
>
>
>
The immediate task is actually to evaluate sound cards for their
suitability, preferably without the added cost and complexity of a DDS
LO and mixer.
Once this evaluation is done, using a mixer and a DDS based LO to
generate a beat frequency is the next step.
Eliminating the mixer and DDS allows a greater number of participants at
this stage than would otherwise be the case.
10Hz resolution whilst avoiding phase truncation spurs is impractical
with a DDS chip by itself.
Depending on the DDS and its clock frequency, the frequency spacing of
phase truncation spur free outputs may be as large as several kHz.
A few divide and mix stages will be required to achieve a spur free
resolution of 10Hz.
A DDS chip with higher resolution phase outputs after truncation such as
the AD99XX series are better in this respect than the earlier AD98XX series.
To broaden participation we need to broaden the scope of the project to
include dual mixer system with statistically independent test sources as
well as the more specialised case where the 2 input frequencies differ
only in phase.
1) Evaluate sound cards for suitablility.
Initially use simple less stable sources and follow up with more stable
test sources for the more promising cards.
Need to measure crosstalk, temporal instability of interchannel phase
shift, system noise etc.
2) Develop robust algorithms for phase extraction.
Use the data produced by the less stable sources and that produced by
the more stable sources
3) Repeat testing using a dual mixer system complete with offset LO.
Test frequencies identical to evaluate system noise floor.
4) Repeat testing using a dual mixer system complete with offset LO.
Test frequencies differ to help the effect of residual crosstalk and
other artifacts.
5) Split the project into 2 branches:
A) where mixer inputs differ only by a phase shift to be measured.
Useful for measuring effect on ADEV of various components and their
phase shift tempcos etc.
B) Where the mixer input test sources are statistically independent.
Useful for measuring pairwise source ADEV etc.
Bruce
PD
Predrag Dukic
Thu, Dec 18, 2008 8:19 PM
Hi, Time -Nuts,
Did anyone try to deliberately allow syncronisation of two
oscillators, by , for example paralleling outputs of two 10811.
I expect to see some benefits from the usual statistics: Phase
noise divided by sqrt of 2 and also decreased amplitude random
frequency jumps.
Aging could also be average of the two.....
Predrag Dukic
Hi, Time -Nuts,
Did anyone try to deliberately allow syncronisation of two
oscillators, by , for example paralleling outputs of two 10811.
I expect to see some benefits from the usual statistics: Phase
noise divided by sqrt of 2 and also decreased amplitude random
frequency jumps.
Aging could also be average of the two.....
Predrag Dukic
RK
Rick Karlquist
Thu, Dec 18, 2008 8:28 PM
Some of Len Cutler's engineers at HP attempted to build
an ensemble of nine 10811 oscillators. It was quite
non-trivial and I'm not sure they ever completed
the project. I doubt whether just letting 10811's
self synchronize would result in satisfactory performance.
Rick Karlquist N6RK
Predrag Dukic wrote:
Hi, Time -Nuts,
Did anyone try to deliberately allow syncronisation of two
oscillators, by , for example paralleling outputs of two 10811.
I expect to see some benefits from the usual statistics: Phase
noise divided by sqrt of 2 and also decreased amplitude random
frequency jumps.
Aging could also be average of the two.....
Predrag Dukic
time-nuts mailing list -- time-nuts@febo.com
To unsubscribe, go to
https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
and follow the instructions there.
Some of Len Cutler's engineers at HP attempted to build
an ensemble of nine 10811 oscillators. It was quite
non-trivial and I'm not sure they ever completed
the project. I doubt whether just letting 10811's
self synchronize would result in satisfactory performance.
Rick Karlquist N6RK
Predrag Dukic wrote:
>
>
> Hi, Time -Nuts,
>
> Did anyone try to deliberately allow syncronisation of two
> oscillators, by , for example paralleling outputs of two 10811.
>
> I expect to see some benefits from the usual statistics: Phase
> noise divided by sqrt of 2 and also decreased amplitude random
> frequency jumps.
>
> Aging could also be average of the two.....
>
> Predrag Dukic
>
>
> _______________________________________________
> time-nuts mailing list -- time-nuts@febo.com
> To unsubscribe, go to
> https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
> and follow the instructions there.
>
>
BG
Bruce Griffiths
Thu, Dec 18, 2008 8:31 PM
Hi, Time -Nuts,
Did anyone try to deliberately allow syncronisation of two
oscillators, by , for example paralleling outputs of two 10811.
I expect to see some benefits from the usual statistics: Phase
noise divided by sqrt of 2 and also decreased amplitude random
frequency jumps.
Aging could also be average of the two.....
Predrag Dukic
Pedrag
First ensure that doing this doesnt overstress the output amplifier cascade.
Then the two oscillators will need to be tuned so that they are very
close in frequency before injection locking can occur.
The higher the reverse isolation of the output amplifier and the higher
the crystal Q the closer the 2 oscillator frequencies will have to be.
The maximum frequency difference tolerable for injection locking to
occur can be estimated from Adlers equation.
Bruce
Predrag Dukic wrote:
> Hi, Time -Nuts,
>
> Did anyone try to deliberately allow syncronisation of two
> oscillators, by , for example paralleling outputs of two 10811.
>
> I expect to see some benefits from the usual statistics: Phase
> noise divided by sqrt of 2 and also decreased amplitude random
> frequency jumps.
>
> Aging could also be average of the two.....
>
> Predrag Dukic
>
>
>
Pedrag
First ensure that doing this doesnt overstress the output amplifier cascade.
Then the two oscillators will need to be tuned so that they are very
close in frequency before injection locking can occur.
The higher the reverse isolation of the output amplifier and the higher
the crystal Q the closer the 2 oscillator frequencies will have to be.
The maximum frequency difference tolerable for injection locking to
occur can be estimated from Adlers equation.
Bruce
BG
Bruce Griffiths
Thu, Dec 18, 2008 8:48 PM
Some of Len Cutler's engineers at HP attempted to build
an ensemble of nine 10811 oscillators. It was quite
non-trivial and I'm not sure they ever completed
the project. I doubt whether just letting 10811's
self synchronize would result in satisfactory performance.
Rick Karlquist N6RK
Predrag Dukic wrote:
Hi, Time -Nuts,
Did anyone try to deliberately allow syncronisation of two
oscillators, by , for example paralleling outputs of two 10811.
I expect to see some benefits from the usual statistics: Phase
noise divided by sqrt of 2 and also decreased amplitude random
frequency jumps.
Aging could also be average of the two.....
Predrag Dukic
Pedrag
You may want to look at:
http://my.ece.ucsb.edu/yorklab/Publications/BioBib/84%20-%20MTT%20May%201997%20Phase%20Noise.pdf
http://repository.kulib.kyoto-u.ac.jp/dspace/bitstream/2433/49985/1/PhysRevLett_98_184101.pdf
to get some idea of the complexities involved in such a scheme if you
intend to reduce the phase noise of an ensemble of mutually coupled
oscillators.
Bruce
Rick Karlquist wrote:
> Some of Len Cutler's engineers at HP attempted to build
> an ensemble of nine 10811 oscillators. It was quite
> non-trivial and I'm not sure they ever completed
> the project. I doubt whether just letting 10811's
> self synchronize would result in satisfactory performance.
>
> Rick Karlquist N6RK
>
>
> Predrag Dukic wrote:
>
>> Hi, Time -Nuts,
>>
>> Did anyone try to deliberately allow syncronisation of two
>> oscillators, by , for example paralleling outputs of two 10811.
>>
>> I expect to see some benefits from the usual statistics: Phase
>> noise divided by sqrt of 2 and also decreased amplitude random
>> frequency jumps.
>>
>> Aging could also be average of the two.....
>>
>> Predrag Dukic
>>
>>
>>
LJ
Lux, James P
Thu, Dec 18, 2008 9:03 PM
-----Original Message-----
From: time-nuts-bounces@febo.com
[mailto:time-nuts-bounces@febo.com] On Behalf Of Predrag Dukic
Sent: Thursday, December 18, 2008 12:20 PM
To: Discussion of precise time and frequency measurement
Subject: [time-nuts] Is oscillator sync always bad?
Hi, Time -Nuts,
Did anyone try to deliberately allow syncronisation of two
oscillators, by , for example paralleling outputs of two 10811.
I expect to see some benefits from the usual statistics: Phase
noise divided by sqrt of 2 and also decreased amplitude random
frequency jumps.
Aging could also be average of the two.....
Predrag Dukic
Allan, et al., did a thing with 8 small oscillators in a ring. I don't recall if they deliberately tried to have them mutually couple, or if it was designed to try and cancel acceleration effects.
James Lux, P.E.
Task Manager, SOMD Software Defined Radios
Flight Communications Systems Section
Jet Propulsion Laboratory
4800 Oak Grove Drive, Mail Stop 161-213
Pasadena, CA, 91109
+1(818)354-2075 phone
+1(818)393-6875 fax
> -----Original Message-----
> From: time-nuts-bounces@febo.com
> [mailto:time-nuts-bounces@febo.com] On Behalf Of Predrag Dukic
> Sent: Thursday, December 18, 2008 12:20 PM
> To: Discussion of precise time and frequency measurement
> Subject: [time-nuts] Is oscillator sync always bad?
>
>
>
> Hi, Time -Nuts,
>
> Did anyone try to deliberately allow syncronisation of two
> oscillators, by , for example paralleling outputs of two 10811.
>
> I expect to see some benefits from the usual statistics: Phase
> noise divided by sqrt of 2 and also decreased amplitude random
> frequency jumps.
>
> Aging could also be average of the two.....
>
> Predrag Dukic
>
>
Allan, et al., did a thing with 8 small oscillators in a ring. I don't recall if they deliberately tried to have them mutually couple, or if it was designed to try and cancel acceleration effects.
James Lux, P.E.
Task Manager, SOMD Software Defined Radios
Flight Communications Systems Section
Jet Propulsion Laboratory
4800 Oak Grove Drive, Mail Stop 161-213
Pasadena, CA, 91109
+1(818)354-2075 phone
+1(818)393-6875 fax
>
LJ
Lux, James P
Thu, Dec 18, 2008 9:14 PM
-----Original Message-----
From: Lux, James P
Sent: Thursday, December 18, 2008 1:04 PM
To: 'Discussion of precise time and frequency measurement'
Subject: RE: [time-nuts] Is oscillator sync always bad?
Allan, et al., did a thing with 8 small oscillators in a
ring. I don't recall if they deliberately tried to have them
mutually couple, or if it was designed to try and cancel
acceleration effects.
> -----Original Message-----
> From: Lux, James P
> Sent: Thursday, December 18, 2008 1:04 PM
> To: 'Discussion of precise time and frequency measurement'
> Subject: RE: [time-nuts] Is oscillator sync always bad?
>
>
>
> Allan, et al., did a thing with 8 small oscillators in a
> ring. I don't recall if they deliberately tried to have them
> mutually couple, or if it was designed to try and cancel
> acceleration effects.
>
>
Found the paper..
Allan, David W., Kusters, John A, and Wheatley, Charles E., III; "CTXO, Clever Time Crystal Oscillator (Clock)", 1999 Joint Meeting EFTF-IEEE IFCS
http://ieeexplore.ieee.org/iel5/6762/18075/00840780.pdf?arnumber=840780 if you have xplore access
PD
Predrag Dukic
Thu, Dec 18, 2008 9:16 PM
Bruce,
I know the math, and possible perils. The main question is still:
Does statistics help? Is it going to be better?
If I do try, I'll use a bunch of 13 MHZ TCXO's, because I have only
3 10811, and more than 30 TCXOs.
Also with a higher number of lower quality oscs improvement could be
easier to see...
Predrag
At 21:31 18.12.2008, you wrote:
Hi, Time -Nuts,
Did anyone try to deliberately allow syncronisation of two
oscillators, by , for example paralleling outputs of two 10811.
I expect to see some benefits from the usual statistics: Phase
noise divided by sqrt of 2 and also decreased amplitude random
frequency jumps.
Aging could also be average of the two.....
Predrag Dukic
Pedrag
First ensure that doing this doesnt overstress the output amplifier cascade.
Then the two oscillators will need to be tuned so that they are very
close in frequency before injection locking can occur.
The higher the reverse isolation of the output amplifier and the higher
the crystal Q the closer the 2 oscillator frequencies will have to be.
The maximum frequency difference tolerable for injection locking to
occur can be estimated from Adlers equation.
Bruce
time-nuts mailing list -- time-nuts@febo.com
To unsubscribe, go to https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
and follow the instructions there.
Bruce,
I know the math, and possible perils. The main question is still:
Does statistics help? Is it going to be better?
If I do try, I'll use a bunch of 13 MHZ TCXO's, because I have only
3 10811, and more than 30 TCXOs.
Also with a higher number of lower quality oscs improvement could be
easier to see...
Predrag
At 21:31 18.12.2008, you wrote:
>Predrag Dukic wrote:
> > Hi, Time -Nuts,
> >
> > Did anyone try to deliberately allow syncronisation of two
> > oscillators, by , for example paralleling outputs of two 10811.
> >
> > I expect to see some benefits from the usual statistics: Phase
> > noise divided by sqrt of 2 and also decreased amplitude random
> > frequency jumps.
> >
> > Aging could also be average of the two.....
> >
> > Predrag Dukic
> >
> >
> >
>Pedrag
>
>First ensure that doing this doesnt overstress the output amplifier cascade.
>Then the two oscillators will need to be tuned so that they are very
>close in frequency before injection locking can occur.
>The higher the reverse isolation of the output amplifier and the higher
>the crystal Q the closer the 2 oscillator frequencies will have to be.
>The maximum frequency difference tolerable for injection locking to
>occur can be estimated from Adlers equation.
>
>Bruce
>
>_______________________________________________
>time-nuts mailing list -- time-nuts@febo.com
>To unsubscribe, go to https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
>and follow the instructions there.
PD
Predrag Dukic
Thu, Dec 18, 2008 9:29 PM
Bruce,
these articles are more or less the answer to my question.
In principle, there is obviously reduction in noise, and the main
concerns are uncoupled frequency difference and phase.
Thanks,
Predrag
At 21:48 18.12.2008, you wrote:
Some of Len Cutler's engineers at HP attempted to build
an ensemble of nine 10811 oscillators. It was quite
non-trivial and I'm not sure they ever completed
the project. I doubt whether just letting 10811's
self synchronize would result in satisfactory performance.
Rick Karlquist N6RK
Predrag Dukic wrote:
Hi, Time -Nuts,
Did anyone try to deliberately allow syncronisation of two
oscillators, by , for example paralleling outputs of two 10811.
I expect to see some benefits from the usual statistics: Phase
noise divided by sqrt of 2 and also decreased amplitude random
frequency jumps.
Aging could also be average of the two.....
Predrag Dukic
Bruce,
these articles are more or less the answer to my question.
In principle, there is obviously reduction in noise, and the main
concerns are uncoupled frequency difference and phase.
Thanks,
Predrag
At 21:48 18.12.2008, you wrote:
>Pedrag
>
>You may want to look at:
>
>http://my.ece.ucsb.edu/yorklab/Publications/BioBib/84%20-%20MTT%20May%201997%20Phase%20Noise.pdf
>
>http://repository.kulib.kyoto-u.ac.jp/dspace/bitstream/2433/49985/1/PhysRevLett_98_184101.pdf
>
>to get some idea of the complexities involved in such a scheme if you
>intend to reduce the phase noise of an ensemble of mutually coupled
>oscillators.
>
>Bruce
>
>Rick Karlquist wrote:
> > Some of Len Cutler's engineers at HP attempted to build
> > an ensemble of nine 10811 oscillators. It was quite
> > non-trivial and I'm not sure they ever completed
> > the project. I doubt whether just letting 10811's
> > self synchronize would result in satisfactory performance.
> >
> > Rick Karlquist N6RK
> >
> >
> > Predrag Dukic wrote:
> >
> >> Hi, Time -Nuts,
> >>
> >> Did anyone try to deliberately allow syncronisation of two
> >> oscillators, by , for example paralleling outputs of two 10811.
> >>
> >> I expect to see some benefits from the usual statistics: Phase
> >> noise divided by sqrt of 2 and also decreased amplitude random
> >> frequency jumps.
> >>
> >> Aging could also be average of the two.....
> >>
> >> Predrag Dukic
> >>
> >>
> >>
>
>
>_______________________________________________
>time-nuts mailing list -- time-nuts@febo.com
>To unsubscribe, go to https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts
>and follow the instructions there.
