I haven't see a nice user's guide for tboltmon. Lots of exciting
opportunities for discovery and enlightenment :)

Setup menu > Position, fill in your known location and hit "save
segment". that should change that dot.

On Wed, Nov 5, 2008 at 3:07 PM, Brooke Clarke brooke@pacific.net wrote:

--
GDB has a 'break' feature; why doesn't it have 'fix' too?

Brooke,

That is covered in my wiki:

http://www.ko4bb.com/dokuwiki/doku.php?id=precision_timing:store_position_on
_trimble_thunderbolt

Courtesy of John Miles :-)

Didier KO4BB

Mine stays yellow, even after completion of self survey.
As mentioned earlier, you need to store it manually.

----- Original Message -----
From: "David C. Partridge" david.partridge@dsl.pipex.com
To: brooke@pacific.net; "'Discussion of precise time and frequency
measurement'" time-nuts@febo.com
Sent: Thursday, November 06, 2008 7:01 AM
Subject: Re: [time-nuts] Thunderbolt Monitor Manual?

The monitor program is described in the ThunderBoltBook 2003 on the Trimble
website:-

http://trl.trimble.com/docushare/dsweb/Get/Document-10001/ThunderBoltBook2003.pdf

On Sat, 8 Nov 2008, Randy wrote:

The best way would be to compare the highest possible frequencies you can
generate with these two sources. I use two 10GHz sources that are each phase
locked to an external 10MHz reference. Then the 10GHz outputs can be compared
using either of these easy methods:

1. look at the DC/IF output of a microwave mixer where the LO and RF ports are
   driven by the two 10GHz sources. Don't overdrive the RF input to a level that
   can burn out your mixer.

2. use a good microwave frequency counter to read one of the 10GHz outputs while
   driving the counter's 10MHz ext ref input with the 10MHz from the other 10MHz
   source. This is very fast but will only give you accuracy readings that are a
   function of the resolution of the counter plus the bounce of the last digit
   owing to sampling and triggering.

3. if you have access to a lab with one or two microwave synthesized signal
   generators, then you can apply the 10MHz sources to the ext ref inputs of each
   of these signal generators and then proceed as in 1) or 2)
   I have done comparison at 26GHz this way so I have a bit more resolution.

73,

Jeffrey Pawlan  WA6KBL

Randy wrote:

Randy, As I have just been doing this, I'll offer my 2 cent's worth.
The problem I found is that the Rb standard, in my case an hp 5065A, has
better short term stability than the gps unit.  So for getting down to
the best accuracy that the Rb unit is capable of requires integration of
some kind over a longish period.    I find using a pen recorder on the
IF output of a mixer, with crude LP filtering on it will permit setting
an Rb standard with confidence down to the 10 e -11 level.

For approximate setting, triggering a double beam scope from your
reference with the time base at it's fastest setting and the x 10
magnifier in,  and the gain on the channel with the unit under test
wound right up to give near vertical zero crossings, will enable you to
see small phase shifts of the unit under test. With the 465B I was using
for example that's 2 nanoseconds per major division  But it gets tiring
watching it for a long time :^)

Fun though!

Dan

----- Original Message -----
From: "Jeffrey Pawlan" jpawlan@pawlan.com
To: hamradio@oz.net; "Discussion of precise time and frequency
measurement" time-nuts@febo.com
Sent: Sunday, November 09, 2008 7:40 AM
Subject: Re: [time-nuts] Checking accuracy of Rubidium standards

Jeffrey-
Details on the 10 GHz sources? Are these commercial or home built?
I've been looking at this kind of scheme on and off for a while now
but other than collecting up some possibly useful ex-commercial
multipliers (10 GHz o/p w/g  assys) I have not progressed any.
DaveB, NZ

This is an interesting thread again.....it may be similar to ones that have
been discussed, but one or two furthur questions occur to me. I have a
Montronics sytem that does comparisons by the multiply and mix process, and
I find (also common to more modern Kethly systems) that the limitation is
around a part in 10^10 where the noise on the phase output makes it not
really usable (without a lot of averaging) being around or in excess of 90
degreees even with a couple of very good OXCOs. How does the 10G comparision
avoid this problem with standard multipliers? I doubt you can go all that
way with low-noise multipliers and have any useful signal left, or have I
missed something. At present I use a phase meter (lock in amps can be quite
good) at the MHz range and datalog the phase drift for several hours. I have
determined that setting "on the nose" is not necessary (for my
applications). It is more useful to know how far a source is "off".
How does the mix down compare with the seemingly more popular "mix down and
timestamp" I understand from previous threads that this has more potential
but might it also be as good even using simpler circuits that the NIST
system.

Thanks for all your efforts inthe background John..... great reading
material !

Alan G3NYK

----- Original Message -----
From: "Jeffrey Pawlan" jpawlan@pawlan.com
To: hamradio@oz.net; "Discussion of precise time and frequency
measurement" time-nuts@febo.com
Sent: Saturday, November 08, 2008 6:40 PM
Subject: Re: [time-nuts] Checking accuracy of Rubidium standards

phase

compared

are

that

while

10MHz

are a

digit

signal

each

https://www.febo.com/cgi-bin/mailman/listinfo/time-nuts

Alan Melia wrote:

Alan

The principal limitation with dual mixer systems is the relatively large
phase shift tempco of the mixers (~ 10ps/C for 10MHz inputs).
If one uses a high end sound card (or equivalent ADC) to timestamp the
beat frequency zero crossings a resolution of better than 1E-12/Tau is
possible when the thermal environment permits.
However a stable low noise offset source frequency is required.
It is also essential to ensure there is sufficient isolation between the
sources being compared to avoid injection locking.

A low end sound card can also be used together with a Collin's style
bandpass limiter/slope amplifier to achieve similar performance if the
slope gain is large enough.

Bruce

Hi,
I have a plan which involves the dividing down of the 10MHz of a GPSDO
and a rubidium (LPRO) to about 1MHz or 100kHZ and applying them to a
XOR or D latch to get a PWM signal that can be averaged for a strip
chart recorder or
12 bit analogue data logger. The DC output gives a range of 5 volts
for one
microsecond or 10 microseconds phase difference and folds back if
this difference is exceeded.
The data from the datalogger is in a format that a spreadsheet can use.
With time and phase measurements I wonder how hard it is to get Allen
variance.
I realise the PWM method requires a low pass filter and this will
prevent short period
variances from being calculated.
cheers, Neville Michie

On 09/11/2008, at 8:43 AM, Alan Melia wrote:

Neville Michie wrote:

Neville

A low pass filter or equivalent means of limiting/defining the
measurement bandwidth is required when measuring ADEV.

The limitation with all such digital phase detectors is the nonlinearity
as the ends of the range are approached.
It is possible to achieve a resolution of around 4E-11/Tau with such
phase detectors if a higher resolution ratiometric ADC is used.

Bruce

J. L. Trantham wrote:

Joe

There is sufficient information available from the sound card samples to
calculate the input signal at any time between 2 samples and in
particular derive the time at which the signal crosses zero.
This is the time stamp for that zero crossing.
The frequency and ADEV of a signal can then be calculated from such a
sequence of time stamps.
However it is necessary to either calibrate the sound card sampling
frequency or lock it to a known frequency.
The method used to interpolate between samples is called WSK (Whittaker
Shannon Kotelnikov) interpolation.

Bruce

Bruce and Joe,

I have been enjoying this discussion too.

2008/11/9 Bruce Griffiths bruce.griffiths@xtra.co.nz:

...

Given Joe's comments, doesn't making ADEV measurements rely on highly
accurately spaced sound card samples and the nature of system based
upon this hardware affect the accuracy of any such based system? Does
the accuracy of the measuring system not have to meet or exceed the
accuracy of the DUT?

73, Steve

Steve Rooke - ZL3TUV & G8KVD
Omnium finis imminet

Steve Rooke wrote:

Steve

When one is measuring the beat frequency between an offset standard and
the DUT, the sound card timebase doesn't have to be more accurate than
the the DUT as it is only measuring the error in the small offset
between the DUT and the offset standard.
However the local standard against which the DUT is being compared does.

For example with a 10MHz DUT and a 10MHz local standard offset by 100Hz
from 10MHz, the sound card only has to measure the  100Hz offset
frequency ( between the DUT and the offset standard) to an accuracy of
1E-7 in order to determine the frequency of the DUT to an accuracy of
1E-12.

When one uses a dual mixer system to compare 2 non offset 10MHz signals,
most of the error contributions from the offset source and ADC sampling
clock are common to both channels and tend to cancel on subtraction.

Bruce

2008/11/9 Bruce Griffiths bruce.griffiths@xtra.co.nz:

So as this is just measuring a beat, there is no requirement for
absolute accuracy in the measuring system, it just needs the reference
frequency to be as accurate as the desired measure of absolute
accuracy. Yes, I can see that.

So 100Hz offset from 10MHz is 1E-5 and if I can measure the 100Hz to
an accuracy of 1E-7 that would give an overall measurement of 1E-12.
So that would mean that the sound card device would somehow have to
sample with at least 1E-7 accuracy. That would mean taking enough
samples of sufficient accuracy to determine this 1E-7 accuracy. A
sound card has a sample frequency of approx 44KHz but to see an offset
of 1E-7 would that not take a fairly long sample time? How would this
affect the ability of such a system to determine ADEV for small tau?

I can see that, assuming both channels are sampled at the same time.

Thanks Bruce.

73, Steve

Steve Rooke - ZL3TUV & G8KVD
Omnium finis imminet

Steve Rooke wrote:

If the signal has sufficiently high SNR the measurement time need not be
very long.
For example if the noise is an additive random Gaussian signal then a 40
dB SNR is equivalent to a period jitter of 0.3%.
For a 100Hz signal the corresponding time jitter is 30us.
If the measurement time is 1 second then the corresponding rms error in
measuring the 100Hz signal is 4.2E-5 or 4.2E-10 of the 10MHz signal.
Thus a measurement time of 420 sec or more would reduce the noise to
1E-12 rms or less.
In practice the beat frequency signal will, with low noise sources, have
a somewhat higher SNR.
Some filtering of the beat signal to reduce the noise will be required.
With a 100Hz beat frequency the shortest value of Tau achievable using
this technique is 10ms.

As long as the requirements of the sampling theorem are satisfied the
choice of sampling rate isnt critical as one can accurately interpolate
between samples to produce as high an effective sampling rate as
required. An effective sampling rate of 10MHz or more is sufficient to
achieve a resolution of 1E-12 or better (at 10MHz) with a 100Hz beat
frequency.
However calculating all the interpolated samples for a 10MHz clock isnt
the most efficient technique since only the zero crossings are of interest.

An alternative technique is to implement a Costas receiver in software.
Such a receiver has both in phase and quadrature outputs allowing the
phase of the receiver output signal to be calculated for each sample.

Bruce

Steve Rooke wrote:

Steve

The effect of the finite resolution of the sound card ADC can be
estimated as follows:
For a 16 bit ADC and a sinusoidal beat frequency signal with 10MHz mixer
inputs a 16 bit ADC has a quantisation noise of about 8.8 ppm of the
amplitude of a full scale sinewave.
This corresponds to a zero crossing measurement jitter of about 140fs
(at 10MHz).
This result is independent of the beat frequency.
The corresponding system noise level due to the finite ADC resolution is
about 1.4E-13/Tau.

This is a gross approximation and the corresponding system noise will be
somewhat larger.
Thus it is advisable to either use an ADC with greater resolution to
achieve such low noise, or to increase the beat frequency signal slew
rate at the zero crossing.
The finite signal bandwidth of the sound card will limit the maximum
slope amplification to around 100X or so.
If one uses a mixer where both the RF and LO ports are saturated then
the beat frequency waveform will be trapezoidal with a slew rate at the
zero crossing about 3X that if the signal were sinusoidal.

Bruce

Joe
J. L. Trantham wrote:

The limitation with using Lissajous figures is that eventually the noise
in the relatively wide oscilloscope bandwidth (due to amplifier noise
and signal phase noise) limits the figure rotation rate that can
reliably discerned. Also the lack of a record of changes limits the
ability to see long term trends/drifts.

However it can be a useful/instructive starting point particularly when
the frequency difference is relatively large but still small enough that
the figure rotation can be seen.

One technique is to use a three cornered hat technique.
If the phase fluctuations of 3 oscillators are statistically
independent, then it is possible to determine the statistical
fluctuations in each of them.

Bruce

Bruce, how does the three cornered hat scheme work?

If I had two LPRO Rubidium oscillators and a TBOLT GPSDO, and I
divided each of them down to
100KHz, then I could compare pairs of them with D latches and record
3 different analogue
signals of phase difference.
One of the LPRO oscillators is in an oven to remove ambient
temperature influence.
If I ran them for several weeks and logged the signals every 10 minutes,
what could I expect to recover from the data and how would I apply
the 3 cornered hat scheme?
I ask this question because this is about where my building program
is taking me.
cheers, Neville Michie

On 10/11/2008, at 1:38 PM, Bruce Griffiths wrote:

Neville Michie wrote:

Neville

When the fluctuations of 3 quantities are independent then comparing 2
of them the individual variances of the add:

VAR(1,2) = VAR(1) + VAR(2)
VAR(1,3) = VAR(1) + VAR(3)
VAR(2,3) = VAR(2)+ VAR(3)

Where
VAR(1,2) denotes the variance of the fluctuations in the difference
between quantities 1 and 2.
VAR(1,3) denotes the variance of the fluctuations in the difference
between quantities 1 and 3.
VAR(2,3) denotes the variance of the fluctuations in the difference
between quantities 2 and 3.
VAR(1) denotes the variance of the quantity 1.
VAR(2) denotes the variance of the quantity 2.
VAR(3) denotes the variance of the quantity 3.

Thus
VAR(1) = (VAR(1,2) + VAR(1,3) - VAR(2,3))/2
VAR(2) = (VAR(1,2) + VAR(2,3) - VAR(1,3))/2
VAR(3) = (VAR(1,3) + VAR(1,3) - VAR(1,2))/2

The same results hold for ADEV (used for characterising the stability of
oscillators as  variance of the phase is divergent for oscillators).

ADEV(1,2) = ADEV(1) + ADEV(2)
ADEV(1,3) = ADEV(1) + ADEV(3)
ADEV(2,3) = ADEV(2) + ADEV(3)

Where
ADEV(1,2) denotes the Allen variance of the fluctuations in the phase
difference between oscillators 1 and 2.
ADEV(1,3) denotes the Allen variance of the fluctuations in the phase
difference between oscillators 1 and 3.
ADEV(2,3) denotes the Allen variance of the fluctuations in the phase
difference between oscillators 2 and 3.
ADEV(1) denotes the Allen variance of the phase fluctuations of
oscillator 1.
ADEV(2) denotes the Allen variance of the phase fluctuations of
oscillator 2.
ADEV(3) denotes the Allen variance of the phase fluctuations of
oscillator 3.

Thus
ADEV(1) = (ADEV(1,2) + ADEV(1,3) - ADEV(2,3))/2
ADEV(2) = (ADEV(1,2) + ADEV(2,3) - ADEV(1,3))/2
ADEV(3) = (ADEV(1,3) + ADEV(1,3) - ADEV(1,2))/2

Note it is essential to measure the relative phase fluctuations between
all 3 oscillator pairs simultaneously.

The limitation is that the oscillators should all have similar ADEV.
If the calculations assign negative values to one or more of the
individual variances then the phase fluctuations for the individual
oscillators may be correlated or at least one of the oscillators may be
much quieter than the others.

Eventually common environmental variations such as temperature pressure
and humidity fluctuations introduce correlations invalidating the above
simplified analysis.
However in this case the theory has been extended to include the effect
of correlations which are adjusted to ensure that the calculated
individual variances are positive definite.

Bruce

Neville Michie wrote:

Neville

If the phase differences are recorded every 10 minutes then the smallest
value of tau for which you can determine AVAR is 10 minutes.

Bruce